JPH098768A - Multipath delay spread measuring instrument - Google Patents

Abstract

PURPOSE: To output the exact absolute value of a reception level in an instrument for measuring the transmitting state of a signal transmission system by using spread spectrum. CONSTITUTION: This measuring instrument for a delayed wave due to the multipath of a radio wave propagation path in digital communication is provided with a signal reception part 21 for receiving a spread spectrum RF signal, a PN signal generating part 24, a mixer 25 for extracting a correlative waveform signal from a reception side PN signal and the spread spectrum RF signal by sliding correlation, an amplitude information arithmetic means 31 for calculating an amplitude information signal from the correlative waveform signal, a temperature measuring means 42 for measuring the temperature of the measuring instrument inside the main body of a reception part, a correction data preserving part 43 for preserving the correction data of a reception level containing temperature characteristics, and a correction processing part 44 for calculating the absolute value of the reception level based on the temperature inside the main body of the reception part, the amplitude information signal and the correction data and outputting the absolute value.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a multipath delay spread measuring device used for designing a service area of digital mobile communication.

[0002]

2. Description of the Related Art Mobile communication such as mobile phones is realized by wireless communication between a base station and a terminal. To actually perform such mobile communication, it is necessary to determine the service area of each base station.

The reason why each of these conditions is different in each service area is that the transmission signal is reflected by a shield / reflector such as a building or a house and is transmitted to the receiving side as a multiple wave passing through a multipath. Because it is done.

However, in the conventional analog system,
Even if the transmission signal is a multiple wave, deterioration of the transmission quality does not pose a problem so much, so that the effective area (service area) design of the base station by measuring the electric field strength of the transmission signal is relatively easy.

However, in recent years, in mobile communication, with the background of efficient use of frequency and digitization of original data,
The transition to digital systems is progressing rapidly. When such a digital system is used, a direct wave (fundamental wave) transmitted from the transmitting side to the receiving side without passing through the shield is delayed by a delayed wave (reflected wave) transmitted by a reflector or the like. The permissible conditions for the effects of waves will become severe.

In digital communication, when the so-called eye pattern opening becomes narrow, the bit error occurrence rate at the time of demodulating into digital data increases and the transmission quality deteriorates. For example, the delay of the reflected wave with respect to the cycle of the transmission speed. 30
It is said that the rate of occurrence of bit errors exceeds the allowable value when it exceeds%. Further, the bit error occurrence rate changes depending on the intensity ratio of the delayed wave to the direct wave and the phase difference.

Therefore, in digital mobile communication, it is necessary to accurately measure the delay time, intensity ratio, phase difference, etc. of the delayed wave with respect to the direct wave, and use these results to determine the service area, transmission rate, etc. You can

For this measurement, it is possible to use a modulated wave with a single pulse having a high transmission power, but it is actually difficult to realize due to regulations of the Radio Law. Therefore, in a device for measuring such a multipath delayed wave, a spread spectrum method is used which can reduce the intensity of a single frequency component by spreading the radio field intensity of a transmission signal in the frequency direction.

FIG. 7 is a diagram for explaining spectrum spreading and despreading of a transmission signal. First, in order to create a spread spectrum signal, an RF signal having a certain frequency fr and having a certain intensity is spread using a signal over a wide frequency band, specifically, a pseudo random code signal (PN signal). Let

On the other hand, when the spread spectrum signal thus created is despread with the PN signal of the same code string used for spreading on the receiving side, the RF signal of frequency fr is reproduced.

Since this spread spectrum signal is a low-strength signal over a wide frequency band, if it is used as a transmission wave, it will not easily interfere with other signals in the same space and spread over a wide frequency band. R from the signal
Since the F signal is reproduced, it is unlikely to be affected by other signals or thermal noise.

By the way, the PN signal is one in which the same pseudo-random code sequence is repeatedly output at a constant cycle. Therefore, in a device that measures a multipath delayed wave, when performing despreading, the code rate of the PN signal on the receiving side is slightly changed to change the code rate of the PN signal on the transmitting side. Each time it is repeated a fixed number of times, both PN signals are synchronized and the RF signal is extracted as a correlation coefficient of both PN signals.

The number of repetitions of the pseudo-random code sequence in this case is called a K factor. Also, despreading is performed with PN signals having slightly different code rates, and for each K factor,
A method that allows the RF signal to be extracted as the correlation coefficient of the PN signal is called a sliding correlation method.

FIG. 8 is an explanatory diagram of sliding correlation. In FIG. 8, the transmission side PN signal and the reception side P
The code rate is slightly different from that of the N signal. Therefore, the code string of the received PN signal is slightly shifted (slipped) from the transmitted PN signal.

Considering the correlation between both PN signals, the correlation coefficient is small when the phases are not synchronized. However, as shown in FIG. 8, the start of the pseudo-random code sequence eventually coincides with the sliding of the PN signal on the receiving side, so the correlation coefficient becomes maximum at this point, and the triangular wave with this point as the apex becomes the correlation waveform. It is formed.

For example, if the transmission side PN signal code rate is 30.
01 Mbps, PN signal code rate on the receiving side is 30M
If bps, factor K = 30 / (30.01-3
0) = 3000, and the above correlation waveform is obtained every 3000 times of the code sequence.

The above description is a correlation waveform of only the direct wave when the RF signal, which also serves as the propagation wave of the spread spectrum signal, is directly transmitted from the transmitter to the receiver. On the other hand, when there is a delayed wave, the delayed wave is transmitted with a time delay with respect to the direct wave, so that the start of the code string of the PN signal is delayed by the delay time.

Here, if the difference between the PN signal code rates on the transmitting and receiving sides is sufficiently small, that is, if the K factor is sufficiently large, the code string of the same reflected wave as the code string from which the correlation waveform is obtained in the direct wave is obtained. , And the start position of the PN code string on the receiving side after the delay time matches. Therefore, at this point, which is delayed by the delay time from the direct wave, the correlation waveform of the delayed wave is obtained.

Further, the time resolution for detecting the delay time is when the delayed wave correlation waveform is detected in the next code string of the direct wave correlation waveform. Therefore, for example, in the case of the above example, the delay resolution is = 1/30 Mbps = 33 ns.
This is 10 m when converted into distance using the propagation velocity of radio waves (300 m / μs). Therefore, in order to increase the delay resolution, the code rate of the PN signal should be increased. However, the PN signal code rate in the above-described delayed wave measurement method is R that the code string modulates.
Limited by the frequency of the F or IF signal.

The reason is that in this case, the RF or IF signal is BPSK modulated by the code string of the PN signal. FIG. 9 is a diagram showing BPSK modulation.

As shown in FIG. 9, for the RF signal, PN
When the sign of is changed between "1" and "0", the phase of the RF signal is inverted and becomes a modulation signal. Therefore, it is understood that the frequency of the RF (or IF) signal must be 3 to 5 times higher than the code rate of the PN signal when producing such a modulated signal.

An apparatus for measuring a multipath delay wave, that is, a multipath delay spread measuring apparatus is constructed by using the techniques such as the spread spectrum, sliding correlation, and BPSK modulation described above. It is a block diagram which shows the receiving part of a path delay spread measuring device.

In FIG. 10, RF means that the RF signal of about 2 GHz shown in FIGS.
The received signal RF is shown in which the transmitted signal is spread by the PN signal of 0.01 Mbps and received by the antenna 59.

In the receiving section of this multipath delay spread measuring apparatus, the spread spectrum received signal RF and the local oscillation signal LO are input to the mixer 51 and converted into an intermediate frequency signal IFa of 140 MHz.

On the other hand, the 30 Mbps PN signal output from the PN signal generator 60 and the 140 MHz intermediate frequency signal IFb are BPSK modulated in the mixer 52 and output as the intermediate frequency signal IFc.

The intermediate frequency signal IFc is further fed to the phase shifter 5
In 3, it is decomposed into the in-phase component and the quadrature component, which are input to the mixer 54 and the mixer 55, respectively. On the other hand, the intermediate frequency signal IFa from the received signal RF is also received by the mixer 54.
Is input to the mixer 55, spectrum despreading is performed in each of the mixers 54 and 55, and at the same time, a correlation waveform is extracted.

The outputs from the mixers 54 and 55 through the low-pass filters 56 and 57 are the in-phase component correlation waveform I and the quadrature component correlation waveform Q, respectively, and these I and Q correlation waveforms are input to the calculator 58. . In the calculator 58, the absolute amplitude value E (= SQRT (I ² + Q
² )) The logarithmically calculated value is calculated and taken out as the amplitude information logE.

Here, each output when the received signal R includes a delayed wave such as a reflected wave is as shown in FIG. FIG. 11 is an explanatory diagram showing each correlation waveform and phase information calculation extracted from a conventional multipath delay spread measuring apparatus.

FIG. 11A shows the in-phase component correlation waveform I, the quadrature component correlation waveform Q, and the absolute amplitude value E of the correlation waveform when the direct wave, the first delayed wave, and the second delayed wave are included. Shows. Here, I1, Q1, and E1 are each correlation waveform of the direct wave, I2, Q2, and E2 are each correlation waveform of the first delayed wave,
I3, Q3 and E3 are the respective correlation waveforms of the second delayed wave.

Further, the phase information (phase angle θ) of each correlation waveform can be obtained from the I and Q correlation waveforms. FIG.
(B) is the phase angle θ of each correlation waveform on the I and Q planes.
Is shown. That is, the phase angle θ is calculated by taking the arc tangent of the I / Q ratio.

[0031]

According to the above-mentioned conventional multipath delay spread measuring apparatus, FIG.
The delay time can be calculated from the output result of (a), and the relative intensity ratio between the delayed waves can also be calculated.

However, in digital mobile communication, in order to prevent the occurrence of bit errors as described above,
Various conditions must be set with a degree of accuracy higher than that of analog mobile communication, and for that purpose, more accurate measurement results are required.

For example, in digital mobile communication, FIG.
As shown in (a), T is a method of transmitting and receiving a plurality of communication channels in one carrier frequency in a time division manner.
DMA communication is often performed.

In this method, when a strong reflected wave (delayed wave) is generated, the fundamental wave (direct wave) is affected as shown in FIG. 12 (b) (the shaded portion of the fundamental wave in the figure). Is the part interfered by the reflected wave), communication is disturbed.

Therefore, how much multiplexing occurs in a certain area is investigated in advance, and the intensity of the delayed wave is not the relative intensity ratio with the direct wave, but the exact absolute intensity value is accurate. It becomes necessary to measure at a position on a different time axis.

However, in the conventional multipath delay spread measuring apparatus shown in FIG. 10, for example, the amplitude information logE cannot be directly used as an absolute reception level because of the temperature characteristics and output characteristics of the analog elements constituting the circuit. . That is, these outputs need to be calibrated, and there is a problem that the absolute intensity value of the detected correlation waveform cannot be accurately measured under the current state.

In addition, in particular, in the device of FIG. 10, in the two mixers 54 and 55 used for despreading, the output temperature drifts due to the difference in the temperature characteristics of the forward voltage of the diodes inside, and the temperature change causes There is also a problem that the DC drift of the IQ signal occurs and the absolute value level of the amplitude information and phase information output becomes inaccurate.

The present invention has been made in consideration of such a situation, and in a device for measuring a transmission state of a signal transmission system transmitted via multipaths using spread spectrum, a direct signal of a transmission signal is transmitted. , It is an object of the present invention to provide a multipath delay spread measuring device capable of outputting an accurate absolute value of each reception level of a delayed wave at that time position.

[0039]

In order to solve the above-mentioned problems, the invention corresponding to claim 1 is a delay wave measuring device by multipath of a radio wave propagation path in digital communication, wherein an RF signal is a PN code. Spread spectrum R modulated by
A signal receiving unit that receives the F signal, a PN signal generating unit that generates a receiving PN signal in the same sequence as the PN code, and a receiving PN
A mixer that mixes a signal containing a signal with a spread spectrum RF signal received by the signal receiving unit, despreads by sliding correlation, and extracts a correlation waveform signal; and an amplitude that outputs an amplitude information signal by calculation on the correlation waveform signal. In the information calculating means, the temperature measuring means for measuring the temperature in the receiver main body of the measuring device including at least the mixer and the amplitude information calculator, and the signal receiver corresponding to the temperature in the receiver main body and the amplitude information signal. The calibration data storage unit that stores the absolute value of the reception level as calibration data, and the temperature and amplitude information signals inside the reception unit main body are input, and these input data and the calibration data read from the calibration data storage unit are input. The multipath delay spread measuring device is provided with a calibration processing unit for calculating and outputting the absolute value of the reception level in the signal receiving unit based on the above.

The invention according to claim 2 is an apparatus for measuring a delayed wave by multipath of a radio wave propagation path in digital mobile communication, wherein a spread spectrum RF signal obtained by modulating a first RF signal with a PN code is used. A signal receiving unit for receiving and a PN for generating a receiving side PN signal of the same sequence as the PN code
A signal generator, a first mixer that mixes the spread spectrum RF signal received by the signal receiver and the reception side PN signal, and despreads by sliding correlation, and a second mixer from the output of the first mixer. A bandpass filter for extracting an RF signal, a first signal generator for generating a local oscillation signal, a second RF signal for mixing the local oscillation signal, and a second mixing for outputting a first intermediate frequency signal. A DC component removing unit that removes a DC component of the first intermediate frequency signal, a logarithmic amplifier that outputs an amplitude information signal and a limiter signal from the output of the DC component removing unit, and a first intermediate frequency signal. A second signal generator for generating a second intermediate frequency signal of the same frequency;
A phase detector that outputs the phase information of the second RF signal from the limiter signal and the second intermediate frequency signal, a temperature measuring unit that includes at least a logarithmic amplifier, and measures the temperature inside the main body of the measuring unit, and a receiving unit. A calibration data storage unit that stores the absolute value of the reception level in the signal reception unit corresponding to the temperature and amplitude information signal in the main body as calibration data, and the temperature and amplitude information signal in the reception unit body are input. A multipath delay spread measuring apparatus comprising: a calibration processing unit that calculates an absolute value of a reception level in a signal receiving unit based on input data and calibration data read from a calibration data storage unit and outputs the absolute value.

[0041]

Therefore, first, in the multipath delay spread measuring apparatus of the invention according to claim 1, the signal receiving section receives the spread spectrum RF signal obtained by modulating the RF signal with the PN code.

Next, the PN signal generator generates a receiving side PN signal in the same sequence as the PN code. Further, the mixer includes a signal including the PN signal on the receiving side and the spread spectrum RF signal received by the signal receiving unit, and despreads by sliding correlation to extract a correlation waveform signal.

Further, the amplitude information calculation means outputs the amplitude information signal by the calculation on the correlation waveform signal. The amplitude information calculating means may be, for example, a log amp, or an arithmetic unit that calculates from the in-phase and quadrature correlation information.

On the other hand, the temperature measuring means measures the temperature in the main body of the receiving portion of the measuring device including at least the mixer and the amplitude information calculating means. Further, the calibration data storage unit stores the absolute value of the reception level in the signal receiving unit corresponding to the temperature and the amplitude information signal in the receiving unit body as the calibration data.

Then, the calibration processing section calculates the absolute value of the reception level in the signal receiving section based on the temperature and amplitude information signals in the receiving section body and the calibration data read from the calibration data storage section, Is output.

Further, in the multipath delay spread measuring apparatus of the invention according to claim 2, the signal receiving section receives the spread spectrum RF signal obtained by modulating the first RF signal with the PN code.

Further, the PN signal generating section generates a receiving side PN signal of the same sequence as the PN code. And the first
The mixer of (1) mixes the spread spectrum RF signal received by the signal receiving unit with the PN signal on the receiving side, and despreads them by sliding correlation.

Further, a band pass filter (pass band filter) is used to output a second RF signal from the output of the first mixer.
The signal is extracted. On the other hand, the first signal generator generates a local oscillation signal.

Next, the second RF is applied by the second mixer.
The signal and the local oscillation signal are mixed and the first intermediate frequency signal is output. Further, the DC component removal unit removes the DC component of the first intermediate frequency signal.

Then, the logarithmic amplifier outputs the amplitude information signal and the limiter signal from the output of the DC component removing section. On the other hand, the second signal generating section generates the second intermediate frequency signal having the same frequency as the first intermediate frequency signal.

Next, the phase detector outputs the phase information of the second RF signal from the limiter signal and the second intermediate frequency signal. On the other hand, the temperature inside the receiver main body of the measuring device including at least a logarithmic amplifier is measured by the temperature measuring means.

In the calibration data storage unit, the absolute value of the reception level in the signal receiving unit corresponding to the temperature and the amplitude information signal in the receiving unit main body is stored as the calibration data. Then, the calibration processing unit calculates and outputs the absolute value of the reception level in the signal receiving unit based on the temperature and amplitude information signal in the receiving unit body and the calibration data read from the calibration data storage unit. .

[0053]

Embodiments of the present invention will be described below. (First Embodiment) FIG. 1 is a block diagram showing a multipath delay spread measuring apparatus according to the first embodiment of the present invention.

This multipath delay spread measuring apparatus comprises a transmitter 10 and a receiver 20. In the transmitter 11, for example, a 10 MHz reference signal Ref
Is output from the oscillator 12 which is a rubidium atomic oscillator, and the reference signal Ref is a carrier signal SG of 2.2 GHz.
It is input to the signal generator 13 that generates 1 and the signal generator 14. For example, 1 output from the signal generator 14
The signal SG2 of 50.05 MHz is supplied to the PN signal generator 15
9 with a code rate of 150.05 Mbps
It is output as the PN signal PN1 of the stage M series.

Then, the carrier signal SG1 and the PN signal P
N1 and the spectrum is spread by the mixer 16, and BPSK
The modulated transmission signal RF is output from the transmission antenna 17.

On the other hand, the receiving section 20 includes a receiving antenna 21.
An oscillator 22, a first signal generator 23, a PN signal generator 24, a first mixer 25, a bandpass filter 26, a second signal generator 27, and a second mixer 28.
A band pass filter 29, a DC component removing unit 30,
Logarithmic amplifier 31, third signal generator 32, phase detector 33, A / D converter 41, temperature sensor 42, calibration data storage unit 43, calibration data processing unit 44, and data display unit. And 45.

In addition, the transmitting unit 10 and the receiving unit 20 have the same configuration as shown in FIG.
Is provided with measuring terminals and switches (not shown) for acquiring each calibration data, which will be described later. The receiving antenna 21 is
A transmission signal propagated via a multipath is received as a reception signal RF by being multiplexed by a shield, a reflector, etc. on the way.

The oscillator 22 comprises a rubidium atomic oscillator and outputs a reference signal Ref of 10 MHz. The first signal generator 23 outputs a signal SG3 of 150 MHz based on the reference signal Ref output from the oscillator 22.

The PN signal generator 24 has a code rate of 150 based on the signal SG3 output from the first signal generator 23.
It outputs a PN signal PN2 of Mbps. Here, this P
The N signal PN2 is the PN used for spread spectrum in the transmission unit 10 except that the code rate is slightly slow.
It is the same as the signal PN1 and has the same code string.

The first mixer 25 mixes the received signal RF received by the antenna 21 with the PN signal PN2 output from the PN signal generator 24, despreads the spectrum by sliding correlation, and uses the bandpass filter 26. A correlation waveform RF1 is output via the. At this time, the K factor is K = 150 / (150.05−150) = 300.
0.

The second signal generator 27 uses, for example, 2189.
It outputs a local oscillation signal LO of 3 MHz. The second mixer 28 mixes the correlation waveform RF1 output from the first mixer 25 and passed through the bandpass filter 26 with the local oscillation signal LO output from the second signal generator 27,
For example, the first intermediate frequency signal IF1 of 10.7 MHz is output. This enables handling with a logarithmic amplifier or the like.

The bandpass filter 29 band-limits the output value of the second mixer 28 and outputs it as the first intermediate frequency signal IF1 of 10.7 MHz. DC component removal unit 30
Is composed of a capacitor 30a, removes the DC component of the first intermediate frequency signal IF1 and outputs it. Therefore, even if a DC drift may occur up to this point, the DC component is removed from the output value of the DC component removing unit 30.

The logarithmic amplifier 31 envelope-detects the first intermediate frequency signal IF1 output from the DC component removing unit 30, logarithmically converts the detected correlation waveform, and outputs it as amplitude information logE.

Further, the logarithmic amplifier 31 is provided with a limiter output section 31a. The limiter output unit 31a
The input first intermediate frequency signal IF1 is output as a limiter output signal lim having a constant amplitude. The limiter output signal lim is converted into a wave having a constant amplitude in the correlation detection range in a portion in which the correlation waveform is detected, regardless of the envelope shape, and becomes a noise in a constant amplitude in the portion in which the correlation waveform is not detected. .

The third signal generator 32 outputs a second intermediate frequency signal IF2 of 10.7 MHz based on the reference signal Ref output from the oscillator 22. Phase detector 3
3 includes a 90 ° phase shifter 34, an in-phase mixer 35, a quadrature-phase mixer 36, and a calculator 37.

The 90 ° phase shifter 34 includes the third signal generator 3
The second intermediate frequency signal IF2 output from 2 is decomposed into the in-phase component and the quadrature component, and the in-phase intermediate frequency signal I
It outputs F0 ° and the quadrature intermediate frequency signal IF90 °.

The in-phase mixer 35 cooperates with a low-pass filter (not shown) provided behind it, and outputs the limiter output signal lim output from the limiter output unit 31a and the 90 ° phase shifter 34. In-phase intermediate frequency signal I
The average value of the product with F0 ° is calculated, and the in-phase information cos θ regarding the correlation waveform is output.

The quadrature mixer 36 cooperates with a low-pass filter (not shown) provided behind the quadrature mixer 36 to output the limiter output signal lim output from the limiter output section 31a.
And the quadrature intermediate frequency signal IF 90 ° output from the 90 ° phase shifter 34 are averaged to output quadrature phase information sin θ regarding the correlation waveform.

The calculator 37 outputs the in-phase phase information cos θ output from the in-phase mixer 35 and the quadrature-phase mixer 3.
The phase information θ is calculated from the arc tangent of the ratio with the quadrature phase information sin θ output from 6 and output.

The A / D converter 41 converts the amplitude information logE output from the logarithmic amplifier 31 into a digital quantity.
The temperature sensor 42 measures the internal temperature of the receiver 20 and outputs it. The output of the temperature sensor 42 is provided so that the temperature of the logarithmic amplifier 31 is reflected.

The calibration data storage unit 43 is composed of an EEPROM and stores the reception level data of the reception signal RF for the output of the amplitude information logE shown in FIG. FIG. 2 is a graph showing the absolute reception level (dBm) of the reception signal RF with respect to the amplitude information logE voltage value (V).

The data shown in FIG. 2 is the absolute reception level (dB) of the reception signal RF with respect to the amplitude information logE voltage value (V) from the logarithmic amplifier 31 for each temperature in the receiving section 20.
The calibration data indicating m), which is actually stored in the calibration data storage unit 43, is each measurement point on the graph.

This calibration data is measured as described below. FIG. 3 is a diagram showing a connection configuration of the transmitting unit 10, the receiving unit 20, and each measuring device at the time of measuring the calibration data.

In FIG. 3, the output of the mixer 15 is input to the attenuator 46 from the measuring terminal P1. The attenuator 46 adjusts the transmission signal RF from the mixer 15 to a required level and then outputs it. The switch SW2 is connected to the output terminal of the attenuator 46, and the attenuator 46 of the switch SW2 is connected.
One end of the other end is connected to the measuring terminal P3 of the receiving unit 20, and the other end is connected to the power meter 47.

The transmission signal RF adjusted to the required level and output from the attenuator 46 is connected to the power meter 47 by the switch SW2, and the signal output is measured as the reception level (dBm) of the reception signal RF. Then, the temperature setting described later is performed, the switch SW2 is connected to the measuring terminal P3 of the receiving unit 20, and the signal output is the received signal RF.
Is input to the receiving unit 20.

On the other hand, the PN signal generator 14 in the transmitter 10
The PN signal PN1 from is output via the measuring terminal P2, and this output signal is input to the receiving unit 20 via the signal line 1 and the measuring terminal P4. Further, a switch SW1 for switching between the PN signal PN1 from the measuring terminal P4 and the PN signal PN2 from the PN signal generator 24 and inputting it to the first mixer 25 is provided inside the receiving unit 20. .

When measuring the calibration data, the measurement terminals P2 and
The signal line 1 is provided between the measuring terminals P4 and the switch SW
1 is switched so that the PN signal PN1 from the measuring terminal P4 is input to the first mixer 25.

That is, if a normal signal is input to obtain the calibration data of the receiving section 20, an output cannot be obtained.
The calibration data can be measured only when the above configuration is provided.

The amplitude information logE voltage value (V) is output from the logarithmic amplifier 31 by each of the above signal inputs. At this time, the receiver 20 must be set to the measurement temperature.

That is, the temperature inside the receiver 20 is adjusted by the temperature heating / cooling means (not shown). In addition, the output of the temperature sensor 42 is received by the receiving unit 20 via the measuring terminal P5.
It is input to the temperature display device 49 connected to the outside.

When the temperature inside the receiver 20 indicated by the temperature display device 49 reaches the temperature for obtaining the calibration data, the power meter 47
Output, that is, the absolute reception level of the received signal RF (dB
m) is measured. Then, the switch SW2 is switched, and the amplitude information logE voltage value (V) is read from the logarithmic amplifier 31 by the voltmeter 48 connected to the outside of the receiving unit 20 via the measuring terminal P6.

Hereinafter, the temperature condition and the output condition are changed and the calibration data shown in FIG. 2 is measured by the same procedure. The data stored in the calibration data storage unit 43 is the calibration data point under each condition thus obtained. In this calibration measurement, the PN signal PN1 input to the first mixer 25 is Transmit signal RF at 10
PN signal PN1 used to spread spectrum
Therefore, the parameters such as the code rate and the phase required to extract the correlation waveform during despreading match, the correlation waveform is reliably extracted, and the level calibration measurement can be reliably performed.

In FIG. 1, the calibration data processing unit 44 has the amplitude information logE output from the A / D converter 41.
And the internal temperature of the receiver 20 measured by the temperature sensor 42 are input.

The calibration data processing unit 44 reads the above-mentioned measured calibration data from the calibration data storage unit 43, and calculates the absolute value level of the reception signal RF based on the amplitude information logE, the internal temperature of the receiver 20 and the calibration data. . Also,
The calibration data processing unit 44 receives the input amplitude information log
E, when the internal temperature of the receiver 20 does not match the calibration data point, linear interpolation is performed between the calibration data points to calculate the absolute value level of the reception signal RF.

The data display unit 45 is the calibration data processing unit 4
The absolute reception level of the reception signal RF calculated in 4 is displayed for each of the direct wave and each delayed wave in the amplitude information logE. Further, for the direct wave indicating the absolute reception level and each delayed wave, the intensity ratio between the direct wave and the delayed wave, the delay time, and the phase information θ output from the calculator 37,
Displays the phase difference etc.

The amplitude information calculating means according to claim 1 is composed of the logarithmic amplifier 31, and the temperature measuring means is composed of the temperature sensor 42. Further, the first signal generator according to claim 2 is configured by, for example, the second signal generator 27, and the second signal generator is configured by, for example, the third signal generator 32. ing.

Next, the operation of the multipath delay spread measuring apparatus of the present embodiment constructed as above will be described. First, in the transmitter 10, the carrier signal S
G1 and PN signal PN1 are BPSK-modulated spread spectrum signals having a center frequency of 2200 MHz, for example, and are output as transmission signals RF.

Next, this transmission signal RF is multiplexed as a result of reflection or the like during signal propagation, and is received by the receiving section 20 as a reception signal RF through a multipath. Therefore, the reception signal RF includes a direct wave in which the output of the transmission unit 10 is directly received, and a delayed wave such as a reflected wave which is reflected on the way and is received with a delay from the direct wave.

FIG. 4 shows an operation in which the received signal RF (RF input signal) is output as a logE waveform which is amplitude information.
A description will be given using (a) and (b). FIG. 4A is an explanatory diagram showing the process until the received signal RF is converted into the first intermediate frequency IF1.

First, the received signal RF (RF input signal) is
Although the spread spectrum signal has a center frequency of 2200 MHz, the PN signal PN2 is supplied to the first mixer 25.
Inverse spreading is performed by mixing with, and the correlation waveform RF1 is extracted. In normal despreading, that is, PN signals PN1 and PN
If 2 is the same, the signal is demodulated over the entire time domain, but in the case of the present embodiment, the sliding correlation in which the code rates of both PN signals are slightly changed is used, so the extracted waveform is It consists of correlation values for both code strings.

The correlation waveform RF1 extracted at this time is shown in FIG. 4 (b). In this example, the direct wave D
W and a delayed wave RW delayed therefrom are shown. Next, in FIG. 4A, the correlation waveform RF1 that has passed through the bandpass filter 26 is a signal of 2200 MHz, which cannot be handled as it is by the logarithmic amplifier 31, and therefore is output from the second signal generator 27. Is down-converted by the local oscillation signal LO. As a result, the correlation waveform becomes the first intermediate frequency signal IF1 having a frequency of 10.7 MHz.

FIG. 4B shows the first intermediate frequency signal IF1 at this time. The DC component of the first intermediate frequency signal IF1 is removed by the capacitor 30a of the DC component removing unit 30, and the logarithmic amplifier 31 outputs the logE waveform as amplitude information as shown in FIG. 4B. .

This amplitude information logE is supplied to the A / D converter 4
It is A / D converted at 1 and input to the calibration processing unit 44.
The amplitude information logE converted into a digital amount is converted into an absolute reception level in the calibration processing unit 44 based on the calibration data read from the calibration data storage unit 43. If the temperature condition and the logE output voltage condition at this time match the condition of the calibration data point stored in the calibration data storage unit 43, the conversion can be performed as it is, but in reality, these conditions do not match. Often not.

If they do not match, interpolation is performed using the calibration data points before and after. FIG. 5 is a diagram illustrating interpolation using calibration data points. As shown in FIG. 5, the internal temperature of the receiver 20 is 55 ° C., the calibration data points are at 10 ° C. intervals,
That is, when only data of 50 ° C. and 60 ° C. is included, for example, calibration data corresponding to the amplitude information logE = 3.55 V is calculated by performing linear interpolation as shown in the equation (1).

Calibration data (55 ° C.) = (53.6-53.1) / 10 (55-50) +53.1 (1) = 53.35 logE When the output voltage is 3.55 V, the reception level is 53.35 dBm as it is. And it is sufficient.

However, since the calibration data points are at 0.05V intervals for the logE output voltage, for example, lo
When the gE output voltage takes a value between 3.55V and 3.60V, the temperature is 55 ° C in the same manner as calculated by the equation (1).
Calculate the calibration data of 3.60V in. And
Based on the calibration data of 3.55 V at 55 ° C. and the calibration data of 3.60 V at 55 ° C., linear interpolation similar to the equation (1) is performed to calculate the reception level for the logE output voltage.

In this way, the amplitude information logE is converted into the reception level as the absolute level displayed in the digi-bell display and input to the data display section 45. On the other hand, the first intermediate frequency signal IF1 is converted into a limiter output lim by the limiter output unit 31a in the logarithmic amplifier 31 and input to the phase detector 33.

The second intermediate frequency signal IF2 output from the third signal generator is decomposed into the in-phase intermediate frequency signal IF0 ° and the quadrature intermediate frequency signal IF90 ° by the 90 ° phase shifter 34. .

In the in-phase mixer 35 and the quadrature-phase mixer 36, the average value of the products of the limiter output lim and the in-phase intermediate frequency signal IF0 °, the limiter output lim and the quadrature intermediate frequency signal IF90 °, is calculated.
In-phase phase information cos θ and quadrature phase information sin θ are output from the in-phase mixer 35 and the quadrature-phase mixer 36, respectively.

Further, in the calculator 37, the phase angle θ is calculated from the composition of the in-phase information cos θ and the quadrature information sin θ. That is, θ = tan ⁻¹ (sin θ /
cos θ). Further, the in-phase waveform output I is I =
It can be calculated by Ecos θ, and the quadrature waveform output Q
Can be calculated by Q = Ecos θ.

Then, this phase angle θ is determined by the data display unit 4
5 is input, and in the data display section, the above-mentioned lo
Using the reception level for the gE output voltage, the direct wave D
The delay time, intensity ratio, phase difference, etc. between W and the delayed wave RW are calculated and displayed.

Next, the calculation of the delay time, the intensity ratio and the phase difference between the direct wave DW and the delayed wave RW will be described with reference to FIG. 4 (b). In FIG. 4B, the direct wave DW is detected at time t1, and the delayed wave RW is subsequently detected at time t2.

Here, the delay time of the delayed wave RW with respect to the direct wave DW is calculated by t2-t1. Also, the direct wave D
The intensity ratio between W and the delayed wave RW is calculated by (absolute reception level corresponding to logE2) / (absolute reception level corresponding to logE1). In addition, direct wave DW and delayed wave RW
The phase difference from 1 is calculated as θ2-θ1.

Since the intensity ratio at this time is calculated by the absolute reception level, it is an absolute value, not a relative value. Then, the data display unit 45 displays and outputs the delay time, the intensity ratio, and the phase difference between the direct wave DW and the delayed wave RW.

Further, the data display section 45 is constructed so as to be able to receive a display instruction input from the outside, and for example, the operator can select a specific delayed wave R displayed on the data display section 45.
When W'is designated, the delay time, intensity ratio, and phase difference between the delayed wave RW 'and the direct wave DW are calculated and displayed.

Further, it is possible to set and input the calculation display condition in advance in the data display section 45, and automatically display each information for the delayed wave which exceeds the set intensity ratio condition and delay time condition. You can also

As described above, the multipath delay spread measuring apparatus according to the present embodiment performs the spectrum despreading by the first mixer 25, converts the correlation waveform output by the logarithmic amplifier 31 into the amplitude information logE, and further receives it. Since the temperature inside the unit 20 is measured and the logE output is calibrated with the calibration data including the temperature characteristic, the device is capable of measuring the transmission state of the signal transmission system transmitted via the multipath using spread spectrum. Therefore, it is possible to output the accurate absolute value of each reception level of the direct wave and the delayed wave of the transmission signal at the position on the time axis.

Therefore, it is possible to set various conditions for communication such as the determination of the service area with high reliability, and it is possible to prevent the occurrence of bit errors in digital mobile communication in advance.

Further, as described above, the multipath delay spread measuring apparatus according to the present embodiment despreads by the first mixer 25, does not divide the correlation waveform output into the in-phase component, and The removing unit 30 causes D due to the temperature change by the first and second mixers 25 and 28.
Since the logarithmic amplifier 31 outputs the amplitude information logE after removing the influence of the C drift, the amplitude information l
It is possible to eliminate the influence of the temperature change of the ogE output, and it is possible to calculate the direct wave DW, the delayed wave RW, and the intensity ratio with higher accuracy.

Therefore, unlike the conventional apparatus, the minimum reception level does not deteriorate, so that the measurement dynamic range can be increased. Further, as described above, the multipath delay spread measuring apparatus according to the present embodiment mixes the PN signal PN2 in the first mixer, that is, the first mixer 25, and extracts the correlation waveform from the BPSK-modulated received signal RF. Since it is configured, when changing the delay resolution, there is no need to change the design of the entire circuit of the receiving unit 20, the first signal generator 23 is changed, and the code rate of the PN signal PN2 from the PN signal generator 24 is changed. The delay resolution can be easily changed only by increasing the value.

In the present embodiment, the use of the mobile communication system for designing the service area of the mobile communication has been described in mind, but the use of the present invention is not limited to this. It can also be used to develop a simulation model for service area design. Further, according to the present invention, since it is possible to provide a measuring device having a high delay resolution and capable of calculating a highly accurate intensity ratio, it is possible to grasp the propagation delay characteristic of an indoor wireless LAN system in a building, an underground mall, etc. Can be used.

Although the limiter output section 31a is part of the function of the logarithmic amplifier 31 (log amplifier) in the present embodiment, the configuration of the present invention is not limited to this.
The limiter output section 31a may be configured separately from the logarithmic amplifier 31, for example, as a discrete one. (Second Embodiment) FIG. 6 is a block diagram showing a multipath delay spread measuring apparatus according to the second embodiment of the present invention. The same parts as those of the conventional apparatus shown in FIG. Detailed description is omitted.

The receiving section of the multipath delay spread measuring apparatus shown in FIG. 6 is different from the multipath shown in FIG. 10 except for the temperature sensor 42, the calibration data storage section 43, the calibration processing section 44, and the data display section 45. It is configured and operates in the same manner as the receiving unit of the delay spread measuring device. That is, the output from the calculator 58 is the amplitude information logE converted for digital.

On the other hand, the temperature sensor 42, the calibration data storage unit 43, the calibration processing unit 44, and the data display unit 45 are
The amplitude information logE output from the computing unit 58 is calibrated and converted into an absolute reception level by being configured and operating in the same manner as in the first embodiment.

Incidentally, the correlation angle θ is calculated from the in-phase component correlation waveform I and the quadrature component correlation waveform Q by the calculation unit and input to the data display unit 45. The display of this portion is shown in FIG.
Omitted from.

The amplitude information calculation means according to claim 1 is composed of the logarithmic amplifier 31, and the temperature measuring means is composed of the temperature sensor 42. As described above, the multipath delay spread measuring apparatus according to the present embodiment performs spectrum despreading by the mixers 54 and 55, converts in-phase and quadrature-phase correlated waveform outputs into amplitude information logE by the calculator 58, and further Since the temperature inside the receiver 20 is measured and the logE output is calibrated with the calibration data including the temperature characteristic, a device for measuring the transmission state of the signal transmission system transmitted via the multipath using spread spectrum. Therefore, it is possible to output accurate absolute values of the reception levels of the direct wave and the delayed wave of the transmission signal at the position on the time axis.

Therefore, it is possible to set various conditions for communication such as the determination of the service area with high reliability, and it is possible to prevent bit error occurrence in digital mobile communication in advance.

The present embodiment can be easily realized by modifying the conventional multipath delay spread measuring device. It should be noted that the present invention is not limited to the above-described embodiments, but can be variously modified without departing from the scope of the invention.

[0119]

As described above in detail, according to the present invention, since the amplitude information logE is calibrated by the calibration data including the temperature characteristic, the transmission state of the signal transmission system transmitted via the multipath can be determined. It is possible to provide a multipath delay spread measuring device capable of outputting an accurate absolute value of each reception level of a direct wave and a delayed wave of a transmission signal at a time position in a device for measuring using spread spectrum.

[Brief description of drawings]

FIG. 1 is a configuration diagram showing a multipath delay spread measuring apparatus according to a first embodiment of the present invention.

FIG. 2 is a graph showing an absolute reception level of a reception signal with respect to an amplitude information voltage value in the multipath delay spread measuring apparatus of the embodiment.

FIG. 3 is a diagram showing a connection configuration of a transmission unit, a reception unit, and each measurement device at the time of measuring calibration data in the multipath delay spread measurement apparatus of the embodiment.

FIG. 4 is an explanatory view showing a process of extracting amplitude information from a received signal in the multipath delay spread measuring apparatus of the embodiment.

FIG. 5 is an explanatory diagram of interpolation using calibration data points in the multipath delay spread measuring apparatus of the embodiment.

FIG. 6 is a configuration diagram showing a multipath delay spread measuring apparatus according to a second embodiment of the present invention.

FIG. 7 is an explanatory diagram of spread spectrum and despread of a transmission signal.

FIG. 8 is an explanatory diagram of sliding correlation.

FIG. 9 is an explanatory diagram of BPSK modulation.

FIG. 10 is a configuration diagram showing a receiver of a conventional multipath delay spread measuring apparatus.

FIG. 11 is an explanatory diagram showing each correlation waveform and phase information calculation extracted from the conventional multipath delay spread measuring apparatus.

FIG. 12 is an explanatory diagram of TDMA communication and interference due to a delayed wave.

[Explanation of symbols]

10 ... Transmission unit, 20 ... Reception unit, 21 ... Reception antenna, 2
2 ... Oscillator, 23 ... 1st signal generator, 24 ... PN signal generator, 25 ... 1st mixer, 26 ... Band pass filter, 27 ... 2nd signal generator, 28 ... 2nd mixer, 2
9 ... Band pass filter, 30 ... DC component removing unit, 31 ...
Logarithmic amplifier, 32 ... Third signal generator, 33 ... Phase detector, 34 ... 90 ° phase shifter, 35 ... In-phase mixer, 36
... Quadrature phase mixer, 41 ... A / D converter, 42 ... Temperature sensor, 43 ... Calibration data storage section, 44 ... Calibration data processing section, 45 ... Data display section.

Claims (2)

[Claims] 1. A device for measuring a delayed wave by multipath of a radio wave propagation path in digital mobile communication, comprising a signal receiving section (21, 59) for receiving a spread spectrum RF signal obtained by modulating an RF signal with a PN code. PN for generating a receiving side PN signal in the same sequence as the PN code
A signal generator (24, 60), a mixer that mixes a signal including the reception side PN signal and a spread spectrum RF signal received by the signal reception unit, despreads by sliding correlation, and extracts a correlation waveform signal. (25, 54, 55), amplitude information calculation means (31, 58) for outputting an amplitude information signal by calculation on the correlation waveform signal, and reception of the measuring device including at least the mixer and the amplitude information calculation means. Temperature measuring means (42) for measuring the temperature in the main body of the unit, and calibration data for storing the absolute value of the reception level in the signal receiving unit corresponding to the temperature in the main unit of the receiving unit and the amplitude information signal as calibration data. The storage unit (43), the temperature inside the receiving unit main body, and the amplitude information signal are input and read from these input data and the calibration data storage unit. Calculating the absolute value of the received level at the signal receiver based on the true data, the multi-path delay spread measuring device, characterized in that a calibration processing unit (44) to be output. 2. A device for measuring a delayed wave by multipath of a radio wave propagation path in digital mobile communication, comprising a signal receiving section (21) for receiving a spread spectrum RF signal obtained by modulating a first RF signal with a PN code. , PN for generating a receiving side PN signal of the same sequence as the PN code
A signal generation section (24), a first mixer (25) for mixing the spread spectrum RF signal received by the signal reception section and the reception side PN signal, and despreading by the sliding correlation; A bandpass filter (26) for extracting a second RF signal from the output of the mixer, a first signal generator (27) for generating a local oscillation signal, and a second RF signal and the local oscillation signal. A second mixer (28) for mixing and outputting a first intermediate frequency signal
A DC component removing unit (30) for removing a DC component of the first intermediate frequency signal; a logarithmic amplifier (31) for outputting an amplitude information signal and a limiter signal from the output of the DC component removing unit; A second signal generator (32) for generating a second intermediate frequency signal having the same frequency as the first intermediate frequency signal; and a phase of the second RF signal from the limiter signal and the second intermediate frequency signal. Phase detector that outputs information (3
3), temperature measuring means (42) for measuring the temperature in the receiver main body of the measuring device including at least the logarithmic amplifier, and the signal reception corresponding to the temperature in the receiver main body and the amplitude information signal. A calibration data storage unit (43) that stores the absolute value of the reception level in the unit as calibration data, and the temperature and the amplitude information signal in the main body of the reception unit are input, and these input data and the calibration data storage unit are input. A multipath delay spread measuring device comprising: a calibration processing unit (44) for calculating and outputting an absolute value of a reception level in the signal receiving unit based on the read calibration data.