JPH0983296A - Filter circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide the filter circuit consisting of capacitors and MOS transistors(TRs) and having a Q multiple circuit suitable for MOS circuit integration. SOLUTION: A voltage is applied to a noninverting input terminal of an operational amplifier 16 from a 1st stage of MOS diodes 8, 7 connecting in series with a constant current source 9 and a voltage is applied to a gate of MOS resistor 15 from a 2nd stage of MOS diodes 11, 12 connected in series with a constant current source 10. The DC operating point is made stable by using the MOS resistance 15. An output of a filter 3 is fed back by a coefficient consisting of a ratio of a capacitance 17 to a capacitance 14 and added to the input 4 to form a Q multiple circuit.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a filter circuit for selecting or removing a desired signal from input signals having various frequency components. More specifically, it relates to a MOS integrated circuit of the filter circuit.

[0002]

2. Description of the Related Art Conventionally, as a method for easily constructing a filter having high selectivity Q, M. et al. E. FIG. VAN VANKE
NBURG, "Analog Filter Desi
gn ", Holt, Rinehart and Win
stone, NEW YORK, 1982, PP217
Q multiplier circuits are known. In general, a filter having a high Q requires a large element value ratio, and therefore, as shown in FIG. 2, feedback is performed by providing feedback with a two-stage configuration of an adder circuit and a low Q filter circuit. Multiplying the Q of the configured low Q filter circuit is
It is a Q multiplication circuit. In FIG. 2, the output (Vout) of a so-called second-order bandpass filter 3 represented by the transfer function of the following Expression 1 is input by multiplying the gain of Ka (Vin).
Is added and is input to the second-order bandpass filter 3. ω is the resonance angular frequency.

[0003]

[Equation 1]

The transfer function of the input Vin and the output Vout of the circuit of FIG.

[0005]

[Equation 2]

From the comparison between Expression 1 and Expression 2, the circuit of FIG. 2 has a Q that is 1 / (1-Ka) times that of the second-order bandpass filter 3 before multiplication.

[0007]

With the use of individual parts, FIG.
In the case of configuring the Ka portion of the circuit (1), it is often realized by a so-called resistance-coupling addition circuit using an operational amplifier and a resistor. However, if a circuit using a resistor and an operational amplifier is to be realized by a MOS integrated circuit that operates with a single power supply, the offset voltage of the operational amplifier is amplified because it has a DC gain, and the operating point shifts. there were. Also, if a capacitance coupling adder circuit is used instead of a resistor, the offset voltage will not be amplified, but in order to set the DC operating point, in parallel with the capacitance, M
There is a problem or problem that an unsuitable high resistance is required for the OS integrated circuit.

[0008]

SUMMARY OF THE INVENTION In order to solve the above-mentioned problems of the prior art, it is an object of the present invention to provide a filter circuit for multiplying Q by an adder circuit suitable for MOS integrated circuit. To do. In order to achieve such an object, in the present invention, the adding circuit is a capacitive coupling adding circuit, and a MOS resistor using a MOS transistor (hereinafter referred to as MOSTr) is used instead of the high resistance for setting the DC operating point. In order to obtain the gate voltage for setting the MOS resistance, first, in the operational amplifier, a constant current is passed through a diode-connected MOSTr to obtain a voltage near a threshold voltage (hereinafter Vth),
The voltage was set as the DC operating point. Here, the diode connection is to connect a gate electrode and a drain electrode in common to obtain a pseudo diode characteristic between the gate electrode and the source electrode.

Next, two diode-connected MOSTrs are connected.
By applying a constant current to those connected in series, almost 2
A voltage of twice the Vth was created and used as the gate voltage of the MOS resistor. Furthermore, to determine the DC operating point of the operational amplifier,
A constant current source is a diode-connected MOSTr, and a constant current source and a diode-connected MOSTr for obtaining a gate voltage of a MOS resistor. In the former, the voltage is taken out from the first stage to obtain a voltage near Vth, and the latter is taken from the second stage. Get the voltage.

According to the present invention, the circuit for multiplying Q of the filter circuit is configured as a capacitive coupling adder circuit, and the resistance for setting the DC operating point of the operational amplifier is realized by a MOS resistance using MOSTr. did. MOS resistance gate
Since the voltage between the source electrodes is the minimum required voltage, which is near Vth, the voltage used for the MOS resistor is
The size of STr can be reduced. Further, the circuit for generating the gate voltage of the MOS resistor and the circuit for generating the voltage at the DC operating point of the operational amplifier are separately provided with the same configuration to prevent interference in the circuit. The voltage generating circuit has a structure in which a current of a constant current source is passed through a diode-connected MOSTr in two stages connected in series. All of the circuits described above are MOS
Easy to be integrated circuit.

[0011]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Preferred embodiments of the present invention will be described in detail below with reference to the drawings. FIG. 1 is a circuit diagram showing an embodiment of a filter circuit according to the present invention. Power supply 1
9 is a P-channel MOS transistor (hereinafter referred to as PMOS
The source electrodes of Tr) 9 and 10 are connected. PMO
Since the constant voltage source 18 is connected between the gate electrode and the source electrode of each of the STr 9 and 10, the PMOSTr
9 and 10 show constant current characteristics regardless of the voltage between the drain electrode and the source electrode. The drain electrodes of the PMOSTrs 9 and 10 each have an N-channel MOS transistor (hereinafter NM) in which a gate electrode and a drain electrode are commonly connected.
OSTr) 8 and 11 are connected. Further NMOS
For the source electrodes of Tr8 and 11, NMOSTr7 and
Twelve drain electrodes and source electrodes are commonly connected. The source electrodes of the NMOSTrs 7 and 12 are connected to the ground potential.

Since the PMOSTrs 9 and 10 are set to have the same size, the NMOSTrs connected in series are connected.
The same current flows through 8 and 7, and 11 and 12. Therefore, N
Potential Va of gate electrode / drain electrode of MOSTr7
And the potential Vb of the gate electrode / drain electrode of the NMOSTr12 are equal, and the so-called threshold voltage of the NMOSTr is Vth.
Then, it becomes almost Vth. Similarly, NMOSTr8
Gate electrode / drain electrode potential Vc and NMOST
The potentials Vd of the gate electrode and the drain electrode of r11 are equal to each other and are approximately 2 × Vth. Since the gate electrode of the NMOSTr15 is connected to the potential Vd, it is similarly about 2 × V.
It becomes th.

The voltage generation circuit 32 is composed of the PMOSTr9, the NMOSTr8 and 7, and the PMOSTr10 and the NMOS
The Tr11 and Tr12 form a voltage generating circuit 31. The non-inverting input terminal of the operational amplifier 16 is connected to the potential Va, and the inverting input terminal of the operational amplifier 16 has capacitors 13, 1
It is commonly connected to one ends of 4, 17 and the source electrode of the NMOSTr 15. The other end of the capacitor 13 is connected to the input terminal 4, and the other end of the capacitor 14 is the output of the operational amplifier 16,
The drain electrode of the NMOSTr15 and the input 1 of the bandpass filter 3 having a quadratic transfer function are commonly connected. The other end of the capacitor 17 is connected to the bandpass filter 3
Output 2 and the output terminal 5 of the filter circuit 6. The bandpass filter 3 has the quadratic transfer function 3 shown in FIG. Although various methods of realizing the transfer function are known, an example is shown in FIG. The filter circuit 3 shown in FIG. 3 is a so-called multiple feedback type bandpass filter, and the transfer function from the input 1 to the output 2 can be expressed by Equation 1.

In FIG. 1, the capacitors 13, 14 and 17 and the operational amplifier 16 constitute a so-called capacitive coupling amplifier circuit. The gain of the capacitive coupling amplifier circuit can be set by the capacitance ratio. In order to make the circuit of FIG. 1 equal to the equivalent circuit of FIG. 2 in terms of alternating current, when the capacitance values of the capacitors 13, 14, 17 are C13, C14, C17, respectively, Ka = C17 / C14, C13 / C14 = 1 A DC operating point will be described. The voltage at the non-inverting input terminal of the operational amplifier 16 is Va, that is, almost Vth. The output of the operational amplifier 16 and the inverting input terminal are NMO
Since the DC-like total feedback is applied by the STr15, they are equal to each other, and the voltage is equal to the voltage of the non-inverting input terminal and is approximately Vth. That is, the potential of the source electrode of the NMOSTr15 becomes almost Vth. NMOS Tr1
In order to turn on 5, it is sufficient to apply a voltage of Vth or more between the source electrode and the gate electrode.

As described above, since the gate electrode of the NMOSTr15 is applied with a voltage of Vth which is almost doubled, a voltage of approximately Vth or more is applied between the source electrode and the gate electrode of the NMOSTr15 and is always turned on. . NM
Almost V between the source electrode and the gate electrode of the OSTr15
Since the voltage of th is a minimum voltage for turning on the NMOST 15r, the resistance value when turned on is large. Therefore, for example, as compared with the case where the power supply voltage is applied to the gate electrode, the transistor is relatively small. A large resistance value can be realized with the size.

Considering only the DC voltage, as described above,
Since Va = Vb and Vc = Vd, the voltage generating circuit portion for Va, Vc, etc. may be only one system. However, NMO
Since the STr15 causes so-called non-saturation operation,
The gate-source capacitance Cgd and the gate-drain capacitance Cgs are large. In particular, Cgd increases by an order of magnitude as compared with the so-called saturation operation. In order to realize a high resistance value for the transistor size, the gate length needs to be extremely longer than the gate width. Therefore, the gate capacitance determined by (gate length x gate width x gate capacitance per unit area) is also It is getting bigger.

For example, when the non-inverting input terminal of the operational amplifier 16 is connected to Va and the gate electrode of the NMOSTr15 is connected to Vc, the output of the operational amplifier 16 is C of the NMOSTr15.
Through gd, leak to the gate electrode of NMOSTr15, N
The voltage is divided by Cgs of the MOSTr8 and Cgs of the NMOSTr7 and appears in Vc and Va. Since Va is connected to the non-inverting input terminal of the operational amplifier 16 and forms a positive feedback loop with the output of the operational amplifier 16, unstable operation such as oscillation is likely to occur.

Therefore, in the present invention, V of the same voltage is applied.
A circuit for separately generating a, Vb and Vc, Vd is provided, one of which has a voltage V of the non-inverting input terminal of the operational amplifier 16.
The other one for generating a is used for generating the gate voltage Vd of the NMOSTr15 to avoid unstable operation.

In the circuit of FIG. 1 and the principle diagram of FIG. 2, not only the transfer function of the bandpass filter shown in FIG. 2 but also the transfer function of the band reject filter expressed by the following mathematical formula 3 It is also applicable to F (s).

[0020]

(Equation 3)

Equation 3 is realized by the circuit shown in FIG. 4 as is well known, but the negative sign of Equation 3 is realized by the aforementioned capacitive coupling amplifier circuit. In principle, by replacing Equation 3 with the transfer function 3 of FIG.
The new transfer function G (s) is

[0022]

(Equation 4)

In this case, Q is increased by (1 + Ka) times from the comparison between Expression 3 and Expression 4. Since the circuit of FIG. 4 passes a direct current with a gain of 1, the input 1 and the output 2 of the filter 3 are the direct current output of the capacitive coupling amplifier of the preceding stage, and the direct current voltage of approximately Vth appears as it is. The DC voltage of the input and output of the operational amplifier 20 used in FIG. 4 does not become 0V, and the operation area is secured.

In addition, the circuit of FIG.
The capacitor 17 is used to feed back only the alternating current. Since the DC component of the output 2 of the filter 3 is not fed back to the operational amplifier 16 in the previous stage, the DC operating point of the operational amplifier 16 is not affected by whether or not the filter 3 passes DC. Therefore, FIG.
It is possible to multiply the Q of a filter that allows direct current to pass.

FIG. 5 shows a filter circuit 6 according to the present invention,
An embodiment applied to the remote control receiving circuit 29 will be shown. Number 1
The infrared light 30 having a light emission cycle of 0 kHz is converted into an electric signal by a photoelectric conversion element 21 such as a photodiode,
It is inputted to the remote control receiving circuit 29 through the input terminal 22. In the remote control receiving circuit 29, the number of light emission cycles is 10 k
Detects whether the infrared light of Hz is incident or not. Generally, the signal at the input terminal 22 is at least 5
It is as weak as 0 μV or less. In the remote control receiving circuit 29, the input signal is amplified by the low noise amplifier 23, the amplitude is then limited by the limiter 24 to a constant value, and only the signal component is extracted by the bandpass filter 6 tuned to the light emission cycle. ，
The detection is performed by the detection circuit 25, and the detected direct current level is output to the output terminal 27 through the comparison circuit 26 which compares the direct current level with a fixed threshold value and outputs a high or low level. When there is a direct current input such as under sunlight, the direct current level setting circuit 28 operates to suppress the variation of the direct current level of the input terminal 22.

The remote control receiving circuit 29 needs a bandpass filter 6 having a high Q, which is tuned to the light emission cycle of the input so as to pass only the input signal. Since the bandpass filter 6 according to the present invention can easily form a filter having a high Q and can be easily realized by a CMOS process, the remote control receiving circuit 2 which is particularly a CMOS integrated circuit is provided.
Suitable for 9.

[0027]

As described above, according to the present invention,
The addition circuit necessary for the Q multiplication circuit that improves the Q of the filter is realized by a capacitance-coupling addition circuit that does not amplify DC, and the DC operating point is set by using MOS resistance instead of high resistance. Suitable for integrated circuits.

Further, the voltage generating circuit for applying the voltage to the gate electrode of the MOS transistor for the MOS resistor and the voltage generating circuit for setting the DC operating point of the adder circuit are separated to prevent interference, so that stable operation is achieved. It is possible to provide a Q multiplication circuit that can be performed.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is an AC equivalent circuit of the circuit of FIG.

FIG. 3 is a circuit diagram of an example of a bandpass filter having a quadratic transfer function.

FIG. 4 is a circuit diagram of an example of a band reject filter having a quadratic transfer function.

FIG. 5 is a circuit diagram showing an embodiment of the present invention.

[Explanation of symbols]

1 Filter input terminal 2 Filter output terminal 3 Filter 4 Q multiplier circuit filter input terminal 5 Q multiplier circuit filter output terminal 6 Q multiplier circuit filter 7, 8, 11, 12, 15 N-channel MOS transistor 9 , 10 P-channel MOS transistors 13, 14, 17 Capacitance 16, 20 Operational amplifier 18, 19 Voltage source 21 Photoelectric conversion element 22 Input terminal of remote control receiving circuit 23 Low noise amplifier 24 Limiter 25 Detection circuit 26 Comparison circuit 27 Remote control receiving Output terminal of the circuit 28 DC level setting circuit 29 Remote control receiving circuit 30 Infrared light 31, 32 Voltage generation circuit

Claims (1)

[Claims] 1. A first voltage generating circuit, a second voltage generating circuit, a source electrode connected to an inverting input terminal of an operational amplifier, a drain electrode connected to an output of the operational amplifier, and a gate electrode connected to the first electrode. A first MOS transistor connected to the output terminal of the voltage generating circuit, an input terminal at one end thereof connected to the inverting input terminal of the operational amplifier at the other end, and an inverting input terminal of the operational amplifier A second capacitor is provided between the output of the operational amplifier, a third capacitor is provided between the output terminal and the inverting input terminal of the operational amplifier, and a third capacitor is provided between the output of the operational amplifier and the output terminal. A filter circuit including a filter circuit having a quadratic transfer function and the operational amplifier whose non-inverting input terminal is connected to the output terminal of the second voltage generating circuit.