JPH09261008A - Filter coefficient arithmetic unit of fir type digital filter for digital beam forming device, fir type digital filter for digital beam forming device and digital beam forming device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To obtain a filter coefficient arithmetic unit of an FIR-type digital filter for a digital beam forming device which can be applied to information transmission using a carrier wave being a radio wave by providing a coefficient arithmetic means calculating the filter coefficient of plural FIR-type digital filters. SOLUTION: DBF circuit 6 of this digital beam forming device is provided with the FIR-type digital filters 9-1 to 9-N filtering and outputting an inputted digital common-mode and orthogonal signal XK (m) based on a load coefficient calculated by an FIR-type digital filter coefficient arithmetic unit 13 and provided with an adder 10. As the DBF circuit 6 and the arithmetic unit 13 are provided, a wide-band beam is formed in the incoming direction of a desired signal being a radio wave and the desired wide-band signal corresponding to the wide-band beam is received to output a desired output signal Z (m) being a base band signal.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method of forming a beam in the direction of a desired signal among incoming propagating wave signals and receiving the signal.
The present invention relates to a digital beam forming apparatus that suppresses nulls in the arrival direction of an interference signal, a filter coefficient calculation device of an FIR digital filter for the digital beam forming apparatus, and an FIR digital filter.

[0002]

2. Description of the Related Art Beam forming technology is an important signal processing technology conventionally used in radar and sonar. However, radar and sonar generally use narrowband signals with a narrow specific bandwidth. As a conventional method for forming a wideband beam for a baseband propagating wave having an infinite specific bandwidth, such as an acoustic signal, a conventional method is disclosed in "Nishikawa et al.," Wideband beam using fan filter. Formation ”, IEICE Technical Report CAS 89-
85, pp. 95-100, 1989 ". This conventional broadband beam forming method forms a broadband beam in a desired direction by using a fan filter (also called a velocity filter) used in seismic wave data processing in exploration of underground resources.

In the information transmission using a carrier wave as a propagating wave signal, there are the following devices as a device for forming a beam of a wide band signal. Here, in this specification, a signal in which the bandwidth of a high frequency (RF) signal is not negligible as compared with the carrier frequency will be referred to as a wideband signal. However, the bandwidth is narrower than 1/2 of the carrier frequency. FIG. 9 shows a related art document 2 “OL Fros.
t, III, “An algorithm of linearly constrained ad
aptive array processing ", Processedings of the IE
EE, Vol. 60, No. 8, pp. 926-935, 19
It is a block diagram of the adaptive digital beam forming apparatus of the 1st prior art example with respect to the above-mentioned wideband signal which applied the adaptive beam former disclosed by "August 1972". First of FIG. 9
In the conventional adaptive digital beam forming apparatus, the array antenna 100 including N antenna elements 1-1 to 1-N having approximately the same directional characteristics and frequency characteristics, and the receivers 2-1 to 2- N and A / D converters 3-1 to 3-
N, mixers 4-1 to 4-N, low-pass digital filters 5-1 to 5-N, digital local oscillator 60, and finite impulse response (hereinafter referred to as FIR).
Type digital filters 9-1 to 9-N, a second coefficient controller 11b, and an adder 10.

A first conventional adaptive digital beam forming apparatus will be described below with reference to FIG. The high frequency signals received by the plurality of N antenna elements 1-k are converted into intermediate frequency signals by the receiver 2-k, and the A / D converter 3-
Each of them is converted into a digital intermediate frequency signal by k (hereinafter, the intermediate frequency is referred to as f _IF ). Where k = 1,
2, ..., N, and the same shall apply hereinafter unless otherwise specified. These digital intermediate frequency signals are the frequency f oscillated by the digital local oscillator 60.
_{The IF} digital sine wave is mixed by the mixer 4-k, and the high-frequency component is cut by the low-pass digital filter 5-k.
Digital in-phase / quadrature signal x that is a complex number from k
_k (m) is output. Here, in the digital in-phase / quadrature signal x _k (m), the real part is the in-phase component and the imaginary part is the quadrature component. In addition, 2 in the line of the drawing indicates a symbol of a complex number.

N digital in-phase / quadrature signals x
₁ (m), x ₂ (m), ..., x _N (m) are FIR
Type digital filters 9-1 to 9-N.
The FIR digital filter 9-k has weighting factors wE _{k, 0} (m) and wE input from the second coefficient controller 11 b.
_{Based on k, 1} (m), ..., wE _{k, q-1} (m), the input digital in-phase / quadrature signal x _k (m) is digitally filtered, and the filtered digital in-phase signal is obtained.・ Quadrature signal x
Output _k (m) to the adder 10. Here, the second coefficient controller 11b uses the digital in-phase / quadrature signal x
_{Based on 1} (m), x ₂ (m), ..., X _N (m), at the output z _es (m) of the adaptive digital beam former,
The weighting factors wE _{k, 0} (m), wE _{k, 1} (m), ..., To extract a wideband desired signal and suppress a wideband interference signal.
By calculating wE _{k, q-1} (m), the weighting factor wE _{k, 0} (m),
Output wE _{k, 1} (m), ..., WE _{k, q-1} (m) to the FIR digital filter 9-k.

These load factors wE _{k, 0} (m), wE _{k, 1}
(M), ..., wE _{k, q−1} (m) are generally complex numbers.
Here, q is a tap length of the FIR digital filter 9-k and is a predetermined integer. Weighting factor wE
The calculation method of _{k, 0} (m), wE _{k, 1} (m), ..., WE _{k, q-1} (m) is, for example, LMS when a reference signal is used.
Known methods such as the (Least Mean Squares) method and the LMS method with direction constraint can be used if the incident direction of the desired signal is known. Here, a configuration in which signals from the antenna elements 1-k having the same directional characteristics are controlled as in the first conventional adaptive digital beam forming apparatus of FIG.

In the adaptive digital beam forming apparatus of the first conventional example, when a wideband signal in which the bandwidth of the target high frequency signal is not negligible compared with the carrier frequency is handled, in order to form a wideband beam or null, Since it is necessary to realize a “weighting coefficient” that depends on frequency instead of a simple weighting coefficient that does not depend on frequency, an FIR digital filter is used.

Further, FIG. 10 can deal with a wideband signal,
As an adaptive digital beam forming apparatus that requires only a small number of weighting factors to be calculated, the prior art document 3 “JF Yan” has been proposed.
g et al., “Wideband adaptive arrays based on the cohe
rent signal-subspace transformation ", IEEE
Alternativeal Confererence on Acoustics, Speech
h, and Signal Processing, pp. 2011-201
FIG. 4 is a block diagram showing a configuration of a second conventional adaptive digital beam forming apparatus disclosed in "4, 1987".
The adaptive digital beam forming apparatus of the second conventional example is
The signal converter 20 performs a non-linear pre-processing called coherent subspace conversion on the received signal to deal with the wideband signal. After this conversion, the adaptive processing is performed by using one weighting factor for each pass without using the FIR digital filter.

[0009]

However, the conventional wideband beam forming method disclosed in the prior art document 1 has a problem that it cannot be applied to information transmission using a carrier wave which is a radio wave. In the case of the first conventional adaptive digital beam forming apparatus having the element space configuration, the weighting factor wE to be calculated in the FIR digital filters 9-1 to 9-N.
_1,0 (m), wE _1,1 (m), ..., wE _{1, q-1} (m), w
The number of E _2,0 (m), ..., WEL _{, q-1} (m) is q × N, and when both the tap length q of the FIR digital filter 9-k and the number N of antenna elements are large. However, there is a problem in that the number of weighting factors to be controlled becomes extremely large, and thus the amount of signal processing calculation for calculating the weighting factor also increases dramatically. Further, in the adaptive digital beam forming apparatus of the second conventional example, the signal converter 20
There is a problem that the coherent subspace conversion in requires a priori knowledge about the incident direction of the signal.

A first object of the present invention is to solve the above problems and to apply an FIR for a digital beam forming apparatus which can be applied to information transmission using a carrier wave which is a radio wave.
It is to provide a filter coefficient calculation device for a digital filter.

A second object of the present invention is to solve the above problems and to apply the FIR for a digital beam forming apparatus which can be applied to information transmission using a carrier wave which is a radio wave.
Type digital filter.

A third object of the present invention is to solve the above problems, to deal with a desired signal in a wide band, to reduce the calculation time of the weighting factor, and to require prior knowledge about the incident direction of the desired signal. It is to provide a digital beam forming device that does not.

[0013]

According to a first aspect of the present invention, there is provided a filter coefficient calculation device for a FIR digital filter for a digital beam forming device according to the present invention, wherein a plurality of antenna elements are arranged on a straight line at predetermined element intervals. A plurality of Fs for a digital beam forming apparatus, which are provided corresponding to the plurality of antenna elements, are formed by using linear array antennas arranged in parallel and form a beam in a direction of a desired incident angle.
A filter coefficient calculation device for an IR digital filter, comprising: a first axis of a normalized time frequency F ₁ normalized by the sampling frequency of the digital beam forming device; and a normalization normalized by the reciprocal of the element spacing. In the two-dimensional frequency plane formed by the second axis of the normalized spatial frequency F ₂ and having a center on the second axis and having a predetermined width, from the negative normalized time frequency to the positive normalized It is characterized by further comprising coefficient calculating means for calculating filter coefficients of the plurality of FIR type digital filters having an amplitude in a desired pass region extending in time frequency.

According to a second aspect of the present invention, there is provided a filter coefficient calculation device for a FIR digital filter for a digital beam forming apparatus, wherein the coefficient calculation means is the filter coefficient calculation device for a FIR digital filter according to the first aspect. By converting the frequency response of a predetermined one-dimensional FIR digital filter into a predetermined variable, the frequency response corresponding to the desired pass region and related to the normalized time frequency F ₁ and the normalized spatial frequency F ₂ is calculated. Then
An impulse response is calculated by performing an inverse Fourier transform on the calculated frequency response, and based on the calculated impulse response, the plurality of Fs having an amplitude in the desired pass region are calculated.
A feature is that the filter coefficient of the IR type digital filter is calculated.

Further, the filter coefficient calculation device of the FIR type digital filter for the digital beam forming apparatus according to claim 3 is the filter coefficient calculation device of the FIR type digital filter according to claim 1 or 2, wherein the coefficient calculation means is used. Has a predetermined amplitude in the desired pass region, has a center on the second axis, and has a predetermined width in the two-dimensional frequency plane, and has a positive normality from a negative normalized time frequency. Of the plurality of F's having a predetermined amount of attenuation in the region where the sidelobe level extending to the signalization time frequency should be lowered.
A feature is that the filter coefficient of the IR type digital filter is calculated.

According to a fourth aspect of the present invention, there is provided a filter coefficient calculation device for a FIR digital filter for a digital beam forming apparatus, wherein the coefficient calculation means is the filter coefficient calculation device for a FIR digital filter according to the second aspect. The pass region of the one-dimensional FIR digital filter is rotated on the two-dimensional frequency plane, and the rotated pass region is translated on the two-dimensional frequency plane to perform variable conversion, and the desired pass region is obtained. Corresponding to the normalized time frequency F ₁ and the normalized spatial frequency F ₂ are calculated, and the impulse response is calculated by inverse Fourier transforming the calculated frequency response, and the calculated impulse response is calculated. Of the plurality of FIR digital filters having an amplitude in the desired pass region based on It is characterized in that the data coefficient is calculated.

According to a fifth aspect of the present invention, there is provided a filter coefficient calculation device for a FIR digital filter for a digital beam forming apparatus, wherein the filter coefficient calculation device for the FIR digital filter is the coefficient calculation means. Responds to the desired pass region by converting the frequency response of the one-dimensional FIR digital filter into a variable so that the beam pattern formed by the digital beam forming device does not change with respect to frequency. The impulse response is calculated by calculating the frequency response related to the normalized time frequency F ₁ and the normalized spatial frequency F _2, and performing the inverse Fourier transform on the calculated frequency response, and based on the calculated impulse response, the desired impulse response is calculated. The plurality of FIs having a predetermined amplitude in the pass region
A feature is that the filter coefficient of the R-type digital filter is calculated.

Further, a FIR type digital filter for a digital beam forming apparatus according to a sixth aspect of the present invention is the FIR type digital filter according to one of the first to fifth aspects. It is characterized in that the calculated filter coefficients of the plurality of FIR type digital filters are set.

According to a seventh aspect of the present invention, there is provided a digital beam forming apparatus, wherein an array antenna comprising a plurality of antenna elements and respective reception signals received by the respective antenna elements are A / D converted, Converting means for outputting each digital signal corresponding to each received signal,
A plurality of FIR type digital filters for filtering and outputting the plurality of digital signals output from the converting means based on a plurality of filter coefficients, and a plurality of filters output from the plurality of FIR type digital filters. A DBF circuit including an adder for adding the post-wave digital signals and outputting the added signal as an output signal,
A plurality of filter coefficients of the plurality of FIR digital filters are calculated by the filter coefficient calculation device according to any one of claims 1 to 5.

Further, in the digital beam forming apparatus according to the present invention of claim 8, an array antenna comprising a plurality of N antenna elements and each received signal received by each of the antenna elements are A / D converted. Then, the conversion means for outputting each digital signal corresponding to each received signal and the plurality of B beams to be formed corresponding to the plurality of N FIR type digital filters for each antenna element, On the other hand, a plurality of N FIR type digital filters are provided correspondingly, and a plurality of (N × B) FIR type digital filters which respectively filter and output the respective digital signals according to a predetermined filter coefficient, , The plurality of N signals output from the plurality of N FIR type digital filters for forming the respective beams, A plurality of B adders for calculating and outputting the plurality of B beams are formed in a plurality of B different directions based on the respective digital signals output from the converting means, and a plurality of B beams corresponding to the beams are formed. Beam forming means for outputting a plurality of beam reception signals, and signal selecting means for selecting and outputting a plurality of M beam reception signals from a plurality of B beam reception signals output from the beam forming means,
A plurality of M multipliers for respectively multiplying the plurality of M beam reception signals input from the signal selection means and a plurality of M weight coefficients input and outputting a signal of a multiplication result, and the signal selection. Based on the plurality of M beam reception signals output from the means, the main beam of the array antenna is directed to the arrival direction of the desired signal and the arrival direction of the interference signal in the predetermined frequency range including at least the frequency of the desired signal. Coefficient control means for calculating the plurality of M weighting factors for zeroing the level of the received signal for each of the multipliers and outputting the plurality of M weighting factors to the corresponding multipliers. And adding means for adding a plurality of M multiplication result signals output from the plurality of M multipliers and outputting as a reception signal.

According to a ninth aspect of the present invention, there is provided the digital beam forming apparatus according to the eighth aspect, wherein the plurality of (N × B) FIR digital filters have a plurality of filter coefficients. It is characterized in that it is calculated by the filter coefficient calculation device described in any one of 1 to 5.

According to a tenth aspect of the present invention, there is provided a digital beam forming apparatus in which an array antenna including a plurality of N antenna elements and respective reception signals received by the respective antenna elements are A / D converted. A plurality of B first beams which are to be formed by a plurality of N weight coefficient units corresponding to the respective antenna elements, and conversion means for outputting each digital signal corresponding to each received signal. A plurality of N weight coefficient units are provided so as to correspond to the first beam, and a plurality of (N) each of the digital signals is output by multiplying by a predetermined first weight coefficient.
XB) weighting factor units and a plurality of B first numbering factors that add and output a plurality of N number of signals output from the plurality of N numbering factor devices for forming each of the first beams. Of the plurality of B beams corresponding to the first beam are formed in a plurality of B different directions based on the respective digital signals output from the conversion means. Beam forming means for outputting a first beam reception signal and a plurality of L FIR type digital filters correspond to each of the first beam reception signals, and a plurality of B second beams to be formed are formed. A plurality of L FIR digital filters are provided so as to correspond to the respective second beams, and the respective first beam reception signals are formed in advance so as to form the plurality of B second beams. Multiple fixed filters A plurality of outputting and filtered by the number (L ×
B) A plurality of L type signals output from the FIR type digital filters and a plurality of L type FIR digital filters for forming each of the second beams are added to obtain each of the second beams. A plurality of B second adders for outputting corresponding second beam reception signals;
Signal selecting means for selecting and outputting a plurality of M second beam reception signals from a plurality of B second beam reception signals output from a plurality of adders, and the plurality of signals input from the signal selecting means. A plurality of M number of multipliers that respectively output the M second beam reception signals and a plurality of M number of second weighting factors that are input and output signals of the multiplication results, and the above-mentioned signal selecting means. Based on the plurality of M second beam reception signals, the main beam of the array antenna is directed to the arrival direction of the desired signal and the reception direction of the interference signal is received in a predetermined frequency range including at least the frequency of the desired signal. The plurality of M second weighting factors that make the signal level zero are calculated for each of the multipliers, and the plurality of M second weighting factors are respectively stored in the corresponding multipliers. Output coefficient control means , Characterized in that an adding means for outputting a plurality of M reception signals by adding the plurality of M multiplication results of the signal output from the multiplier.

The digital beam forming apparatus according to claim 11 is the digital beam forming apparatus according to claim 10, wherein the plurality of (L × B) FIR digital filters have a plurality of filter coefficients. It is characterized in that it is calculated by the filter coefficient calculation device described in any one of 1 to 5.

Further, in a digital beam forming apparatus according to a twelfth aspect, in the digital beam forming apparatus according to the eighth aspect, the coefficient controlling means,
Each of the weighting factors is calculated so that the envelope of the signal output from the digital beam forming device is kept constant.

[0025]

BEST MODE FOR CARRYING OUT THE INVENTION

<First Embodiment> FIG. 1 is a block diagram showing the arrangement of a digital beam forming apparatus according to the first embodiment of the present invention. The digital beam forming apparatus of the first embodiment is similar to the adaptive digital beam forming apparatus of the first conventional example shown in FIG. 9, except that the second coefficient controller 11b is replaced by FIR.
A digital filter coefficient calculator 13 is provided. Here, in the digital beam forming apparatus according to the first embodiment, the FIR type digital filter coefficient calculator 13 has a frequency response when an incident angle θ which is a direction for forming a beam is given, as will be described later in detail. G (F _1, F ₂₎ by calculating the arithmetic frequency response G impulses by inverse Fourier transform (F _1, F ₂₎ response g (m _1, m ₂₎
And further calculates and outputs the weighting factors w _{k , s} based on the impulse response g (m ₁ , m ₂ ), and the FIR digital filters 9-1 to 9-N respectively output the weighting factors. Based on w _{k, s} , the input digital in-phase
The quadrature signal x _k (m) is filtered and output to the adder 10. As a result, the adder 10 has a plurality of N
The filtered digital in-phase / quadrature signals x _k (m) are added, and the added signal is used as an output signal z (m) in which a wideband desired signal is extracted and a wideband interference signal is suppressed. Output. Here, k = 1, 2, ..., N, and the same applies hereinafter unless otherwise specified in this specification. Further, m in parentheses () in the digital in-phase / quadrature signal x _k (m) represents time (time number).

The digital beam forming apparatus according to the first embodiment will be described in detail below with reference to FIG. In the digital beam forming apparatus of the first embodiment, the high-frequency signals received by the plurality of N antenna elements 1-k are respectively received by the receiver 2-k similarly to the adaptive digital beam forming apparatus of the first conventional example. Converted to an intermediate frequency signal with
Each of the signals is converted into a digital intermediate frequency signal by the A / D converter 3-k. Then, these digital intermediate frequency signals are the frequency f _IF oscillated by the digital local oscillator 60.
Is mixed with the digital sine wave of the low-pass digital filter 5-k and the high-frequency component is cut by the low-pass digital filter 5-k.
To complex digital in-phase / quadrature signal x _k (m)
Is output. Then, the digital in-phase / quadrature signal x _k
(M) is the FIR type digital filters 9-1 to 9-
It is input to the FIR type digital filter 9-k of the DBF circuit 6 composed of N and the adder 10. Here, in the first embodiment, the array antenna 100 is a linear array antenna in which antenna elements 1-1 to 1-N are juxtaposed on a straight line at a predetermined element interval d.

FIR type digital filter coefficient calculator 1
As will be described later in detail, when an incident angle θ, which is a direction for forming a beam, is given, the reference numeral 3 extracts a desired signal in a wide band from the output signal z (m) of the digital beam forming device and causes wide band interference. The weighting factors w _{k, 0} , w _{k, 1} , ..., W _{k, q-1} are calculated so as to suppress the signals, and the weighting factors w _{k , 0} , w _{k, 1} , ..., W _{k, q- 1} is output to the FIR digital filter 9-k.

In the DBF circuit 6, the FIR digital filter 9-k has the weighting factors w _{k, 0} and w input from the FIR digital filter coefficient calculator 13.
_{k, 1,} ..., w _k, based on the _q-1, a digital in-phase and quadrature signals x _k (m) are digitally filtered version of the input, the digital in-phase and quadrature signals x _k after filtering ( m) is the adder 1
Output to 0. The adder 10 adds a plurality of N filtered digital in-phase / quadrature signals x _k (m) input from each FIR digital filter 9-k, and outputs the added signal as a wide band desired signal. The output signal z (m) in which the interference signal in the extracted and wide band is suppressed is output.

Next, the configuration of the FIR type digital filter 9-k will be described with reference to FIG. As shown in FIG. 2, the FIR digital filter 9-k includes (q-1) delay units 91-1 to 91- (q-1) and q multipliers 92-1 to 92-q And (q-1) adders 93
-1 to 93- (q-1). Where FIR
In the type digital filter 9-k, q is called a tap length, and is set to an odd number in the first embodiment. And
The digital in-phase / quadrature signal x _k (m) input to the FIR digital filter 9-k is input to the delay unit 91-1 and the multiplier 92-1. The weighting coefficient w _{k, s-1} input to the FIR digital filter 9-k is input to the multiplier 92-s (s = 1, 2,..., Q).

In the FIR type digital filter 9-k, the delay device 91-s (s = 1, 2, ..., Q-1) inputs the digital in-phase / quadrature signal x _k (m-s + 1).
Is delayed by one sample period, and a signal x _k (ms) delayed by one sample period is output to the delay unit 91- (s + 1) and the multiplier 92- (s + 1). The multiplier 92-1 is
The input digital in-phase / quadrature signal x _k (m) is multiplied by the weight coefficient w _{k, 0} and output to the adder 93-1.
The multiplier 92-s (s = 2, 3,..., Q) multiplies the input signal x _k (m−s + 1) by the weight coefficient w _{k, s−1,} and adds the signal x _k (m−s + 1) to the adder 93- (s). Output to -1). Adder 93-
1 is a signal input from the multiplier 92-1 and the multiplier 92-1
-2 is added to the signal inputted from the adder 93-2.
Output to The adder 93-s (s = 2, 3,..., Q-
2) adds the signal input from the adder 93- (s-1) and the signal input from the multiplier 92- (s + 1), and outputs the result to the adder 93- (s + 1). Adder 93-
(Q-1) adds the signal input from the adder 93- (q-2) and the signal input from the multiplier 92-q and outputs the result to the adder 10. Here, in the first embodiment, (q-1) adders 93-1 to 93- (q-
1) is provided, but the present invention is not limited to this, and (q)
-1) In place of the adders 93-1 to 93- (q-1), q output signals output from the multipliers 92-1 to 92-q are collectively added to adder 10 It may be configured by using one adder for outputting to. Further, q output signals output from the multipliers 92-1 to 92-q may be directly output to the adder 10 without using such an adder.

The FIR type digital filter 9-k having the above-mentioned structure is provided with the respective weighting factors w _{k, 0} , w _{k, 1} inputted from the FIR type digital filter coefficient calculator 13.
, W _{k, q−1} based on w _{k, q −1, the} digital in-phase / quadrature signal x input from the low-pass digital filter 5-k.
_Digitally filter _k (m) and output to adder 10. The DBF circuit 6 including the FIR digital filters 9-1 to 9-N configured as described above can be considered as a two-dimensional digital filter. In addition, F of FIG.
The structure of the IR type digital filters 9-1 to 9-N is a direct type, but other configurations such as a cascade type and a lattice type may be used. However, FIR type digital filter 9-k
The calculation method of the weighting factor w _{k, s} input to is different depending on the structure of the FIR digital filter 9-k.

Next, a method of calculating the weighting coefficient w _{k, s in} the FIR type digital filter coefficient calculator 13 will be described. First, as shown in FIG. 1, the incident angle θ of the desired signal is defined by the angle between the perpendicular of the line in which the antenna elements 1-1 to 1-N are juxtaposed and the incident direction of the desired signal.
Further, the normalized time frequency F _{1 is} calculated by using the frequency f ₁ of the desired signal which is a high frequency signal, the carrier frequency f _{c of the} carrier signal and the sampling frequency f _s of the digital beam forming apparatus.
Is defined by the following equation 1. Further, using the incident angle θ, the speed of light c, and the frequency f _{1 of the} desired signal, the denormalized spatial frequency f ₂
Is defined as Equation _2, and the denormalized spatial frequency f ₂ is normalized by the reciprocal of the element spacing d of the antenna elements 1-1 to 1-N as represented by Equation 3, and the normalized spatial frequency F ₂ Is defined. Here, the reciprocal of the element spacing d of the antenna elements 1 /
d is called the spatial sampling frequency. That is,
The normalized time frequency F ₁ is a frequency obtained by normalizing the difference between the frequency f ₁ and the carrier frequency f _{c at} a certain time with the sampling frequency f _s , and the normalized spatial frequency F ₂ is the incident angle θ.
Is a spatial frequency normalized on the juxtaposed line of the antenna element 1-k at a certain time.

[0033]

[Number 1] _{F 1 = (f 1 -f c} ) / f s

F ₂ = (f ₁ sin θ) / c

## EQU3 ## F ₂ = df ₂ = (df ₁ sin θ) / c = ｛(dsin
_{θ) / c} (f s} F 1 + f c)

The transfer characteristic H (f ₁ , θ) when the DBF circuit 6 is regarded as a two-dimensional digital filter can be expressed by the following equation 4, and the transfer characteristic H (f ₁ , θ) is , The frequency response G (F ₁ , F ₁ shown in Equation 5 using the normalized time frequency F ₁ and the normalized spatial frequency F ₂ defined as above.
₂ ) can be converted to. That is, the frequency response G
(F ₁ , F ₂ ) is a frequency response on the two-dimensional frequency plane of the normalized time frequency F ₁ and the normalized spatial frequency F ₂ .

[0035]

(Equation 4)

(Equation 5)

Here, in the equation 4, T _s is a sampling interval in the DBF circuit 6, and is represented by the reciprocal of the sampling frequency f _s in the DBF circuit 6, that is, T _s = 1 / f _s . Further, in the expressions 4 and 5, q is the tap length of the FIR type digital filter 9-k as described above. Equations (4) and (5) above are equations that hold when the element spacing d is assumed to be constant. This method has a limitation that it can be applied only to a linear array in which the element spacing d is constant. However, when the antenna elements 1-1 to 1-N are arranged on the plane in a grid pattern at equal intervals in the vertical and horizontal directions, it can be dealt with by expanding the equations 4 and 5. That is, the present invention can be applied to a case where the digital filter is three-dimensional (two-dimensional space-time).

The DBF circuit 6 has desired characteristics on a frequency-angle plane for forming a wideband beam in a desired direction.
By converting the normalized time frequency F ₁ and the normalized spatial frequency F ₂ plane into desired characteristics on the two-dimensional frequency plane, and approximating the characteristics of the two-dimensional digital filter to the desired characteristics on the two-dimensional frequency plane. It was realized. In the processing described above, the inventors of the present invention use the broadband DBF circuit 6 that processes a radio wave for transmitting information using a carrier wave as shown in FIG.
The desired characteristic in the frequency-angle plane shown in (a) is 2
It was found that the desired pass band shown in FIG. 3B is expressed on the dimensional frequency plane.

Here, in order to form a wide band multi-beam and then select the beam to form an adaptive wide band beam or a wide band null, the phase characteristic of the circuit forming each beam is linear with respect to frequency. , And the slopes must all be the same, but the method described below can satisfy the demand.

[0039] Therefore, first, the digital in-phase and quadrature signals x _k (m) result of studying in detail inputted to DBF circuit 6, a digital-phase and quadrature signal corresponds to a signal at an incident angle θ x _k (m) It was found that the two-dimensional spectrum of time and space at the input of the DBF circuit 6 is on the straight line represented by the following equation 6 on the two-dimensional frequency plane. FIG. 5 is a diagram schematically showing the two-dimensional spectrum Sp.

[0040]

[Equation 6] F ₂ = {(d sin θ) / c} (f _s F ₁ + f _c ).

That is, the straight line expressed by the equation 6 is shown in FIG.
It is a straight line located at the center in the width direction of the desired pass band of the rectangle shown in FIG. Therefore, the desired rectangular passband shown in FIG. 3B can be expressed by the following inequality shown in Equation 7.
The first axis of the normalized time frequency F ₁ normalized by the sampling frequency of the digital beam former and the second axis of the normalized spatial frequency F ₂ normalized by the reciprocal of the element spacing.
In the two-dimensional frequency plane formed by the axis and the center of the second axis, and has a predetermined width and extends from the negative normalized time frequency to the positive normalized time frequency.

[0042]

Equation 7] {(dsinθ) / c} ( f s F 1 + f c) -ε ≦ F 2 ≦ {(ds
_{inθ) / c} (f s} F 1 + f c) + ε

Here, ε in the equation (7) is a predetermined small number corresponding to the width of the rectangular desired pass band shown in FIG. 3 (b). That is, the amplitude characteristic D in the desired pass band of the rectangular 3 shown by the number _{7 (b) (F 1,} F 2) is = 1, the desired passband in the external amplitude characteristic D (F _1, F ₂ If the DBF circuit 6 that is a two-dimensional digital filter that can approximately realize the amplitude characteristic of) = 0 can be configured, the DBF circuit 6 that forms a broadband beam in the direction of the incident angle θ can be configured. Here, in the first embodiment, F
Even in the vicinity of ₁ = ± 0.5, the amplitude characteristic D (F ₁ , F ₂ ) =
It was set to 0.

Next, an amplitude characteristic having an amplitude in the above-mentioned rectangular desired pass band, that is, an amplitude characteristic D (F ₁ , F ₂ ) = in the rectangular desired pass band shown in FIG. 1 and the amplitude characteristic D (F ₁ , F ₂ ) = outside the desired passband.
A two-dimensional digital filter that can approximately realize an amplitude characteristic of 0 was studied. The result will be described with reference to FIG.

First, based on the number of antenna elements N, the impulse response length Ni of a one-dimensional linear phase FIR type narrow band low pass digital filter (hereinafter referred to as FIR type low pass digital filter) is defined as follows. To do.
That is, when the number of antenna elements N is an odd number, Ni = N
If the number of antenna elements N is an even number, then Ni = N-1
And As a result, the impulse response length Ni is always set to an odd number. Next, the FI with an impulse response length of Ni
The impulse response p (m) of the R type narrow band low pass digital filter is obtained. Here, the band width of the FIR type narrow band low pass digital filter is preferably as narrow as possible. There are several known methods for calculating the impulse response p (m), but the simplest method is to set p (m) = 1 / Ni.

Here, for simplicity, only the zero-phase portion, which does not affect the amplitude characteristic and excludes the linear phase component, is considered. Therefore, the frequency response P (F ₂ ) of the FIR type low pass digital filter obtained here can be expressed by the following equation 8. Here, the frequency response P (F ₂ ) of Equation 8 is
Since the phase is zero, the relationship of p (m) = p (-m) is established.

[0047]

(Equation 8)

Here, in equation (8), T _m (x) is a Chebyshev polynomial of the first kind of the m-th order, and T ₀ (x) =
_{1, T 1 (x) =} x, T m (x) = 2xT m-1 (x) -T
_m-2 (x); m = 2, 3, ..., T _m (cosx) = cos (m
x). The frequency response P (F ₂ ) of Equation 8 can be expressed as shown in FIG. 4A when expressed on the two-dimensional frequency plane.

Next, the frequency response P (F ₂ ) of the equation 8 shown in FIG. 4A is rotated on the two-dimensional frequency plane as shown in FIG. 4B. This rotation is performed as follows. That is, when cos (2πF ₂ ) in the equation 8 is replaced with the function R (F ₁ , F ₂ ) represented by the following equation 9, the frequency characteristic can be rotated approximately.

[0050]

## EQU9 ## R (F ₁ , F ₂ ) = R _c (F ₁ ) cos (2πF ₂ ) −R _s (F ₁ ).
sin (2πF ₂ )

(Equation 10) Here, a ₀ = sin (uπ) / (uπ), a _m = 2 · (−1) ^m usin
(uπ) / {π (u ² −m ² )}

[Equation 11] Here, b _m = 2 · (−1) ^m msin (uπ) / {π (u ² −m ² )}

[Equation 12] u =-(df _s / c) sin θ

Here, the function R _c (F ₁ ) in the equation 9 is the equation 1
0, and the function R _s (F ₁ ) is expressed by Equation 11. Further, u in Expressions 10 and 11 is a constant represented by Expression 12. Further, the c _m in number 11 centered at m = 0, a suitable window function of length 2L + 1 (except for rectangular window). Here, the rotation angle is represented by tan ⁻¹ (−u). The above-described rotation method of the frequency characteristic is a known method, and is described in the related art document 4
"Hasegawa et al.," Passband variable 2D based on spectrum conversion
The FIR fan filter is described in detail in "Technical Research Report CAS 89-158 of the Institute of Electronics, Information and Communication Engineers".

Next, the rotated frequency characteristic is translated along the F ₂ axis. The translation amount F _{2shif t} is given by the following _equation 13.

[0053]

[ _Equation 13] F 2 _shift = (df _c / c) sin θ

Frequency response G translated as described above
(F ₁ , F ₂ ) can be expressed by the following equation 14.

[0055]

[Equation 14]

Here, when the number N of antenna elements is an even number, it is expressed by the following _expression 15 by further multiplying the right side of _expression 14 by cos {π (F ₂ −F _{2 shift} )}.

[0057]

(Equation 15)

Next, the impulse response is obtained based on the frequency response of the equation 14 or the equation 15, and the weight coefficient w _{k, s} is calculated. To do so, first of all, an appropriate score I expressed as a power of 2
Then, the time frequency F ₁ and the spatial frequency F ₂ are equally divided on the two-dimensional frequency plane between −0.5 and 0.5. Number 14 or number 15 on those points, respectively
An impulse response g (m ₁ , m ₂ ) is obtained by obtaining a frequency response G (F ₁ , F ₂ ) represented by and performing an inverse discrete (fast) Fourier transform of the frequency response G (F ₁ , F ₂ ). . Then, using the impulse response g (m ₁ , m ₂ ) obtained as described above,
The weight coefficient w _{k, s} is given by the following equation 16. Where m ₁
Is a variable related to time, and m ₂ is a variable related to space.

[0059]

W _{k, s} = g (s− (q−1) / 2, −k + (N−1) / 2 + 1) k = 1,2, ..., N; s = 0,1, ..., q -1

Here, the range of m ₁ and m ₂ is -I / 2 to I
/ 2 and the number of antenna elements N is an even number,
(N-1) / 2 in the above equation is N / 2. Regarding the time direction (m ₁ ), g (m ₁ , m ₂ ) will be cut off midway according to the value of q. Based on the above principle, the DBF circuit 6 is
The amplitude characteristic D (F ₁ , F ₂ ) = 1 is approximately in the rectangular desired pass band of FIG. 3B shown by the above-mentioned equation 7, and the amplitude characteristic D is approximately outside the desired pass band. (F ₁ , F ₂ ) = 0
By operating as a two-dimensional digital filter having an amplitude characteristic, it is possible to form a broadband beam in the direction of the incident angle θ of the desired signal.

In order to execute the DBF processing based on the above principle, in the first embodiment, the FIR type digital filter coefficient calculator 13 uses the FIR angle on the basis of the incident angle θ which is the direction in which the beam is to be formed. The frequency response G (F ₁ , F ₂ ) of the two-dimensional digital filter is calculated by rotating the pass characteristic of the low-pass type digital filter on the two-dimensional frequency plane and then translating the frequency response G (F ₁ , F ₂ ). The impulse response g (m ₁ , m ₂ ) is calculated by performing an inverse discrete Fourier transform on (F ₁ , F ₂ ), and the weighting factor w _k, is calculated based on the impulse response g (m ₁ , m ₂ ) _{. s}
Is calculated. Then, in the DBF circuit 6, each of the FIR digital filters 9-1 to 9-N digitally filters the input digital in-phase / quadrature signal x _k based on the weighting coefficient w _{k, s} , The digital in-phase / quadrature signal x _k after filtering is output, and the adder 10 has a plurality of N
The filtered digital in-phase / quadrature signals x _k are added and output. As a result, the DBF circuit 6 operates as a two-dimensional digital filter having the above-mentioned amplitude characteristic D (F ₁ , F ₂ ), so that when the incident angle θ of the desired signal is given, a wideband beam in that direction is obtained. Can be formed, and an output signal z (m) that is a desired signal of a wide band corresponding to the wide band beam is output.

In the digital beam forming apparatus according to the first embodiment described in detail above, the desired signal, which is a wideband propagation wave signal, is received by the N antenna elements 1-1 to 1-N together with the wideband interference signal. Antenna element 1 received
The received signals received at -1 to 1-N are respectively A /
The digital signals are converted into digital signals by the D converters 3-1 to 3-N, low-pass filtered by the low-pass digital filter 5-k, and then wide band by the FIR digital filters 9-1 to 9-N. , W _{k, q−1} are calculated based on the weighting factors w _{k, 0} , w _{k, 1} , ..., W _{k, q−1} calculated so as to suppress a wideband interference signal having an incident angle different from that of the desired signal. After being waved, they are combined to extract a desired signal, and a desired output signal z (m) in which an interference signal is suppressed is output as an output signal of the digital beam forming apparatus.

In the digital beam forming apparatus according to the first embodiment described above, the FIR digital filter coefficient calculator 13 does not use the optimization method or solve the simultaneous equations, and the frequency response G (F ₁ , F ₂ ) Is calculated and the frequency response G (F ₁ , F ₂ ) is inverse-Fourier-transformed to calculate the weighting factor w _{k, s} , so that the weighting factor w _{k, s} can be calculated at high speed. .

In the first embodiment described above, the DBF circuit 6 receives the digital in-phase / quadrature signal x _k (m) which is input based on the weighting coefficient w _{k, s} calculated by the FIR digital filter coefficient calculator 13. ) Is provided and FIR digital filters 9-1 to 9-N and the adder 10 are provided. Therefore, the amplitude characteristic D (F ₁ ,
It operates as a two-dimensional digital filter having F ₂ ). As a result, the DBF circuit 6 can form a wideband beam in the direction of the incident angle θ of the desired signal and output an output signal z (m) that is a wideband desired signal corresponding to the wideband beam. You can

The digital beam forming apparatus according to the first embodiment described above includes the DBF circuit 6 and the FIR type digital filter coefficient calculator 13 described above. Therefore, in information transmission using a carrier wave, which is a radio wave, It is possible to form a broadband beam in the direction of arrival of the desired signal that is
A wideband desired signal corresponding to the wideband beam is received, and a desired output signal z (m) that is a baseband signal is received.
Can be output.

In the digital beam forming apparatus according to the first embodiment described above, the DBF circuit 6 has a linear phase characteristic, and the slope of the phase characteristic is determined by the value of the tap length q.
Therefore, when the present method is used to form a wideband multibeam suitable for an adaptive digital beam forming apparatus described later, it is possible to form a wideband multibeam by aligning the tap length q in each single beam DBF circuit. it can.

<Second Embodiment> FIG. 6 is a block diagram of an adaptive digital beam forming apparatus according to the second embodiment of the present invention. The adaptive digital beam forming apparatus according to the second embodiment will be described below with reference to FIG. The adaptive digital beam forming apparatus according to the second embodiment shown in FIG. 6 is the same as the adaptive digital beam forming apparatus according to the second conventional example shown in FIG. 10, except that the signal converter 20 is replaced by a first multi-beam former 7a and a signal. And a selector 8.

The adaptive digital beam forming apparatus according to the second embodiment will be described in detail below with reference to FIG.
Here, in the second embodiment of FIG. 6, the same components as those of the first embodiment of FIG. 1 and the second conventional example of FIG. 10 are denoted by the same reference numerals. In the adaptive digital beam forming apparatus according to the second embodiment, the high frequency signals received by the N antenna elements 1-k are respectively received by the receiver 2-k similarly to the adaptive digital beam forming apparatus according to the first embodiment. Converted into an intermediate frequency signal by the A / D converter 3-k
Are converted into digital intermediate frequency signals respectively. Then, these digital intermediate frequency signals are respectively mixed with the digital sine wave of the frequency f _IF oscillated by the digital local oscillator 60 by the mixer 4-k, and the high frequency component is cut by the low pass digital filter 5-k,
Complex low-pass digital filter 5-k digital in-phase / quadrature signal x ₁ (m) x ₂ (m), ..., x
_N (m) is output and the signal is input to the first multi-beam former 7a.

The first multi-beam former 7a is a FIR
Is a kind of spatial filter bank provided with digital filters 7-1-1 to 7-BN and adders 10-1 to 10-B. Here, the digital in-phase / quadrature signal x
₁ (m) is the FIR type digital filter 7-1-1,
7-2-1, 7-3-1, ..., 7-B-1,
The digital in-phase / quadrature signal x ₂ (m) is transmitted to the FIR digital filters 7-1-2, 7-2-2, 7-3-2,
, 7-B-2, and the digital in-phase / quadrature signal x _k (m) (k = 3, 4, ..., N) is input to FI.
R type digital filter 7-1-k, 7-2-k, 7-
3-k, ..., 7-B-k.

More specifically, each of the FIR digital filters 7-1-k (k = 1, 2, ..., N) receives an input digital in-phase / quadrature signal x _k (m).
Is digitally filtered based on a predetermined coefficient and output to the adder 10-1. Then, the adder 10-1 adds each digital in-phase / quadrature signal x _k (m) after filtering,
The signal y ₁ (m) after addition is output to the signal selector 8. Here, the weighting coefficient which is each filter coefficient of the FIR type digital filter 7-1-k uses a method similar to that of the FIR type digital filter coefficient calculator 13 of the first embodiment, and a weighting factor of a wide band in a predetermined direction is obtained. A signal y ₁ corresponding to a broadband signal that forms a beam and is incident from the direction of the beam
Set to output to (m). However, in the present invention, each weighting factor of the FIR digital filter 7-1-k uses another method to form a broadband beam in a predetermined direction, and a broadband signal incident from the direction of the beam is formed. May be set to output to the signal y ₁ (m) corresponding to.

Further, the FIR type digital filter 7-2
Each −k digitally filters the input digital in-phase / quadrature signal x _k (m) based on a predetermined coefficient and outputs the result to the adder 10-2. Then, the adder 10-
2 adds the digital in-phase / quadrature signals x _k (m) after filtering and outputs the added signal y ₂ (m) to the signal selector 8. Here, the FIR type digital filter 7-2-
Each coefficient of k is set so as to form a wide band beam in a predetermined direction and output a signal y ₂ (m) corresponding to a wide band signal incident from the beam direction. In the same manner, the FIR digital filters 7-b-k respectively output digital in-phase / quadrature signals x _k (m) after filtering.
, And each of the adders 10-b (b = 3, 4, ..., B) outputs a signal y _b (m) corresponding to a predetermined beam to the signal selector 8. Here, in order to make the phase characteristics of the above-mentioned beams formed by the first multi-beam former 7a uniform and to realize distortion-free transmission, the FIR
The type digital filters 7-1-1 to 7-BN need to have the same linear phase characteristic with respect to frequency. As the FIR digital filter having such characteristics, a direct type configuration, a cascade type configuration, a lattice type configuration, and the like can be considered, and any configuration may be used in the second embodiment.

The signal selector 8 outputs a plurality of B signals y ₁ (m), y output from the first multi-beam former 7a.
₂ (m), ..., and selects M signals from y _B (m), the selected signal _{yn 1 (m), yn 2} (m), ..., yn
It outputs _M (m) to the multipliers 12-1 to 12-M, respectively. Here, the selection method in the signal selector 8 is as follows.
In the second embodiment, M signals are selected in descending order of power. However, the present invention is not limited to this, and another method such as selecting a signal having power equal to or higher than a predetermined threshold may be used. Further, the number M of signals to be selected is an integer equal to or less than the number of beams B and is determined according to the number of antenna elements N and the number of expected interference waves, but preferably from half the number of antenna elements N to 1 It is set to about / 4.

[0073] The multiplier 12-1 to 12-M each weighting factor wB ₁ of _{(m), wB 2 (m} ), ..., wB M (m) at the output of the adaptive digital beam forming apparatus, extracts a desired signal Then, the first coefficient controller 11a adaptively controls so as to suppress the interference signal. The number of weighting factors to be controlled is M. Each of these load factors wB ₁ (m), wB
_{2 (m), ..., wB} M (m) are generally complex numbers. The output signals of the multipliers 12-1 to 12-M are added by the adder 10 and the output signal z (m) which is the desired arrival signal in which the interference signal is suppressed is output.

Here, in the adaptive digital beam forming apparatus of the second embodiment, the first coefficient controller 11a is
The weighting factors wB ₁ (m) and wB that keep the envelope of the output signal z (m) constant by executing the processing described below.
₂ (m), ..., and calculates a wB _M (m). In this way, the weighting factor w is set so that the envelope of the output signal z (m) is kept constant.
_{B 1 (m), wB 2} (m), ..., the processing for calculating the wB _M (m) is called the CMA processors, the CMA treatment, for example, FM signals, FSK signals, effective for such PSK signal Therefore, there is an advantage that a reference signal is not required.

First, the load factors wB ₁ (m), wB
₂ (m), ..., the evaluation function J for calculating the wB _M (m)
Is defined as Equation 17, and the signal vector Y (m) at time m and the weighting factors wB ₁ (m), wB ₂ (m), ..., W
The vector W (m) consisting of B _M (m) is calculated as
8 is defined as in Expression 19.

[0076]

[Equation 17] J = (1/4) E [│z (m) │ ² −σ ² ]

Y (m) = [yn ₁ (m), yn ₂ (m), ..., yn _M (m)] ^T

W (m) = [wB ₁ (m), wB ₂ (m), ..., wB _M (m)] ^T

Here, in Expression 17, E [·] is an expected value and σ is a desired envelope value. Further, in Expressions 18 and 19, [·] ^T represents a transposed matrix. A vector that minimizes the evaluation function J of Expression 17 cannot be analytically obtained. Therefore, using the gradient method, the coefficient vector W
(M) is updated sequentially. The updating equation of the coefficient vector W (m) in this case can be expressed by the following Expression 20.
Further, the output signal z (m) is expressed by Expression 21.

[0078]

(20) W (m + 1) = W (m) −μ [{│z (m) │ ² −
σ ² } z (m) Y * (m)]

Equation 21] ^{z (m) = W T (} m) Y (m)

In Expression 20, μ is an appropriate positive constant, and μ
Is set so that the coefficient vector W (m) converges. Also, * represents a complex conjugate. The initial value W (0) of the coefficient vector W (m) is set to an appropriate value other than 0 for the weighting factor of the multiplier 12 to which the signal with the maximum power of the output signal of the signal selector 8 is input. , And other coefficients are set to 0.

Thus, the multiplier 12-
If the weighting factors of 1 to 12-M are updated, the FM signal, the FSK signal, and the PSK in which the envelope of the desired reception signal is constant
In the case of a signal or the like, the interference signal is suppressed and the desired signal is extracted. At this time, a wideband beam is formed in the incident direction of the desired signal, and a wideband null is formed in the incident direction of the interference signal. Although this method limits the types of signals, it does not require prior knowledge about the incident direction of the signal and is more practical than the coefficient calculation method with constraint conditions.
In the second embodiment, the load coefficient is updated according to the equation (20).

Next, the result of the simulation performed to confirm the operation of the adaptive digital beam forming apparatus according to the second embodiment will be described. FIG. 7 is a graph showing a radiation pattern of the array antenna 100 which is the result of the simulation. The graph of FIG. 7 shows the relative power with respect to the angle indicating the beam emission direction. Here, the angle indicating the radiation direction of the beam is the antenna element 1-
1 to 1-N are represented by an angle with respect to a vertical line of juxtaposed lines, and the relative power on the vertical axis is shown by standardizing the power of the main beam. In the graph of FIG. 7, the frequency f is set to the carrier frequency f _c and the frequency f =
And when set to 0.95f _c, the frequency f = 1.05f _c
The simulation is performed for three cases in which the above setting is made, and the respective cases are shown.

As is clear from the graph of FIG. 7, in both the case of the frequency f = f _c , the case of the frequency f = 0.95f _{c and} the case of the frequency f = 1.05f _c , the desired value is obtained. The main beam can be formed in the direction of 40 degrees which is the arrival direction of the signal, and the null can be formed in the direction of 60 degrees of the arrival direction of the interference signal. That is, a beam having a relatively wide band having a band of 10% with respect to the carrier frequency f _c can be formed in the incident direction of the desired signal, and 10% with respect to the carrier frequency f _c in the incident direction of the interference signal. It is shown that it is possible to form a relatively wide band null having a band of. That is, according to the adaptive digital beamforming apparatus of the second embodiment,
It is shown that a desired signal in a relatively wide band can be extracted and an interference signal in a relatively wide band can be suppressed.

The adaptive digital beam forming apparatus of the second embodiment having the beam space type structure configured as described above requires calculation of multi-beam forming processing.
The number of weighting factors to be controlled can be extremely reduced as compared with the adaptive beam forming apparatus for wide band having the element space type configuration of the first conventional example shown in FIG. This is advantageous because the amount of calculation and the amount of hardware required to control the weighting factor can be greatly reduced.

Further, the adaptive digital beam forming apparatus of the second embodiment described above can form a wide band beam without requiring any prior knowledge about the incident direction of the signal, and the multi-beam processing itself can be performed linearly. It can be anything. That is, it is superior to the adaptive digital beam forming apparatus of the second conventional example in the above points.

<Third Embodiment> FIG. 8 is a block diagram of an adaptive digital beam forming apparatus according to the third embodiment of the present invention. The adaptive digital beam forming apparatus according to the third embodiment will be described below with reference to FIG.

The adaptive digital beam forming apparatus according to the third embodiment shown in FIG. 8 is the same as the adaptive digital beam forming apparatus according to the second embodiment shown in FIG. 6, instead of the first multi-beam former 7a. The multi-beam former 7c is provided. Here, the third multi-beam former 7
c is a plurality of second multi-beam formers 7b (B × 2)
FIR digital filters 7c-1-1 to 7c
-B-2 and adders 10-1 to 10-B. In the third embodiment of FIG. 8, the same components as those of the second embodiment of FIG. 6 are designated by the same reference numerals.

In the adaptive digital beam forming apparatus of the third embodiment, the high frequency signals received by the N antenna elements 1-k are the same as those of the adaptive digital beam forming apparatus of the first and second embodiments, respectively. , Are converted into intermediate frequency signals by the receiver 2-k, and converted into digital intermediate frequency signals by the A / D converter 3-k. Then, these digital intermediate frequency signals are respectively mixed with the digital sine wave of the frequency f _IF oscillated by the digital local oscillator 60 by the mixer 4-k, and the high frequency component is cut by the low pass digital filter 5-k, A low-pass digital filter 5-k outputs a complex digital in-phase / quadrature signal x ₁ (m) x ₂ (m), ..., X _N (m), and the signal is a third multi-beam former. 7c is input.

The third multi-beam former 7c of the adaptive digital beam former of FIG. 8 will be described in detail with reference to FIG. The third multi-beam former 7c
In FIG. 11, the second multi-beam former 7b includes a plurality of (B × N) weight coefficient units 71-1−, as shown in FIG.
1 to 71-BN and a plurality of B adders 72-1 to 7
2-B. Then, in the second multi-beam former 7b, the digital in-phase / quadrature signal x ₁ (m) is input to the weight coefficient units 71-1-1 to 71-B-1,
The digital in-phase / quadrature signal x ₂ (m) is applied to the weight coefficient unit 71.
-1-2 to 71-B-2. Similarly, the digital in-phase / quadrature signal x _k (m) (k = 3, 4, ...,
N) is input to the load coefficient units 71-1-k to 71-B-k. As shown in FIG. 11, the load coefficient unit 71-1-k
(K = 1, 2, ..., N) multiplies the input digital in-phase / quadrature signal x _k (m) by the weighting coefficient α _1k, and adds the multiplication result x _k (m) · α _1k to the adder. 72-1 is output.
The adder 72-1 adds a plurality of N multiplication results x _k (m) · α _1k input from each weighting coefficient unit 71-1 -k, and outputs a signal y ₁ (m) as the addition result. It outputs to the FIR type digital filter 7c-1-1 and the FIR type digital filter 7c-B-2. Here, each load coefficient unit 71
Each weighting factor α _1k in -1-k is set to a predetermined value in advance so that a beam is formed in a predetermined direction and a signal y ₁ (m) corresponding to the beam is output from the adder 72-1. It is set.

Further, the weight coefficient unit 71-2-k receives the input digital in-phase / quadrature signal x _k (m) and the weight coefficient α.
_2k is multiplied and the multiplication result x _k (m) · α _2k is added by the adder 7
Output to 2-2. The adder 72-2 is used for each load coefficient unit 7
Multiple N multiplication results x input from 1-2k
_The signal y, which is the addition result, is obtained by adding _k (m) · α _2k.
2 (m) is output to the FIR type digital filter 7c-1-2 and the FIR type digital filter 7c-2-1.
Here, the load coefficient α in each load coefficient unit 71-2-k
_2k is set in advance to a predetermined value so that a beam is formed in a predetermined direction and a signal y ₂ (m) corresponding to the beam is output from the adder 72-1.

Further, the load coefficient unit 71-b-k (b = 3,
4, ..., B) are similarly input digital in-phase /
The quadrature signal x _k (m) is multiplied by the _weighting factor α _bk, and the multiplication result x _k (m) · α _bk is output to the adder 72-b. The adder 72-b adds a plurality of N multiplication results x _k (m) · α _bk input from each load coefficient unit 71-b-k,
The signal y _b (m) which is the addition result is output to the FIR digital filter 7c- (b-1) -2 and the FIR digital filter 7c-b-1. Here, each _weighting factor α _bk in each _weighting factor unit 71-b-k forms each beam in a predetermined direction and outputs a signal y corresponding to the beam.
It is preset to a predetermined value so that _b (m) is output from each adder 72-b. The second multi-beam former 7b configured as described above forms a beam in each of a plurality of B different preset directions, and a plurality of B signals y _b (m) corresponding to the beam. (B = 1,
2, ..., B) are output.

Here, a plurality of B signals y ₁ (m), y ₂ (m), output from the second multi-beam former 7b,
, Y _B (m) can be expressed by the following formula 22 using the _weighting factor α _bk . This number 22 holds even when the second multi-beam former 7b is configured by using the fast Fourier transform (FFT). However, in this case, the number of antenna elements N is set to the number of beams B.

[0092]

(Equation 22)

Further, the radiation power intensity H _b (f, θ) of the beam when the beam is formed by using the second multi-beam former 7b can be expressed by the following equation 23.

[0094]

(Equation 23)

In Equation 23, b is the beam number, d is the element spacing between adjacent antenna elements 1-k, and a linear array with equal spacing is assumed. Also, f
_c is the carrier frequency and λ _c is the wavelength of the carrier. Further, θ is the incident angle of the desired signal, and the antenna element 1-k
Is defined by the angle between the perpendicular of the juxtaposed lines and the incident direction of the desired signal. Here, considering the case where the second multi-beam former 7b is configured by using the fast Fourier transform (FFT), the _weighting factor α _bk can be expressed by the following equation 24.

[0096]

Α _bk = (1 / B) · exp {j2π (b-1) (k-1) /
B}, k = 1, 2, 3, ..., N

In Expression 24, b = 1 corresponds to a beam having an incident angle θ = 0 °. When 0 ° is substituted for the incident angle θ of the equation 23, the radiated power intensity H ₁ (f, θ) represented by the equation 23 becomes a constant that does not depend on the frequency f. Therefore, the broadband signal incident at the incident angle θ = 0 ° is received as it is. But,
The radiation power intensity H _b (f, θ) of beams other than b = 1 depends on the frequency f, and the reception power of a signal having a frequency apart from the carrier frequency f _c is attenuated. That is, the incident angle θ = 0
Since the second multi-beam former 7b operates as a kind of band pass filter for a signal incident from a direction other than °, a wide band signal having a wider frequency range than the pass band of the band pass filter. Among them, a signal outside the pass band having a frequency away from the carrier frequency f _c is attenuated and output as a narrow band signal. Here, the pass band of the band pass filter is determined corresponding to the incident angle θ. In this way, the load coefficient unit 71-b-k preset with the load coefficient α _bk as a value that does not depend on frequency is set.
The second multi-beam former 7b configured using
Since it operates as a kind of band pass filter for a signal incident from a direction other than the incident angle θ = 0 °, in the third embodiment in which the desired signal has a wide band, it is narrower than the desired signal. The signals y ₁ (m), y ₂ (m), ..., Y _N (m), which are a plurality of B beam reception signals in the band, are output.

In addition, the third multi-beam former 7 of FIG.
In c, one pair of FIR digital filters and one adder are provided for each of the B wideband beams to be formed. That is, the FIR digital filters 7c-1-1 and 7c-1-2 and the adder 10-1 are provided corresponding to the first wideband beam to be formed. Then, the FIR digital filter 7c-1-1 digitally filters the input narrow band beam reception signal y ₁ (m) with a weighting factor which is a preset filter factor. Output to the adder 10-1, and the FIR digital filter 7c
-1-2 digitally filters the input narrow band beam reception signal y ₂ (m) with a weighting coefficient which is a preset filter coefficient, and outputs it to the adder 10-1. To do. The adder 10-1 adds the input two filtered signals y ₁ (m) and y ₂ (m), and adds the added signal v ₁
(M) is output to the signal selector 8. Here, each weighting factor of the FIR digital filters 7c-1-1 and 7c-1-2 forms a first wideband beam in a predetermined direction and is a wideband received signal corresponding to the beam. The signal v ₁ (m) is set to be output from the adder 10-1.

The FIR type digital filter 7c-2-corresponding to the second wide band beam to be formed.
1, 7c-2-2 and an adder 10-2 are provided. Then, the FIR digital filter 7c-2-1 digitally filters the input narrow band beam reception signal y ₂ (m) with a weighting coefficient which is a preset filter coefficient. The FIR digital filter 7c-2-2 outputs the signal to the adder 10-2, and the signal y ₃ (m), which is the narrow-band beam reception signal that is input, with a weighting coefficient that is a preset filter coefficient. It is digitally filtered and output to the adder 10-2. The adder 10-2 adds the two input signals y ₂ (m) and y ₃ (m) after filtering and outputs the added signal v ₂ (m) to the signal selector 8. Here, the FIR digital filter 7c-2-
Each of the weighting factors 1, 7c-2-2 forms a second wideband beam in a predetermined direction, and adds the signal v ₂ (m), which is a wideband received signal corresponding to the beam, to the adder 10 −
It is set to output from 2.

Similarly, the b-th to be formed (b = 3,
4, ..., B-1) corresponding to the broadband beam
Type digital filters 7c-b-1, 7c-b-2 and an adder 10-b are provided. Then, the FIR digital filter 7c-b-1 digitally filters the input narrow band beam reception signal _yb (m) with a weighting factor which is a preset filter factor. Output to the adder 10-b, and the FIR digital filter 7
c-b-2 digitally filters the input narrow band beam reception signal y _{b + 1} (m) with a weighting coefficient which is a preset filter coefficient, and adds it to the adder 10- b
Output to The adder 10-b adds the two input signals y _b (m) and y _{b + 1} (m) after filtering, and outputs the added signal v _b (m) to the signal selector 8. To do. Where FI
The respective weighting factors of the R-type digital filters 7c-b-1 and 7c-b-2 form a b-th wide band beam in a predetermined direction, and a signal v which is a wide band received signal corresponding to the beam. _b (m) is set to be output from the adder 10-b.

The FIR digital filter 7c-B- corresponding to the B-th broadband beam to be formed.
1, 7c-B-2 and an adder 10-B are provided. Then, the FIR digital filter 7c-B-1 receives the signal y _B (m) which is the narrow-band beam reception signal to be input.
Is digitally filtered by a weighting coefficient which is a preset filter coefficient and is output to the adder 10-B. The FIR digital filter 7c-B-2 is a narrowband beam reception signal to be input. A certain signal y ₁ (m) is digitally filtered by a preset weighting coefficient, which is a filter coefficient, and output to the adder 10-B. The adder 10-B adds the two input signals y _B (m) and y ₁ (m) after filtering, and outputs the added signal v _B (m) to the signal selector 8. Here, the FIR digital filter 7c-B-
Each weighting factor of 1 and 7c-B-2 forms a B-th wide band beam in a predetermined direction, and a signal v _B (m) which is a wide band received signal corresponding to the beam is added by the adder 10 −
It is set to output from B.

That is, in the third multi-beam former 7c of FIG. 8, the second multi-beam former 7b cannot cover the frequency band of the wide band desired signal as described above. Therefore, after the second multi-beam former 7b, a pair of FIR type digital filters having a predetermined weighting factor value corresponding to a wide-band beam to be formed and an adder are connected to each other so that the wide-band multi-beam is formed. Forming a beam. In FIG. 8, the second multi-beam former 7
Two adjacent beams are selected from the beams formed in b to form a broadband beam in a predetermined direction. However, the invention need not be two adjacent beams. The number of selections is not limited to two, and two or more signals among the signals y _k (m) output from the second multi-beam former 7b are selected to form a wideband beam. Good.

Here, the FIR digital filter 7c
The weighting factors, which are the filter coefficients of −b-1 and 7c-b-2, form a beam in a predetermined direction by using the same method as that of the FIR type digital filter coefficient calculator 13 of the first embodiment. The signal v _b (m) corresponding to the broadband signal incident from the beam direction may be set to be output, or another method may be used. The FIR digital filters 7c-b-1 and 7c-b-2 may have a direct configuration, a cascade configuration, a lattice configuration, or the like.
Any configuration may be used in the third embodiment.

As in the second embodiment, the signal selector 8 outputs the B signals v ₁ (m), v ₂ (m), ..., V _B output from the third multi-beam former 7c. Select M signals from (m) and select the selected signals yn ₁ (m), yn
₂ (m), ..., Yn _M (m) are respectively multiplied by the multiplier 12-1.
To 12-M.

The multipliers 12-1 to 12-M input the respective signals yn, respectively, as in the second embodiment.
_{1 (m), yn 2 (} m), ..., yn M (m) and the respective weighting factor _{wB 1 (m), wB 2} (m), ..., and output by multiplying the wB _M (m), adding The multiplier 10 is a multiplier 12-1 to 12-M.
Output signals z (m), which are desired incoming signals with interference signals suppressed, are output.

The adaptive digital beam forming apparatus of the third embodiment configured as described above has the same effects as the adaptive digital beam forming apparatus of the second embodiment, and at the same time, the adaptive digital beam forming apparatus of the second embodiment is used. The amount of calculation in wideband multi-beam forming can be further reduced as compared with the beam forming apparatus.

<Fourth Embodiment> Next, an adaptive digital beam forming apparatus according to the fourth embodiment will be described. The adaptive digital beam forming apparatus according to the fourth embodiment is
The configuration is the same as that of the adaptive digital beam forming apparatus according to the first embodiment of FIG. 1, but the calculation method of the weight coefficient w _{k, s} is different. That is, in the adaptive digital beam forming apparatus according to the first embodiment described above, a variable that rotates the pass region of the one-dimensional linear phase FIR narrow band low-pass digital filter on the two-dimensional frequency plane and then translates the same. By performing the conversion, a two-dimensional digital filter D
Although the amplitude characteristic of the BF circuit 6 is approximated to a desired characteristic to calculate the weighting coefficient w _{k, s} , in the adaptive digital beam forming apparatus of the fourth embodiment, the one-dimensional zero-phase FIR type low-pass digital filter is used. The frequency response of (hereinafter referred to as a one-dimensional original filter) is converted into a predetermined straight line corresponding to the beam forming direction on the two-dimensional frequency plane in which the frequency in the beam direction formed by the digital beam forming device in the predetermined direction. As described above, by performing variable conversion, the amplitude characteristic of the DBF circuit 6 is approximated to a desired characteristic, and the weight coefficient w _{k, s} is calculated. The configuration is similar to that of the first embodiment except the above points. Hereinafter, the load factor calculation method of the fourth embodiment will be described in detail.

Here, the variable conversion applied to the one-dimensional zero-phase FIR type low-pass digital filter will be described first, and then the calculation method of the weighting factor w _{k, s} of the DBF circuit 6 will be described.

The variable conversion in the fourth embodiment is the first
The cos appearing in the number 8 embodiment of the equation of the frequency response at the F ₂ = F (2πF), suitable four functions cos (2πF _1), s
This is performed by replacing the function S (F ₁ , F ₂ ) represented by in (2πF ₁ ), cos (2πF ₂ ), and sin (2πF ₂ ). here,
The function S (F ₁ , F ₂ ) has the amplitude characteristic D _T (F ₁ ,
This is a function that approximates F ₂ ) and will be described in detail later.

[0110]

[Expression 25] D _T (F ₁ , F ₂ ) = cos [2πF ₂ / {(f _s / f _c ) F ₁ +
1} −2πF _2shift ]

Here, the F 2 _shift in _equation 25 is the first
In the shift amount represented by the same formula as the equation 13 in the embodiment of
It is determined by the angle of incidence θ corresponding to the direction in which the beam is to be formed. Further, p (m) in the equation 8 is the impulse response of the one-dimensional original filter in the fourth embodiment. The amplitude characteristic D _T (F ₁ , F ₂ ) of the equation 25 can be derived by using the equations 6 and 13, and an arbitrary one-dimensional frequency F (actually, some frequencies) of the one-dimensional prototype filter can be derived. Is excluded) to a straight line on the two-dimensional frequency plane represented by the following equation 26.

[0112]

[ _Equation 26] F = F ₂ / {(f _s / f _c ) F ₁ +1} −F 2 _shift

For example, the one-dimensional frequency F = 0 of the one-dimensional original filter is a straight line F ₂ = F _2shift {(f
_s / f _c ) F ₁ +1}. Therefore, if this mapping is applied to the one-dimensional FIR type narrow band low-pass digital filter used in the first embodiment, the pass band is a straight line F ₂ = F _2shift {(f _s / It will appear in the vicinity of f _c ) F ₁ +1}. In other words, in order to form a beam in the direction of the incident angle θ, the passband is F on the two-dimensional frequency plane.
₂ = (df _c sin θ / c) {(f _s / f _c ) F ₁ +1} appears near,
The shift amount F ₂ shift is determined according to _Equation 13, and the one-dimensional FI
First for R type narrow band low pass digital filter
Instead of the variable conversion method used in the above embodiment, the mapping of Equation 25 or Equation 26 may be performed. FIG. 12 shows an example when mapping is performed by using Expression 26. In this example, assuming that the spacing d between antenna elements is d = 0.45λ _c and the beam forming direction θ = 40 °, the one-dimensional frequency F is between +0.5 and −0.5, and the spacing is 0.1. It is divided and shown for each one-dimensional frequency F.

If the above-mentioned mapping is ideally performed, each equal-amplitude line of the amplitude characteristic of the two-dimensional digital filter obtained by such variable conversion will follow the straight line of the equation 6 in the first embodiment. . On the other hand, the straight line represented by Expression 6 corresponds to a straight line parallel to the frequency axis at the angle θ on the frequency-angle plane. Therefore, the fact that the equal-amplitude line of the amplitude characteristic of the two-dimensional digital filter on the two-dimensional frequency plane follows Expression 6 means that the equal-amplitude line is parallel to the frequency axis on the corresponding frequency-angle plane. This means that the directional characteristic of the DBF circuit 6 does not depend on the frequency. That is, the directional characteristic of the DBF circuit 6 as the two-dimensional digital filter obtained by such variable conversion depends only on the incident angle θ and does not depend on the frequency. In other words, in the fourth embodiment, the one-dimensional FIR narrow band low-pass digital filter is subjected to variable conversion so that the directivity of the DBF circuit 6 as a two-dimensional digital filter does not depend on frequency. .

Next, the amplitude characteristic D _T (F ₁ , F ₂ ) of Equation 25 is obtained.
A method of obtaining the function S (F ₁ , F ₂ ) that approximates This approximation is based on the window function method in which the impulse response to the frequency characteristic represented by the equation 25 is truncated by a two-dimensional window function (Prior Art Document 5, “Nishikawa et al.,“ Fan Filter Design by Fourier Transform Method ”, IEICE). In the 6th Digital Signal Processing Symposium Proceedings, Lecture No. B3-2, pp.287-292, 1991 ", it is called the Fourier series method.). The specific procedure is shown in the following procedures SS1 to SS4.

(Procedure SS1) The normalized temporal frequency F ₁ and the normalized spatial frequency F ₂ are -0.5≤F ₁ <0.5,-
Sampling is performed at appropriate intervals in the range of 0.5 ≦ F ₂ <0.5, and each sampled normalized time frequency F ₁
And the amplitude characteristic D for each normalized spatial frequency F ₂ .
Calculate each frequency response value of _T (F ₁ , F ₂ ). When the fast Fourier transform is used later, the number of sampling points J
(Even number) is preferably set to a value that can be represented by a power of 2 such as 128 points. (Procedure SS2) Amplitude characteristic D _T (F ₁ ,
Each frequency response value of F ₂ ) is subjected to two-dimensional inverse discrete (fast) Fourier transform to obtain an impulse response (complex number) for the frequency response value of the amplitude characteristic D _T (F ₁ , F ₂ ). The impulse response is called an ideal impulse response, and h _TI (m ₁ ,
m ₂ ). Here, (m ₁ , m ₂ ) = (0, 0) is set to be the center of the ideal impulse response h _TI (m ₁ , m ₂ ). The arguments m ₁ and m ₂ are −J / 2 and (−J /
2) +1, ...,-1,0,1, ..., (J / 2) -1. (Procedure SS3) The ideal impulse response h _TI (m ₁ , m ₂ ) is truncated by the two-dimensional window function w _2D (m ₁ , m ₂ ). That is, the ideal impulse response h _TI (m ₁ , m ₂ ) and the two-dimensional window function w _2D
Take the product of (m ₁ , m ₂ ). Here, the two-dimensional window function w
_2D (m ₁ , m ₂ ) is represented by the following equation 27, for example.

[0117]

[Expression 27] w _2D (m ₁ , m ₂ ) = w _1D (m ₁ ) w _1D (m ₂ )

[0118] In Equation 27, w _1D (m _i), to the truncation number of terms L _a is an odd number, an argument m _i is the following equation 28
If the center of the window function is m _i = 0 (i =
1 and 2) is represented by an appropriate window function other than a rectangular window such as a Hamming window function and does not satisfy Eq. 28,
It is a window function whose function value is 0.

[0119]

(28) − (L _a −1) / 2 ≦ m _i ≦ (L _a −1) / 2

[0120] Here, the truncation number of terms L _a, differs from the number 10, number 11 truncation number of terms L in the first embodiment. The impulse response h _T ′ (m ₁ , m ₂ ) obtained by this operation is expressed by the following equation 29.

[0121]

H ′ _T (m ₁ , m ₂ ) = h _TI (m ₁ , m ₂ ) w _2D (m ₁ , m ₂ ) where − (L _a −1) / 2 ≦ m _i ≦ ( L _a -1) / 2, i
= 1,2

(Procedure SS4) Scaling is performed. This is the procedure S so as not to deteriorate the characteristics of the DBF circuit 6.
This is an operation of setting the maximum value and the minimum value of the frequency response value of the impulse response h _T '(m ₁ , m ₂ ) obtained in S3 to 1 and -1, respectively. Therefore, the frequency response value of the impulse response h _T '(m ₁ , m ₂ ) obtained in (procedure SS3) is calculated, and the maximum value h _T ' _max and the minimum value h _T ' _min thereof are obtained. Then, the scaling constants c ₁ and c ₂ represented by the following formula 30 are calculated, and these scaling constants c ₁ and c ₂ are used to calculate the following formula 31
Alternatively, the scaled two-dimensional impulse response h _T (m ₁ , m ₂ ) represented by Expression 32 is obtained. In the two-dimensional impulse response h _T (m ₁ , m ₂ ), h _T (m ₁ , m ₂ ) = h _T * (−
m ₁ , −m ₂ ) and therefore its frequency response S
(F ₁ , F ₂ ) is a real number, and cos (2πF ₁ ), sin
(2πF ₁ ), cos (2πF ₂ ), sin (2πF ₂ )
Function. Here, * of h _T * (− m ₁ , −m ₂ )
Represents a conjugate complex number.

[0123]

[Equation 30] c ₁ = 2 / (h ′ _Tmax −h ′ _Tmin ), c ₂ = c ₁
h ' _Tmax -1

When (m ₁ , m ₂ ) ≠ (0,0), h _T (m ₁ ,
m ₂ ) = c ₁ h ′ _T (m ₁ , m ₂ )

When (m ₁ , m ₂ ) = (0, 0), h _T (m ₁ ,
m ₂ ) = c ₁ h ′ _T (0,0) −c ₂

Next, a method of calculating the weighting factor in the DBF circuit 6 using the two-dimensional impulse response h _T (m ₁ , m ₂ ) obtained by the above procedure will be described. In the following calculation method, the frequency response of the two-dimensional impulse response h _T (m ₁ , m ₂ ) becomes the function S (F ₁ , F ₂ ). The calculation method of the weighting factor in the DBF circuit 6 includes the following steps S1 to S3.

<Step S1> First, the frequency response P (F) of the one-dimensional linear phase FIR type low-pass digital filter whose impulse response length is M _c (odd number) is obtained. 1
The frequency response P (F) of the dimensional linear phase FIR type low-pass digital filter is represented by the following equation 32 by setting N _i = M _c and F ₂ = F in equation 8. Here, the fourth
In the embodiment, the number of taps M _c is set smaller than the number N of elements in order to obtain good characteristics.

[0126]

(Equation 32)

<Step S2> Next, variable conversion is performed in which cos (2πF) on the right side of Expression 32 is replaced by the function S (F ₁ , F ₂ ) expressed by Expression 34. Therefore, the frequency response G (F ₁ , F ₂ ) after replacement is represented by the following expression 33. here,
In Expression 33, only the zero phase portion is shown. Similarly, in the mathematical expressions described below, only the zero phase portion is shown.

[0128]

[Equation 33]

(Equation 34)

<Step S3> Next, similarly to the first embodiment, the frequency response G (F ₁ , F ₂ ) represented by the equation 33 is expressed.
Impulse response g (m ₁ , m ₂ ) is calculated, and the load coefficient w _{n, k} is _obtained from the impulse response g (m ₁ , m ₂ ).

[0130]

W _{n, k} = g (k− (q−1) / 2, −n + (N−1) / 2 + 1) where n = 1, 2, ..., N; k = 0, 1, …, Q-1
And q is the number of taps.

The equi-amplitude line of the amplitude characteristic on the frequency-angle plane of the DBF circuit 6 obtained by the above method appears approximately parallel to the frequency axis on the frequency-angle plane. Therefore, the characteristic of the one-dimensional original filter can be set to a bandwidth according to a desired beam width instead of a narrow band, and the side lobe level can be lowered by increasing the attenuation amount of the corresponding stop band in a specific frequency range. As a result, the DBF circuit 6 of the fourth embodiment can form a wide-angle and wide-angle beam as will be described later in detail, and further, within a range of a predetermined direction, the sidelobe level can be obtained. Can be lower than the side lobe level in other regions. It should be noted that the method of the prior art document 5 cannot lower the side lobe level within the range of the predetermined direction below the side lobe level of other areas.

Further, the phase characteristic of the DBF circuit 6 in the fourth embodiment is linear, and the slope of the phase shift characteristic is determined by the value of the tap number q. Therefore, when a wide band multi-beam is formed using this method, each single beam DB
The number of taps q may be made uniform in the F circuit.

Next, a method of calculating the weighting factor of the DBF circuit 6 so that the side lobe level in the range of the predetermined direction is made lower than the side lobe levels of other regions will be described. In the following description, it is assumed that the direction in which a beam is formed is an angle θ _0, and the angle θ in a direction range in which a side lobe level is to be lowered is θ ₁ ≦ θ ≦ θ ₂ . Here, it is assumed that θ ₀ > θ ₂ . That is, the direction range in which the side lobe level is desired to be reduced is smaller than θ ₀ . However, the present invention is not limited to this, and θ ₀ <θ ₁ may be set, and a plurality of direction ranges in which the side lobe level is desired to be reduced may be provided.

By using the variable transformations described above, FIG.
The desired characteristic in the frequency-angle plane shown in FIG. 3 (a) can be expressed in the two-dimensional frequency plane as shown in FIG. 13 (b). That is, the one-dimensional digital filter to be designed is not the low-pass digital filter in the first embodiment, but is generally a linear phase FIR type band-pass digital filter having complex coefficients (hereinafter referred to as one-dimensional complex coefficient band-pass digital filter). It is). The one-dimensional complex coefficient band pass digital filter has a narrow band pass band centered on the frequency F ₀ expressed by the following formula 36 and has a frequency range F expressed by the following formula 37.
The side lobe level in ₀₁ ≤ F ≤ F ₀₂ is set to be lower than the side lobe levels in other stopbands. This is described, for example, in Prior Art Document 6 “Akiho Nishihara et al.,“ R
Design of Complex Digital Filter by emez Algorithm ”, IEICE Technical Report, CAS88-20, p
p.37-42, June 1988 ”can be used.

[0135]

F ₀ = (df _c / c) sin θ ₀

F _0i = (df _c / c) sin θ _i , where i = 1, 2

Regarding the frequency response, only the zero phase part is considered. Frequency response P of one-dimensional zero-phase FIR complex coefficient digital filter with impulse response length M _c (odd)
_c (F) can be expressed as in Eq. Here, the impulse response p _c (m) _{is, p c (m) = p} c * - a relationship of (m) (where * denotes the complex conjugate.). Also, the number of taps M _c
Is selected in the same manner as step S1 of the method for calculating the weighting factor in the DBF circuit 6.

[0137]

(38)

Here, U _m (cos (2πF)) in the equation 38
Is a Chebyshev function of the second kind, and
There is a relationship of 40 and 41.

[0139]

[Formula 39] U ₀ (cos (2πF)) = 0

[Formula 40] U ₁ (cos (2πF)) = sin (2πF)

[Formula 41] U _m (cos (2πF)) = 2 cos (2πF) U _m-1 (cos (2
πF)) − U _m−2 (cos (2πF)) where m = 2, 3,.

The difference between the expression 38 and the expression 8 in the first embodiment is that the frequency response P _c (F) includes sin (2πF). Therefore, cos (2πF) of Equation 38
Substituting the function S _c (F ₁ , F ₂ ) into, and sin (2πF)
The variable conversion for substituting the function S _s (F ₁ , F ₂ ) into Here, the function S _c (F ₁ , F ₂ ) and the function S _s (F ₁ , F ₂ ) are the D _TC (F ₁ , F ₂ ) of the equation 42 and the D _TS (F ₁ , F ₂ ) of the equation 43, respectively. ) Appropriate four functions cos (2πF ₁ ), sin (2πF ₁ ), cos
It is a function represented by (2πF ₂ ), sin (2πF ₂ ), and is represented by the following formula 42 and formula 43.

[0141]

(42) S _c (F ₁ , F ₂ ) ≈D _TC (F ₁ , F ₂ ) = cos [2πF _{2
/ {(f s / f c} ) F 1 +1}]

S _s (F ₁ , F ₂ ) ≈D _TS (F ₁ , F ₂ ) = sin [2πF _{2
/ {(f s / f c} ) F 1 +1}]

Here, D in the equations 42 and 43
_{In Equation} 25, _TC (F ₁ , F ₂ ) and D _TS (F ₁ , F ₂ ) are F _2shift = 0 and F _2shift = 0.25, respectively. It can be obtained by approximation using the method of step SS4).

The frequency response G _c (F ₁ , F ₂ ) of the two-dimensional digital filter obtained as described above can be expressed by the following equation 44. The frequency response of this number 44 G _c (F ₁ , F ₂ )
Using, the impulse response is calculated in the same manner as in the first embodiment, and the weighting factor of the DBF circuit 6 is obtained.

[0144]

[Equation 44]

As described above, since the variable conversion is performed using the equation 25, it becomes possible to reduce the sidelobe level in a specific direction without using the conventional optimization method.

For wide-angle beam forming, the range in the beam forming direction is set to θ ₁ ≦ θ ≦ θ ₂ and the range of the pass band of the one-dimensional complex coefficient band pass digital filter is given by
The frequency range represented by 7 may be set to F ₀₁ ≦ F ≦ F ₀₂ . Therefore, in the fourth embodiment, a wide-angle beam can be formed and the side lobe level in a specific direction can be lowered.

In the above description, θ ₀ > θ ₂ , that is, the range in which the side lobe level is desired to be lowered is smaller than θ _0. However, when θ ₀ <θ ₂ and the side lobe level should be lowered. It is also possible to provide a plurality of. That is, in the fourth embodiment, at least one direction in which the side lobe level should be lowered can be set, and the direction can be set at any position.

In the above fourth embodiment, since the variable conversion is performed using the equation 25 or the equation 26, the straight line corresponding to the incident angle θ on the frequency-angle plane is expressed by the equation on the two-dimensional frequency plane. It is converted into a predetermined straight line represented by 6. Therefore, on the frequency-angle plane, the equal amplitude lines of the DBF circuit 6 as the two-dimensional digital filter are parallel to the frequency axis. Therefore, the directional characteristic of the DBF circuit 6 obtained by such variable conversion can be made to depend only on the incident angle θ and not on the frequency. In other words,
In the fourth embodiment, variable conversion is applied to the FIR type narrow band low pass digital filter so that the directional characteristic of the DBF circuit 6 does not depend on frequency.

As described in detail above, in the fourth embodiment, an arbitrary frequency in the beam direction formed by the digital beam forming apparatus in the predetermined direction is represented by the formula 6 on the two-dimensional frequency plane. By performing variable conversion on the frequency response of the one-dimensional linear phase FIR digital filter so as to be converted to the straight line of, the normalized time frequency F ₁ and the normalized spatial frequency F ₂ corresponding to the desired pass region are obtained. And the frequency response of the two-dimensional digital filter is calculated, and the calculated frequency response is inverse-Fourier-transformed to calculate the impulse response of the two-dimensional digital filter, and the two-dimensional digital filter is calculated based on the calculated impulse response. So that each has an amplitude in the desired pass region, each FIR type digital filter 9-1
The weighting factors w _{k, s} of 9 to N are calculated. As a result, the adaptive digital beam forming apparatus according to the fourth embodiment can form a wide-angle beam with a wide band, and lowers the side lobe level in other than the side lobe level in other regions in a predetermined direction. You can

Next, the result of the simulation in the fourth embodiment will be described. FIG. 14 shows a wideband beam θ = 56 ° using the DBF circuit 6 of the fourth embodiment.
7 is a graph showing the directional characteristics of the DBF circuit 6 when it is formed in the direction. The graph of FIG. 14, f = 0.95f _c,
For three frequencies of f = f _c and f = 1.05f _c ,
The relative electric power with respect to the angle θ is shown. As is clear from FIG. 14, the directional characteristics at the above-mentioned three frequencies are substantially the same. That is, from the graph of FIG.
It can be seen that the DBF circuit 6 according to the embodiment can form a beam that does not depend on the frequency of a relatively wide band of ± 5% with respect to the center frequency f = f _c .

FIG. 15 shows the DBF of the fourth embodiment.
A wideband beam is formed in the direction of θ = 60 ° by using the circuit 6, and the side lobe level in the direction range of −50 ° ≦ θ ≦ −20 ° is about 2 than the side lobe level of other regions.
It is a graph which shows the directivity when it is lowered by 0 dB. Also in FIG. 15, similarly to FIG. 14, f = 0.95f _c ,
For three frequencies of f = f _c and f = 1.05f _c ,
The relative electric power with respect to the angle θ is shown. As is clear from FIG. 15, by using the DBF circuit 6 of the fourth embodiment, a wide band beam can be formed in the direction of θ = 60 ° and the side at −50 ° ≦ θ ≦ −20 ° can be formed. It can be seen that the lobe level can be set lower than the side lobe levels of other regions by about 20 dB.

Further, in FIG. 16, the above-mentioned weighting factor calculation method is used for the single beam former (DBF circuit) which constitutes the first multi-beam former 7a, and adaptive beam forming explained in the second embodiment is performed. The directional characteristic when the interference wave is suppressed when the method is used is shown. In the figure, S1 indicates a desired wave with an incident angle of 25 °, and I1 and I2 indicate an interference wave with an incident angle of 35 ° and an interference wave with an incident angle of −50 °, respectively. Also in FIG. 16, FIG.
4, f = 0.95f _c , f = f _c and f = 1.0
For three frequencies of 5f _c, shows the relative power for the angle theta. As is apparent from FIG. 16, by using the DBF circuit 6 of the fourth embodiment, a wide band beam can be formed in the desired wave direction of θ = 25 °,
Moreover, nulls can be formed in the arrival directions of the interference wave with the incident angle of 35 ° and the interference wave with the incident angle of −50 °. Here, any of the graphs in FIGS. 14 to 16 has a sampling frequency.
f _s was set to 0.2f _c .

In the DBF circuit 6 of the fourth embodiment described above, the side lobe level can be controlled,
In addition, it is possible to lower only a part (a range of a certain direction) of the side lobe level. In addition, the broadband DB that constitutes the first multi-beam former 7a in the second embodiment.
It can be used as the F circuit 6 and can form a null in the direction of the interference wave even when the incident angles of the desired wave and the interference wave are distant from each other. Therefore, the DB of the fourth embodiment
Although the F circuit 6 can be used for forming a single beam, it is preferably used as the DBF circuit 6 forming the first multi-beam former 7a.

<Modification> In the above first and second embodiments, the case where the propagating wave signal is a radio wave and the information transmission using the carrier wave is described, but the present invention is not limited to this, and the carrier wave is not used. It can also be applied to the case of baseband transmission. In this case, since the frequency conversion operation is unnecessary, the receiver 2-k is configured and the digital in-phase / quadrature signals x ₁ (m), x ₂ (m), ..., X _N (m) are real-valued. May be taken. Even with the above configuration, the same effects as those of the first and second embodiments are obtained. However, in this case, it is necessary to properly change the shape of the pass band of the two-dimensional digital filter.

In the above-described first to fourth embodiments, the digital in-phase / quadrature signal x _{k is} converted from the digital intermediate frequency signal.
Although the mixer 4-k and the low-pass digital filter 5-k are used as the means for generating (m), other methods may of course be used. For example, it can be realized by using a digital Hilbert transformer and a mixer. In addition, analog signal processing enables analog in-phase and analog in-phase
A quadrature signal may be generated and then sampling and A / D conversion may be performed. Even with the above configuration, the same effects as those of the first, second, and third embodiments can be obtained.

In the adaptive digital beam forming apparatus of the second and third embodiments, the CMA process for calculating the weighting coefficient w _{k, s} is executed so that the envelope of the output signal z (m) is kept constant. However, the present invention is not limited to this,
If the reference signal is obtained, it may be configured to execute other processing such as LMS processing. Even with the above configuration, the same effects as those of the second and third embodiments can be obtained.

The adaptive digital beam forming apparatus according to the first to fourth embodiments described above has the antenna elements 1-1 to 1-
Although the array antenna 100, which is a linear array antenna in which N is arranged in a straight line, is used, the present invention is not limited to this, and an array antenna in which antenna elements are two-dimensionally arranged may be used. Good. Even with the above configuration, the same effects as those of the first to fourth embodiments are obtained.

[0158]

According to the first aspect of the present invention, there is provided a filter coefficient computing device, which comprises a normalized time frequency F ₁ normalized by a sampling frequency of the digital beam forming device,
Normalized spatial frequency F normalized by the reciprocal of the above element spacing
_{Since the} coefficient calculating means for calculating the filter coefficients of the plurality of FIR type digital filters having the amplitude in the desired passing region on the two-dimensional frequency plane formed by ₂ and ₂ is provided, it is possible to transmit information using a carrier wave which is a radio wave. FI for applicable digital beam former
The filter coefficients of the R-type digital filter can be calculated.

The filter coefficient calculation device of the FIR digital filter for a digital beam forming device according to claim 2 is the filter coefficient calculation device according to claim 1, wherein the coefficient calculation means is a predetermined one-dimensional FIR. The frequency response of the digital filter is converted into a predetermined variable to calculate the frequency response corresponding to the desired pass region and with respect to the normalized time frequency F ₁ and the normalized spatial frequency F ₂ . The impulse response is calculated by inverse Fourier transforming the frequency response, and the filter coefficients of the plurality of FIR digital filters having the amplitude in the desired pass region are calculated based on the calculated impulse response. Fast and easy filter coefficient of FIR type digital filter for forming device Can be calculated.

Further, the filter coefficient calculation device of the FIR type digital filter for the digital beam forming apparatus according to claim 3 is the filter coefficient calculation device of the FIR type digital filter according to claim 1 or 2, wherein the coefficient calculation means is used. In the two-dimensional frequency plane has a predetermined amplitude in the desired pass region, has a predetermined width centered on the second axis, and has a predetermined width. Of the plurality of F's having a predetermined amount of attenuation in the region where the sidelobe level extending to the signalization time frequency should be lowered.
Since the filter coefficient of the IR type digital filter is calculated, when the arrival directions of the desired wave and the interference wave are assumed, a beam is formed in the arrival direction of the desired wave and a null is formed in the arrival direction of the interference wave. The filter coefficient of the FIR type digital filter for the digital beam forming apparatus can be calculated at high speed and easily.

The filter coefficient calculation device of the FIR type digital filter for the digital beam forming apparatus according to claim 4 is the filter coefficient calculation device of the FIR type digital filter according to claim 2, wherein the coefficient calculation means is: The pass region of the one-dimensional FIR digital filter is rotated on the two-dimensional frequency plane, and the rotated pass region is translated on the two-dimensional frequency plane to perform variable conversion, and thus the desired pass region. Corresponding to the above-mentioned normalized time frequency F ₁ and the above-mentioned normalized spatial frequency F ₂ are calculated, and the impulse response is calculated by inverse Fourier transforming the calculated frequency response, and the calculated impulse response is calculated. Of the plurality of FIR digital filters having an amplitude in the desired pass region based on The data coefficient is calculated. As a result, the filter coefficient of the FIR type digital filter for forming the beam in the arrival direction of the desired wave and forming the null in the vicinity of the beam can be calculated at high speed and easily.

According to a fifth aspect of the present invention, there is provided a filter coefficient calculation device for a FIR digital filter for a digital beam forming apparatus according to the second or third aspect, wherein the coefficient calculation means is the filter coefficient calculation device. Responds to the desired pass region by converting the frequency response of the one-dimensional FIR digital filter into a variable so that the beam pattern formed by the digital beam forming device does not change with frequency, and The impulse response is calculated by calculating the frequency response related to the normalized time frequency F ₁ and the normalized spatial frequency F _2, and performing the inverse Fourier transform on the calculated frequency response, and based on the calculated impulse response, the desired impulse response is calculated. The plurality of FIs having a predetermined amplitude in the pass region
The filter coefficient of the R type digital filter is calculated. As a result, when the arrival directions of the desired wave and the interference wave are assumed, a FIR digital type for a digital beam forming apparatus that forms a beam in the arrival direction of the desired wave and forms a null in the arrival direction of the interference wave. The filter coefficient of the filter can be calculated quickly and easily.

An FIR type digital filter for a digital beam forming apparatus according to claim 6 of the present invention comprises:
Since the filter coefficient is calculated and set by the filter coefficient calculating device according to claim 1, it is used for a digital beam forming device used for information transmission using a carrier wave which is a radio wave. You can

According to a seventh aspect of the present invention, there is provided a digital beam forming apparatus, wherein the array antenna, the converting means, and the plurality of digital signals output from the converting means are filtered based on a plurality of filter coefficients. A plurality of FIR digital filters for wave output and the FI
A DB including an adder for adding and outputting a plurality of filtered digital signals output from the R-type digital filter
An F circuit, wherein the plurality of filter coefficients of the plurality of FIR digital filters are one of claims 1 to 5.
Since it is calculated by the filter coefficient calculating device described in No. 3, it is possible to form a wide band beam in a desired direction and output a received signal which is a wide band desired signal corresponding to the beam.

Since the digital beam forming apparatus according to claim 8 of the present invention comprises the plurality of beam forming means and the signal selecting means, it forms a wide band beam to be calculated by the coefficient controlling means. The load factor for doing so can be reduced compared to the conventional example. As a result, the structure of the coefficient control means can be simplified.

The digital beam forming apparatus according to claim 9 is the digital beam forming apparatus according to claim 8, wherein the plurality of (N × B) FIR digital filters have a plurality of filter coefficients. Since it is calculated by the filter coefficient calculation device described in any one of 1 to 5, the filter coefficient of the FIR type digital filter can be calculated at high speed and easily.

According to a tenth aspect of the present invention, there is provided a digital beam forming apparatus comprising: beam forming means for outputting a plurality of B first beam reception signals corresponding to the first beam; and a plurality of (L × B) beam forming means. The FIR type digital filter, the plurality of B second adders, and the signal selecting means are provided, so that the calculation amount of the filter coefficient is further increased as compared with the digital beam forming apparatus according to claim 8. Can be reduced.

The digital beam forming apparatus according to claim 11 is the digital beam forming apparatus according to claim 10, wherein the plurality of (L × B) FIR digital filters have a plurality of filter coefficients. Alternatively, since it is calculated by the filter coefficient calculation device described in 2, the filter coefficient of the FIR type digital filter can be calculated at high speed.

Further, the digital beam forming apparatus according to claim 12 is the digital beam forming apparatus according to claim 8, 9, 10 or 11, wherein the coefficient control means is
Since each of the weighting factors is calculated so that the envelope of the signal output from the digital beam forming device is kept constant, the FM signal and the FSK signal in which the envelope of the transmission signal transmitted from the transmitter is constant, When receiving a PSK signal or the like, the interference signal can be suppressed and the desired signal can be extracted without requiring prior knowledge about the incident direction of the desired signal.

[Brief description of drawings]

FIG. 1 is a block diagram illustrating a configuration of a digital beam forming apparatus according to a first embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration of an FIR digital filter of the digital beam forming apparatus of FIG.

3A is a graph showing a beam in a desired direction on a frequency-angle plane, for explaining the operation principle of the DBF circuit 6 in FIG. 1, and FIG. It is a graph which shows the desired passband when the beam of the desired direction in an angle plane is converted into a two-dimensional frequency plane.

FIG. 4A is a graph showing the frequency response P (F ₂ ) of the FIR digital filter represented by Equation 8 on a two-dimensional frequency plane, and FIG. 4B is a graph showing the frequency shown in FIG. It is a graph in which the response P (F ₂ ) is rotated by a predetermined angle on the two-dimensional frequency plane, and (c) shows the rotated frequency response P (F ₂ ) shown in (b). FIG. 4 is a graph when the above-mentioned parallel movement is made to match the desired pass band shown in FIG.

FIG. 5 is a two-dimensional spectrum Sp on a two-dimensional frequency plane.
It is the figure which showed typically.

FIG. 6 is a block diagram of an adaptive digital beam forming apparatus according to a second embodiment of the present invention.

7 is a graph showing a radiation pattern of a radiation beam emitted by the adaptive digital beam forming apparatus of FIG.

FIG. 8 is a block diagram of an adaptive digital beam forming apparatus according to a third embodiment of the present invention.

FIG. 9 is a block diagram showing a configuration of an adaptive digital beam forming apparatus of a first conventional example.

FIG. 10 is a block diagram showing a configuration of an adaptive digital beam forming apparatus of a second conventional example.

11 is a block diagram showing a configuration of a second multi-beam former 7b of FIG.

FIG. 12 is a graph when Expression 6 is expressed on a two-dimensional frequency plane using the one-dimensional frequency F as a parameter in the fourth embodiment of the present invention.

FIG. 13A is a DBF in the fourth embodiment.
6 is a graph showing a beam in a desired direction and a region in which a side lobe level should be lowered in a frequency-angle plane, for explaining the operating principle of the circuit 6, and FIG. 7 is a graph showing a desired pass band and a region where the side lobe level should be lowered when the beam in the direction and the region where the side lobe level should be lowered are converted into a two-dimensional frequency plane.

FIG. 14 shows an incident angle θ = 56 in the fourth embodiment.
9 is a graph showing directional characteristics when a broadband beam is formed in the direction of °.

FIG. 15 shows an incident angle θ = 60 in the fourth embodiment.
A wide-angle beam is formed in the direction of °, and the incident angle θ is -50.
It is a graph which shows the directional characteristic when the side lobe level in the direction of ° to -20 ° is lowered.

FIG. 16 In the fourth embodiment, the incident angle θ of the desired wave is set to 25 °, and the incident angle θ of the interference wave is set to 35 ° and −50.
9 is a graph showing the directional characteristics when the weighting factor is calculated so that the interference signal is suppressed, where ° is set.

[Explanation of symbols]

1-1 to 1-N ... Antenna element, 2-1 to 2-N ... Receiver, 3-1 to 3-N ... A / D converter, 4-1 to 4-N ... Mixer, 5-1 to 5-N ... Low-pass digital filter, 6 ... DBF circuit, 7a ... First multi-beam former, 7b ... Second multi-beam former, 7c ... Third multi-beam former, 7-1-1 To 7-BN, 9-1 to 9-N, 7c-
1-1 to 7c-B-2 ... FIR type digital filter, 8 ... Signal selector, 10, 10-1 to 10-B, 93-1 to 93- (q
-1) ... Adder, 11a ... First coefficient controller, 11b ... Second coefficient controller, 12-1 to 12-M, 92-1 to 92-q ... Multiplier, 13 ... FIR digital filter Coefficient calculator, 60 ... Digital local oscillator, 91-1 to 91- (q-1) ... Delay device, 100 ... Array antenna.

Continuation of front page (51) Int.Cl. ⁶ Identification number Office reference number FI Technical indication location H01Q 25/00 H01Q 25/00 H03H 17/02 601 9274-5J H03H 17/02 601C (72) Inventor Abdeseram CLOUCHE JEDY 5 Sanrai-ya, Seiya-cho, Soraku-gun, Kyoto Pref. 5 Hiratani, A.R. Optical Communications Research Institute, Inc. (72) Ryu Miura 5 Sanrai-tani, Seiraku-cho, Seika-cho, Kyoto Pref. ATR Optical Communication Research Institute (72) Inventor Yoshio Karasawa 5 Seiraya, Seika-cho, Soraku-gun, Kyoto Pref. 5 Hiratani, ATR Optical Communication Research Institute, Inc.

Claims (12)

[Claims] 1. A linear array antenna in which a plurality of antenna elements are juxtaposed on one straight line at predetermined element intervals,
Provided corresponding to the plurality of antenna elements,
A filter coefficient calculation device for a plurality of FIR digital filters for forming a beam in a direction of a desired incident angle, the normalization time frequency being normalized by a sampling frequency of the digital beam forming device. In the two-dimensional frequency plane formed by the first axis of F _{1 and} the second axis of the normalized spatial frequency F ₂ that is normalized by the reciprocal of the element spacing, the center is located on the second axis. Then, a coefficient calculation for calculating the filter coefficient of the plurality of FIR digital filters having a predetermined amplitude in a desired pass region having a predetermined width and extending from the negative normalized time frequency to the positive normalized time frequency An apparatus for calculating a filter coefficient of an FIR type digital filter for a digital beam forming apparatus, characterized by comprising: 2. The filter coefficient calculation device for a FIR digital filter according to claim 1, wherein the coefficient calculation means converts the frequency response of a predetermined one-dimensional FIR digital filter by a predetermined variable to obtain the desired pass. The frequency response corresponding to the region and related to the normalized time frequency F ₁ and the normalized spatial frequency F ₂ is calculated, and the impulse response is calculated by performing an inverse Fourier transform on the calculated frequency response. A filter coefficient calculation device for a FIR digital filter for a digital beam forming device, which calculates a filter coefficient of the plurality of FIR digital filters having an amplitude in the desired pass region based on an impulse response. 3. The filter coefficient calculation device for an FIR digital filter according to claim 1, wherein the coefficient calculation means is in the two-dimensional frequency plane.
Side lobes having a predetermined amplitude in the desired pass region, a center on the second axis, a predetermined width, and extending from a negative normalized time frequency to a positive normalized time frequency. A filter coefficient calculation device for a FIR digital filter for a digital beam forming device, which calculates filter coefficients of the plurality of FIR digital filters having a predetermined amount of attenuation in a region where the level should be lowered. 4. The filter coefficient calculation device for a FIR digital filter according to claim 2, wherein the coefficient calculation means rotates the pass region of the one-dimensional FIR digital filter on the two-dimensional frequency plane, A frequency response regarding the normalized time frequency F ₁ and the normalized spatial frequency F ₂ corresponding to the desired pass region is obtained by performing variable conversion by translating the rotated pass region in parallel on the two-dimensional frequency plane. To calculate an impulse response by performing an inverse Fourier transform on the calculated frequency response, and based on the calculated impulse response, the plurality of FIs having an amplitude in the desired pass region.
FIR for digital beam former characterized by calculating filter coefficient of R type digital filter
-Type digital filter filter coefficient calculator. 5. The filter coefficient calculation device for an FIR digital filter according to claim 2, wherein the coefficient calculation means is configured so that the beam pattern formed by the digital beam forming device does not change with respect to frequency. By performing variable conversion of the frequency response of the one-dimensional FIR digital filter, the frequency response corresponding to the desired pass region and regarding the normalized time frequency F ₁ and the normalized spatial frequency F ₂ is calculated, Calculating an impulse response by performing an inverse Fourier transform on the obtained frequency response, and calculating filter coefficients of the plurality of FIR type digital filters having a predetermined amplitude in the desired pass region based on the calculated impulse response. FIR digital filter for a characteristic digital beam former Filter coefficient calculation device. 6. F according to one of claims 1 to 5.
An FIR digital filter for a digital beam forming apparatus, wherein the filter coefficients of the plurality of FIR digital filters calculated by the filter coefficient calculation device of the IR digital filter are set. 7. An array antenna comprising a plurality of antenna elements, and conversion means for A / D converting each reception signal received by each antenna element and outputting each digital signal corresponding to each reception signal. A plurality of FIR type digital filters for filtering and outputting the plurality of digital signals output from the converting means based on a plurality of filter coefficients, and a plurality of output types from the plurality of FIR type digital filters. A DBF circuit including an adder for adding the filtered digital signals and outputting the added signal as an output signal, wherein a plurality of filter coefficients of the plurality of FIR digital filters are provided. 5. A digital beam forming apparatus which is calculated by the filter coefficient calculating apparatus according to any one of 5 above. 8. An array antenna composed of a plurality of N antenna elements and A / D-converted each received signal received by each antenna element to output each digital signal corresponding to each received signal. A plurality of N FIR type digital filters correspond to the conversion means and the antenna elements, and a plurality N of FIR type digital filters correspond to each of the plurality of B beams to be formed. A plurality of (N × B) FIR type digital filters which are respectively provided to filter each digital signal according to a predetermined filter coefficient and output the same, and a plurality N to form each beam.
N output from each FIR digital filter
A plurality of B adders that add and output a plurality of signals, form beams in a plurality of B different directions based on the respective digital signals output from the conversion means, and form the beams into the beams. Beam forming means for outputting a corresponding plurality of B beam reception signals, and signal selecting means for selecting and outputting a plurality of M beam reception signals from the plurality of B beam reception signals output from the beam forming means A plurality of M multipliers for respectively multiplying the plurality M of beam reception signals input from the signal selection means and a plurality of M weight coefficients input, and outputting a signal of a multiplication result; Arrival of the desired signal from the main beams of the array antenna in a predetermined frequency range including at least the frequency of the desired signal based on the plurality of M beam reception signals output from the selection means. A plurality of M weighting factors for directing the received signal in the direction of arrival of the interference signal to make the level of the received signal zero are calculated for each of the multipliers, and the plurality of M weighting factors are respectively corresponded. And a coefficient control unit for outputting to each of the multipliers, and an adding unit for adding a plurality of M multiplication result signals output from the plurality of M multipliers and outputting as a reception signal. Digital beam former. 9. The digital beam forming apparatus according to claim 8, wherein the plurality of (N × B) FIR digital filters have a plurality of filter coefficients. A digital beam forming apparatus characterized by being calculated by a coefficient calculating device. 10. An array antenna composed of a plurality of N antenna elements, and A / D-converts each reception signal received by each antenna element, and outputs each digital signal corresponding to each reception signal. A plurality of N weighting factors correspond to the converting means and a plurality of N weighting factors for each of the antenna elements, and a plurality of N weighting factors for each of the first beams of the B first beams to be formed. A plurality (N × B) which are provided so as to correspond to each other, and which multiply the respective digital signals by a predetermined first weighting factor and output the result.
Number of weighting factors and a plurality of N numbering factors output from the plurality of N numbering factors for forming each of the first beams.
A plurality of B first adders for adding and outputting a plurality of signals, and forming a first beam in each of a plurality of B different directions based on each digital signal output from the conversion means. Beam forming means for outputting a plurality B of first beam reception signals corresponding to the first beam, and a plurality of L FIRs for each of the first beam reception signals.
Type B digital filter is compatible and should be formed
A plurality of L FIR type digital filters are provided so as to correspond to the respective second beams of the second beams, and each of the first beam reception signals is supplied to the plurality of B second beams. A plurality of (L × B) FIR digital filters that are filtered with a plurality of predetermined filter coefficients so as to form a beam and output, and a plurality of L number of L-type digital filters for forming each of the second beams. A plurality of L signals output from the FIR type digital filter are added to each second beam corresponding to each second beam.
B second adders for outputting the beam reception signals, and M second beam reception signals from the B second beam reception signals output from the B adders. And a plurality of M second beam reception signals input from the signal selection unit and a plurality of M second weighting factors input and multiplied. Multiple M outputs the resulting signal
Number of multipliers and a plurality of M second beam reception signals output from the signal selecting means, the main beam of the array antenna is set to a desired signal in a predetermined frequency range including at least the frequency of the desired signal. Of the plurality of Ms so that the level of the received signal in the arrival direction of the interference signal and in the arrival direction of the interference signal is zero.
The second weighting factors are calculated for each of the multipliers,
Coefficient control means for outputting the plurality of M second load coefficients to the corresponding multipliers, and a plurality of M multiplication result signals output from the plurality M of multipliers are added and received. A digital beam forming apparatus comprising: an addition unit that outputs a signal. 11. The digital beam forming apparatus according to claim 10, wherein the plurality of (L × B) FIR digital filters have a plurality of filter coefficients. A digital beam forming apparatus characterized by being calculated by a coefficient calculating device. 12. The coefficient control means calculates each of the weighting factors so that the envelope of a signal output from the digital beam forming device is kept constant. Alternatively, the digital beam forming apparatus according to item 11.