JP3107781B2 - Array antenna control method and control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method and apparatus for controlling an array antenna for a wireless communication system having a common transmission / reception array antenna composed of a plurality of antenna elements and having different transmission / reception frequencies.

[0002]

2. Description of the Related Art With the explosion of mobile communication systems,
The radio wave environment is becoming more severe. Since the adaptive array antenna directs the directivity zero to the direction of arrival of the interference wave, it is effective in reducing frequency-selective fading and removing co-channel interference. In a mobile communication system, it is important not only to eliminate interference from other stations at the time of reception, but also to perform transmission so as not to interfere with other communication systems. For example, it is known that the number of zone repetitions can be reduced if an antenna pattern can be formed so that a zero point is adaptively oriented in the direction of an interference wave in both transmission and reception in a cellular base station (for example, see Prior Art Document 1 “Daigane,“ Cellular ”). Improvement of Frequency Utilization Efficiency by Controlling Antenna Directivity of Base Station ", Proc. Of the IEICE Spring 1993 National Convention, B-397, March 1993.)

In order to form a pattern adaptively at the time of transmission, a weight coefficient at the time of reception is required. However,
When the frequency is different between reception and transmission as in a frequency division multiplexing system, if the weighting factor at the time of reception is used as it is at the time of transmission, the zero point direction is shifted, and the transmission characteristic deteriorates (for example, Prior Art Document 2 "Takatoshi Aoyagi et al.," Characteristics of LMS Adaptive Array Antenna During Transmission ", IEICE Technical Report, A.P86
-95, 1986 ". ).

[0004]

One method for solving such a problem is disclosed in prior art document 3 “I. Chiba et al.,“ T.
ransmitting null beam forming with beam space adap
tive array antennas ", Proceedings of IEEE Vehicular
Technology Conference, Stockholm, Sweden, pp. 1498-1502, June 1994 ". Here, a simple transmission load coefficient calculation method utilizing the configuration of the beam space type adaptive array antenna is proposed. However, in the method of the prior art document 3, in a multi-beam pattern at a transmission frequency used for obtaining a load at the time of transmission, the main beam direction is the same as that at the time of reception, but the directivity characteristics other than the main beam are different from those at the time of reception. In some cases, a zero point is not formed in the direction of the interference wave because the characteristic has a compressed or expanded form.

The present inventors have proposed a beam space type adaptive array (hereinafter, referred to as a conventional example) corresponding to a propagation wave signal having a wide fractional bandwidth, for example, as disclosed in Japanese Unexamined Patent Publication No.
No. 261008. In the multi-beam forming unit, a fixed direction broad band multi-beam having a beam pattern approximately independent of frequency is formed,
The following adaptive processing unit is simplified. The conventional example is characterized by fast convergence. By using a wide-band multi-beamforming circuit that has the characteristic that the beam pattern does not depend on frequency, even if the frequency differs between reception and transmission (the signal has a narrow bandwidth), almost the same beam pattern is formed during transmission and reception. It will be possible to do so. However, the conventional example is a processing system for a signal with a wide relative bandwidth, and the digital multi-beam forming circuit requires a large amount of signal processing operation with a tapped delay line circuit configuration, and requires a high sampling frequency and high-speed processing. On the other hand, although the signals targeted in the present invention are separated in frequency between transmission and reception, each signal itself is a signal having a narrow relative bandwidth. Therefore, there is a problem that it is wasteful to use the configuration of the conventional example as it is.

SUMMARY OF THE INVENTION An object of the present invention is to solve the above-mentioned problems, and in a wireless communication system having a common transmitting and receiving array antenna and having different transmitting and receiving frequencies, a main beam is directed in a desired wave direction for both transmitting and receiving, and is always in an interference wave direction. An object of the present invention is to provide a method and a device for controlling an array antenna, which can be controlled so as to direct a zero point and which has a simple circuit configuration as compared with the conventional example.

[0007]

According to a first aspect of the present invention, there is provided a method for controlling an array antenna, comprising three or more odd-numbered N antenna elements which are linearly arranged at regular intervals at predetermined intervals. A shared transmitting / receiving array antenna, receiving means for receiving the odd N received signals received by the odd N antenna elements of the transmitting / receiving array antenna, and receiving the odd N received signals received by the receiving means. Receiving multi-beam forming circuit for forming predetermined M multi-beam signals by multiplying each of the multi-beam signals by a predetermined receiving weighting coefficient; and M multi-beam forming circuits formed by the receiving multi-beam forming circuit. Select and output I multi-beam signals having higher power or higher than a predetermined threshold power among the signals Multiplying means for multiplying each of the I multi-beam signals selected by the selecting means with a predetermined adaptive control weighting factor and adding the multiplication results to output a signal of the addition result as a reception signal Addition means, distribution and multiplication means for dividing the input transmission signal into odd N transmission signals and multiplying by a predetermined transmission weight coefficient to output odd N transmission signals, and output from the multiplication and addition means A predetermined adaptive control algorithm is used to direct the main beam of the array antenna in the direction of arrival of the desired wave and to reduce the level of the received signal in the direction of arrival of the interference wave to zero using a predetermined adaptive control algorithm. Coefficient calculating means for calculating adaptive control load coefficients respectively corresponding to the plurality of multi-beam signals and setting them in the multiplying and adding means; Transmitting means for transmitting by amplifying a transmission signal and outputting the amplified signal to the odd number N of antenna elements, wherein the control means for an array antenna for a wireless communication system having mutually different transmission / reception frequencies is provided. Utilizing coefficients of a predetermined broadband digital multi-beam forming circuit having a frequency-independent beam pattern having a beam forming direction, based on a receiving frequency and a transmitting frequency, a weight coefficient of the receiving multi-beam forming circuit and a transmitting coefficient are determined. A first step of calculating and setting a multi-beam forming weight coefficient with respect to a frequency; a multi-beam forming weight coefficient with respect to a transmitting multi-beam forming circuit and a transmitting frequency calculated by the first step; Based on the adaptive control weighting factor calculated by the means. And a second step of calculating and setting the transmission weight coefficient of the distribution multiplying means so as to form a main beam in the direction of the desired wave and to form a zero point in the direction of arrival of the interference wave together with the received signal. It is characterized by the following.

According to a second aspect of the present invention, there is provided a control apparatus for an array antenna, comprising: a shared transmission / reception array comprising three or more odd-numbered N antenna elements which are linearly arranged at regular intervals at predetermined intervals. An antenna; receiving means for receiving the odd N received signals received by the odd N antenna elements of the transmission / reception array antenna; and a multi-beam for each of the odd N received signals received by the receiving means. A receiving multi-beam forming circuit that forms a predetermined M multi-beam signals by multiplying a predetermined receiving weighting coefficient for each signal, and among the M multi-beam signals formed by the receiving multi-beam forming circuit, Selecting means for selecting and outputting I multi-beam signals having greater power or having power greater than a predetermined threshold power; I selected by said selection means
Multiplying and adding means for multiplying each of the multi-beam signals by a predetermined adaptive control weighting factor and adding the multiplication result to output a signal of the addition result as a reception signal; Distribution multiplication means for dividing the transmission signals into a plurality of transmission signals and multiplying the transmission signal by a predetermined transmission weight coefficient to output odd N transmission signals; and a predetermined adaptive control algorithm based on the reception signals output from the multiplication and addition means. For directing the main beam of the array antenna in the direction of arrival of the desired wave and making the level of the received signal in the direction of arrival of the interference wave zero, for adaptive control respectively corresponding to the I multi-beam signals. A coefficient calculating means for calculating a weighting coefficient to set the multiplying and adding means, and an odd N transmitting signal output from the distribution and multiplying means for amplifying the odd N transmitting elements. The control apparatus for the array antenna for the transmission means and a transmission and reception frequency are different from each other a wireless communication system that transmits by outputting to,
Utilizing the coefficients of a predetermined broadband digital multi-beam forming circuit having a frequency-independent beam pattern having a predetermined multi-beam forming direction, a receiving frequency,
First calculating means for calculating and setting a weighting factor of the receiving multi-beam forming circuit and a multi-beam forming weighting factor for the transmitting frequency based on the transmitting frequency; and a receiving multi-factor calculated by the first calculating means. Based on the beam forming circuit load coefficient and the multi-beam forming load coefficient for the transmission frequency, and the adaptive control load coefficient calculated by the coefficient calculating means, both the transmission signal and the reception signal form a main beam in a desired wave direction. And a second calculating means for calculating and setting the transmission load coefficient of the distribution multiplying means so as to form a zero point in the direction of arrival of the interference wave.

[0009]

[0010]

[0011]

Embodiments of the present invention will be described below with reference to the drawings.

FIG. 1 is a block diagram showing the configuration of a radio transmission / reception system including an array antenna control device according to an embodiment of the present invention. In the embodiment according to the present invention,
A wideband digital multi-beam forming circuit (hereinafter referred to as a digital beam forming circuit) having a frequency-independent Is referred to as a DBF circuit) for transmission / reception adaptive beamforming. It is noted that for a signal with a narrow bandwidth used in the present embodiment, it is not necessary to use a tapped delay line circuit as a DBF circuit, and it is sufficient to consider only transmission and reception frequency points. The receiving side is an adaptive array having a beam space configuration, and the weighting factor of the DBF circuit is calculated from the weighting factor of a virtual broadband DBF circuit 9 having a frequency-independent beam pattern. The filter coefficients of the virtual wideband DBF circuit 9 can be calculated by a conventional method. The tapped delay line circuit in the virtual broadband DBF circuit 9 can be regarded as a load coefficient that changes according to the frequency. Since the element interval and the frequency of the transmission / reception signal are known, the weight coefficient of the DBF circuit 9 at that frequency can be obtained in advance. The weighting factor on the transmitting side can be obtained by calculating a simple matrix product from the adaptive weighting factor on the receiving side and the weighting factor of the DBF circuit 5 at the previously calculated transmission frequency. Since the transmission and reception have the same multi-beam pattern, the transmission and reception can have almost the same zero depth. Therefore, the problem of the conventional method (Prior Art Document 3) that a zero point may not be formed in the direction of an interference wave during transmission is solved. Note that the virtual wideband DBF circuit 9 does not process the signal, but only uses it to calculate the weighting factor of the DBF circuit for the transmission / reception frequency.

In this specification, where n = 1, 2, 2,
, N, m = 1, 2,..., M, i = 1,
2, ..., I, and the same applies hereinafter unless otherwise specified in this specification. The digital in-phase / quadrature signal x
k in parenthesis () in _n (k) is time (time number)
Represents Furthermore, in this specification, an interference wave is used as an unnecessary wave in a broad sense.

In FIG. 1, an array antenna 100 is
The number N of antenna elements is 1 at a predetermined element interval d.
This is an evenly spaced linear array antenna formed side by side on a straight line. The number N of elements is set to an odd number due to the restriction of the characteristic approximation method of the broadband DBF circuit having the frequency-independent beam pattern (for details, see Japanese Patent Application Laid-Open No. 9-261008). As shown in FIG. 1, the signal incident direction θ is defined by the angle between the perpendicular of the line where the antenna elements 1-1 to 1-N are juxtaposed and the direction in which the desired signal is incident. Let the reception frequency be f _RX . The receiving side has a beam space configuration.

First, the configuration of FIG. 1 will be described. After the transmission baseband signal is input to the in-phase distributor 3, the transmission baseband signals distributed to a plurality of N and distributed in-phase are weight coefficients w _{TX1 to} w _TXN calculated by the transmission weight coefficient calculator 32, respectively. Multipliers 4-1 to 4-
N, and the D / A converter 2 of the transmission / reception devices 2-1 to 2-N
5 to the transmitter 26. The transmitter 26 includes a digital modulator and a transmission power amplifier, and performs digital modulation such as PSK modulation on a carrier signal having a predetermined radio transmission frequency f _TX in accordance with the input two rows of transmission baseband signals. Antenna elements 1-1 to 1- of array antenna 100 via circulator 20
N, and radiates as electromagnetic waves from antenna elements 1-1 to 1-N toward free space at an angle θ.

On the other hand, the received signals received by the antenna elements 1-1 to 1-N are transmitted and received by the transmitting and receiving devices 2-1 to 2-N.
The signal is input to each receiver 21 via each of the N circulators 20. Each receiver 21 includes a low-noise amplifier, an intermediate frequency converter for converting a radio reception frequency f _RX to an intermediate frequency f _IF , an intermediate frequency amplifier, and the like. / D converter 22. Next, the A / D converter 22 converts the input analog reception intermediate frequency signal into a digital reception intermediate frequency signal,
The IQ signal converter 23 outputs the received intermediate frequency signal to the signal converter 23. The IQ signal converter 23 converts the input received intermediate frequency signal into a complex reception baseband signal x _n composed of two signals, in-phase and quadrature, which are orthogonal to each other.
(K) and output to the DBF circuit 5. Furthermore, D
The BF circuit 5 forms a multi-beam forming circuit, and receives N input baseband signals x _n (k)
A plurality of M multi-beams are formed by multiplying using the load coefficient matrix R calculated by the multi-beam load coefficient calculator 31, and the M multi-beam signals y _m (k) resulting from the multiplication are used as beam selectors. 6 is output. The beam selector 6 receives the M multi-beam signals y
_m (k) with higher power (where I ≦ M
It is. ) Are selected, and the selected I received signals y _ri (k) are multiplied by multipliers 7-1 to 7 having reception weight coefficients w _{RX1 to} w _RXI calculated by the CMA coefficient calculator 33, respectively. and outputs to the adder 8 and CMA coefficient calculator 33 through the -I, outputs the selected beam number m of multi-beam signal y _m selected (k) to transmit the load coefficient multiplier 32. The adder 8 adds the input I received baseband signals, outputs one received baseband signal z (k) to an external device, and outputs the received baseband signal z (k) to the CMA coefficient calculator 33. In FIG. 1, in-phase and quadrature signals are obtained after A / D conversion of the received intermediate frequency signal, but the order of these two processes may be reversed.

Here, the beam selector 6 receives the input M
Although I received signals having higher power are selected from the multi-beam signals y _m (k), the present invention is not limited to this, and I received signals having power equal to or higher than a predetermined threshold value are selected. A signal may be received (variation example). In the present embodiment, by providing the beam selector 6, the control of the multi-beam formation of the present embodiment can be reliably performed by using the received signal of higher power. Further, the beam selector 6 may not be provided.

The DBF circuit 5 on the receiving side calculates not the FFT / DFT but the coefficients of the virtual broadband DBF circuit 9 of FIG. 2 having a frequency-independent beam pattern. Also, C
The MA coefficient calculator 33 is, for example, a conventional constant modulus algorithm (hereinafter, referred to as a CM algorithm or CMA; for example, specifically, Takeo Ohgane et al., “Selective fading compensation characteristic of CMA adaptive array in land mobile communication”, Electronic Information Communication) Journal, Vo
l. J73-B-II, No. 10, pp 489-497
reference. ), The received baseband signal z (k) output from the adder 8 and the output signal y of the beam selector 6
Based on _ri (k), the main beam of the array antenna 100 is directed to a desired direction of a desired wave, and a plurality of reception signals corresponding to the respective beams are set so that the reception level in the arrival direction of an interference wave such as an interference wave becomes zero. The I reception weight coefficients w _RXi are calculated as follows. That is, as described below, in a communication system using a signal wave of a desired wave having a constant envelope, the CM algorithm converts the waveform of the envelope changed by the influence of an interference wave such as an interference wave into a desired shape. The reception level in the radiation pattern of the array antenna 100 in the direction of arrival of the interference wave is reduced to zero by the conversion into.

Now, r selected by the beam selector 6
The received signal at the time k of the ith beam is represented by y _ri (k)
Then, the complex number _weighting factor to be applied to the received signal y _ri (k) is defined as w _RXi (k). Here, the array antenna 1
The synthesized electric field Y synthesized using 00 can be expressed by the following equation (3).

(Equation 1) Here, assuming that the shape of the desired envelope of the signal wave is a predetermined constant value P ₀ for simplicity, a complex number weighting factor w for setting the envelope of the signal of the composite electric field to a constant value P _0.
As is well known, _RXi (k) is obtained by calculating the complex weighting coefficient w that minimizes the evaluation function F shown in the following _Expressions 2 and 3.
_This is equivalent to determining _RXi (k).

F = (| Y | ² −P ₀ ² ) ² Here, the following equation is obtained by substituting the composite electric field Y expressed by Equation 1 into Equation 2.

(Equation 3)

Therefore, the complex weight w _RXi (k) is _calculated by the following equation using the complex weight w _RXi (k) at the next time.
And the received signal y _ri (k + 1) at the next time is calculated, whereby the envelope of the signal wave can be set to a desired shape and the reception level in the radiation pattern in the arrival direction of the interference wave can be reduced to zero. .

## EQU4 ## w _RXi (k + 1) = w _RXi (k) -μy _ri (k)
* · (| Y | ² −P ₀ ² ) · Y where μ is a constant determined by the system,
y _ri (k) * is a complex conjugate of the received signal y _ri (k) represented by a complex number. When the CM algorithm is used, it is possible to form zeros in the radiation pattern by subtracting 1 from the number of beams of the multi-beam, as is well known.

As described above, the CMA coefficient calculator 33
Is based on the output signal y _ri (k) of the beam selector 6 and the received baseband signal z (k) output from the adder 8 operating as an in-phase combiner using a known CM algorithm. A plurality of I reception weight coefficients w _Rxi are calculated for the reception signals corresponding to the respective beams such that the main beam is directed in a desired direction of the desired wave and the reception level in the arrival direction of the interference wave such as the interference wave is made zero. Then, it outputs to the multipliers 7-1 to 7-I and the transmission weight coefficient calculator 32.

In the present embodiment, the CMA coefficient calculator 33 calculates a plurality of I reception weight coefficients w _Rxi for received signals using a known CM algorithm, but the present invention is not limited to this. , LMS (Least Mean Squar
es) The reception weight coefficient w _Rxi may be calculated using an adaptive algorithm such as an algorithm.

On the transmitting side, on the other hand, as will be described in detail later, the filter coefficient calculator 30 generates a broadband DBF circuit 9 (having a frequency-independent beam pattern) based on the transmission frequency f _TX and the reception frequency f _RX. The filter coefficient of the FIR digital filter A _mn (z) in FIG. 2) is calculated, and the multi-beam weighting coefficient calculator 31 calculates the calculated DB
Based on the filter coefficient of the FIR type digital filter A _mn (z) of the F circuit 9, the DB for the reception frequency f _RX
Load coefficient of F circuit 5 and DBF for transmission frequency f _TX
Calculate the weighting factor of the circuit. Transmission load coefficient calculator 32
Are the weight coefficient of the DBF circuit for the transmission frequency f _TX calculated by the multi-beam weight coefficient calculator 31 and the reception weight coefficient W _RX = calculated by the CMA coefficient calculator 33.
[W _RX1 , w _{R X2} ,..., W _RXI ] Based on ^T , W _TX = [w
_TX1, w _TX2, ..., to calculate the w _TXN] ^T. In this embodiment, the multi-beam pattern for reception and transmission is θ = ± 9
They are almost the same except for the vicinity of 0 °. Therefore, almost the same zero depth can be obtained in transmission and reception. Further, both the reception and transmission DBF circuits are of a narrow band type. That is, a simple weighting operation is performed instead of a convolution operation as in a broadband system. Although the circuit configuration of the present embodiment of FIG. 1 is similar to the conventional example, the reception weight coefficient and the transmission weight coefficient of the reception-side DBF circuit are based on the filter coefficient of the broadband DBF circuit 9 having a frequency-independent beam pattern. It is different in that it is calculated.

Next, a method of calculating the weight coefficient of the receiving DBF circuit 5 will be described in detail. FIG. 2 is a block diagram showing the configuration of the virtual wideband DBF circuit 9. FIG. 3 is a virtual finite impulse response type (hereinafter, the finite impulse response type is referred to as an FIR type) which is the delay line circuit with taps in FIG. FIG. 3 is a block diagram showing the configuration of a digital filter 11-mn, where each virtual FIR digital filter 11-m is shown.
−n is a transfer function A _mn (z) (n = 1, 2,..., N; m =
1, 2,..., M). In the virtual FIR digital filter 11- _{mn of} FIG. 3, the number of taps is K, and the coefficients of the multipliers 92-1 through 92-K in the tapped delay line circuit having the transfer function A _mn (z) are a _{m, n, 0} , am _{, n, 1} , ..., am _{, n, K-1} . Also,
z ^-1 is a delay unit for one sampling interval when a signal is processed by the wideband DBF circuit 9 in FIG. It should be noted that the circuits of FIGS. 2 and 3 are not used for actual signal processing in the present invention. All that is required are the coefficients am _{, n, 0} , am _{, n, 1} , ..., am _{, n, K-1} . The circuits of FIGS. 2 and 3 are only used for calculating the weighting factors of both the receiving and transmitting DBF circuits of FIG. In addition, the following FIG.
In the description of the configuration of FIG. 3 and FIG. 3, it is assumed that signal processing is performed for the sake of convenience contrary to the above points.

The virtual DBF circuit 9 in FIG. 2 is a kind of spatial filter bank including FIR digital filters 11-1-1 to 11-MN and adders 10-1 to 10-M. . Here, the digital phase of the received signal
The quadrature signal x ₁ (k) is output from the FIR digital filter 1
1-1-1, 11-2-1,..., 11-M-1, and the digital in-phase / quadrature signal x ₂ (k) of the received signal
Are FIR digital filters 11-1-2, 11-
2-2,..., 11-M-2, and similarly, the digital in-phase / quadrature signal x _n (k) of the received signal is
R type digital filters 11-1-n, 11-2-n,
.., 11-Mn (n = 3, 4,..., N). Next, each FIR type digital filter 11-m-
n is the digital in-phase of the input received signal.
The orthogonal signal x _n (k) is digitally filtered with a predetermined filter coefficient described later in detail, and output to the adder 10-m.
The adder 10-m adds the filtered digital in-phase / quadrature signals x _m (k) and outputs the added signal y _m (k).

Each of the virtual FIR digital filters 11-mn shown in FIG. 3 is a so-called transversal filter, and is configured as follows. As shown in FIG. 3, the FIR digital filter 11-mn has:
(K-1) delay units 91-1 to 91- (K-1)
And K multipliers 92-1 to 92-K, and (K-1)
It comprises the adders 93-1 to 93- (K-1).
Here, K is called a tap length in the FIR digital filter 11-mn, and is set to an odd number in the present embodiment. Then, the FIR digital filter 1
1-mn digital in-phase / quadrature signal x _n
(M) is input to the delay unit 91-1 and the multiplier 92-1. The FIR digital filter 11-mn
Load factor is input to a _{m, n, s-1} to the multiplier 92-s (s
= 1, 2,..., K). In the FIR digital filter 11-mn, the delay unit 91-s (s =
1, 2,..., K-1) are input digital in-phase
The orthogonal signal x _n (k-s + 1) is delayed by one sample period, and the signal x _n (k-s) delayed by one sample period is output to the delay unit 91- (s + 1) and the multiplier 92- (s + 1). . The multiplier 92-1 multiplies the input digital in-phase / quadrature signal x _n (k) by a weighting coefficient am _{, n, 0} ,
Output to adder 93-1. Multiplier 92-s (s = 2,
,..., K) are multiplied by the input signal x _n (k−s + 1) and the weighting factors am _{, n, s−1} to form an adder 93− (s−).
Output to 1). Adder 93-1 adds the signal input from multiplier 92-1 and the signal input from multiplier 92-2, and outputs the result to adder 93-2. Adder 93
−s (s = 2, 3,..., K−2) is added to the adder 93− (s
-1) and the signal input from the multiplier 92- (s + 1) are added to form an adder 93- (s +
Output to 1). The adder 93- (K-1)
3- Add the signal input from (K-2) and the signal input from multiplier 92-K and output. The FIR digital filter 11-mn constructed as described above.
Outputs digitally filtered digital in-phase / quadrature signal x _n (k) with a predetermined transfer function A _mn (z).

The filter coefficient calculator 30 shown in FIG. 1 calculates the weight coefficients am _{, n, 0} , am _{, n, 1} ,..., _{Am, n} of the virtual FIR digital filter 11-mn shown in FIG. _{, K-1} are disclosed in
The calculation is performed according to a known calculation method disclosed in Japanese Patent Application Publication No. 61008. This calculation method will be described in detail in the last part of the detailed description of the invention. The parameter required at that time is the number N of antenna elements of the array antenna 100.
2, the element spacing d, the angles θ of the plurality of M beam forming directions, the number of beams M, the carrier frequency f _c0 and the sampling frequency f _s0 when the signal is processed by the virtual broadband DBF circuit 9 in FIG. It is a number K. Here, f _c0 and f _s0 are the load coefficients a _{m, n, 0} , a _{m, n, 1} ,.
_{It is} only used when calculating _{m, n, and K-1} and is a virtual one that has nothing to do with the sampling frequency in the signal processing of FIG. The setting of the carrier frequency f _c0 is, for example, f _c0 is f
_It is assumed to be between _RX and _fTX . That is, in the preferred embodiment, f _c0 = (f _RX + f _TX ) / 2. From the viewpoint of the degree of approximation of the characteristics of the virtual broadband DBF circuit 9 in FIG. 2, the sampling frequency f _s0 is preferably about twice as large as | f _RX −f _TX |. The coincidence of the transmission and reception beam patterns greatly depends on the degree of approximation of the characteristics of the virtual broadband DBF circuit 9. Furthermore, the distance d between the antenna elements about half the wavelength corresponding to the frequency (f _c0 + 0.5f _s0) is desirable (for details, JP-A 9-261
See No. 008. ). Since the virtual wideband DBF circuit 9 in FIG. 2 does not actually perform signal processing, the number of taps K has no relation to the signal processing calculation amount in FIG. The standard of this value is about the same as the number of elements.

The input / output relationship of the DBF circuit 5 on the receiving side in FIG. 1 is expressed by the following equation.

## EQU5 ## Y (k) = RX (k)

X (k) = [x ₁ (k), x ₂ (k),..., X _N (k)] ^T

Y (k) = [y ₁ (k), y ₂ (k),..., Y _M (k)] ^T
Here, X (k) is an input signal vector to the DBF circuit 5 on the receiving side, Y (k) is an output signal vector from the DBF circuit 5, and R is a weighting factor of the DBF circuit 5 on the receiving side. It is a matrix. The weight coefficient matrix R is a frequency response value of the virtual wideband DBF circuit 9 in FIG. 2 at a frequency f = f _RX . Referring to FIG. 2, it is represented by the following equation.

[0029]

(Equation 8) Here, A _mn (f) means a frequency response value of A _mn (z) with respect to the frequency f. That is,

(Equation 9) It is. DBF circuit 5 on the receiving side based on Equations 8 and 9
Is calculated.

Therefore, the filter coefficient calculator 30 shown in FIG.
The FIR digital filter 11-mn is calculated using the calculation method described in the last part of the detailed description of the invention.
After calculating the coefficients am _{, n, 0} , am _{, n, 1} ,..., _{Am, n, K−1} , the coefficients are output to the multi-beam load coefficient calculator 31. In response, the multi-beam weighting factor calculator 31 calculates
Is used to calculate the weighting coefficient matrix R in the receiving-side DBF circuit 5 shown in Expression 8 and output to the receiving-side DBF circuit 5.

Next, a method of calculating the transmission weight coefficient in the multipliers 4-1 to 4-N will be described in detail. The transmission weight coefficients w _TX1 , w _TX2 ,..., W _TXN are calculated using the reception side weight coefficient w _RXi . As a preparation, first, a load coefficient equivalently given to each of the antenna elements 1-1 to 1-N on the receiving side is obtained. The beam number selected in the beam space adaptive array on the receiving side is denoted by ｛r ₁ ,
r ₂ ,..., r _I }. At this time, the relationship between the output signal Y (k) from the DBF circuit 5 and the output signal Y _S (k) from the beam selector 6 as a beam selector is expressed by the following equation.

## EQU10 ## Y _S (k) = SY (k)

Y _s (k) = [y _r1 (k), y _r2 (k),
.., Y _rI (k)] ^T where S is a beam selection matrix whose size is I
× M, and the (i, r _i ) element is 1 (i = 1, 2,...,
I), while the other elements are zero. Further, the output signal z (k) from the adder 8 after the multiplication operation of the multipliers 7-1 to 7-I is as follows.

[Number 12] ^{_{z (k) = W RX T}} Y S (k)

[ _Expression 13] W _RX = [w _RX1 , w _RX2 ,..., W _RXl ] ^T

From the above equations (5), (10) and (12), the relationship between the input signal X (k) to the DBF circuit 5 on the receiving side and the adaptive array output signal z (k) can be expressed by the following equation.

[ _Mathematical formula-see _{original document} ] z (k) = W _RX ^T SRX (k) Accordingly, the weight coefficient matrix W _RXeq given equivalently to each antenna element on the receiving side can be expressed by the following equation.

[ _Equation 15] W _RXeq = W _RX ^T SR

The transmission weighting factor is calculated by the DBF of the receiving side in Equation (16).
The weight coefficient matrix R of the circuit 5 must be replaced with that at the transmission frequency f _TX . D at transmission frequency f _TX
Assuming that the weighting coefficient matrix of the BF circuit is T, the transmission weighting coefficient matrix W _TX can be expressed by the following equation.

[Equation 16] W _TX = W _RX ^T ST

W _TX = [w _TX1 , w _TX2 ,..., W _TXN ] ^T Here, the matrix T can be represented by the following equation.

(Equation 18) Since each element of the matrix T can be calculated in advance using Equation 9, the reception weight coefficient W _RX and the beam selection matrix S
, The transmission weight coefficient matrix W _TX can be calculated adaptively. The matrix T is only used to calculate the transmission weight coefficient matrix W _TX , and the DBF circuit on the transmission side does not actually appear in the signal processing circuit.

Therefore, the multi-beam weighting factor calculator 31
Is a coefficient a of the FIR digital filter 11-mn calculated by the filter coefficient calculator 30.
_{Based on m, n, 0} , am _{, n, 1} ,..., am _{, n, K−1} , the weight coefficient matrix T of the DBF circuit at the transmission frequency f _TX is calculated using Expressions 18 and 9. Then, the calculation result is output to the transmission load coefficient calculator 32. Next, in response to this, the transmission weight coefficient calculator 32 calculates the transmission weight coefficient matrix W _TX using Expression 16, and outputs and sets the transmission weight coefficient matrix W _TX to the multipliers 4-1 to 4-N.

[0035]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS With respect to the radio transmission / reception system configured as described above, the present inventor compared directional characteristics at the time of reception and at the time of transmission by computer simulation. Here, the number of antenna elements N = 17, the reception frequency f _RX = 1.9 GHz, and the transmission frequency f _TX = 2.1 GHz. Desired wave incident angle is 1
5 ° (signal power-to-noise power ratio SNR = 20 dB), two interference waves, incident angles of −25 ° and 35 °, and both interference signal power-to-noise power ratio INR = 17 dB. None of these are QPSK signals independent of each other and their bandwidths are sufficiently smaller than the reception frequency. The distance d between the antenna elements was set to d = (0.9c) / (f _RX + f _TX ). It is assumed that each element pattern is isotropic and has no mutual coupling.

Here, the virtual broadband DBF circuit 9 of FIG. 2 having a frequency-independent beam pattern was designed by the method disclosed in Japanese Patent Application Laid-Open No. 9-261008 as described above.
The direction of the multi-beam and the number M are such that the direction of the main beam of a certain beam is approximately the zero point direction of another beam. As a result, the multi-beam number M = 11, and the multi-beam directions are 0 °, ± 10 °, ± 20 °, ± 31 °, ± 43.
゜, ± 59 °. Number of taps K = 21, virtual carrier frequency f _c0 required for design = (f _RX + f _TX ) / 2
= 2 GHz, virtual sampling frequency f _s0 = 2 | f
_RX- f _TX | = 0.2f _c0 = 0.4 GHz. FIG. 4 shows directivity characteristics of a beam in the 20 ° direction in which the power of the output signal becomes maximum at the time of reception among the transmission and reception multi-beams. Except for the vicinity of θ = ± 90 °, they are almost the same. As described above, the adaptive algorithm is a CMA (Constant Modul
us Algorithm). From the output signals from the DBF circuit 5 on the receiving side, a beam having power equal to or more than 1/10 of the maximum power was selected (this selection method is not an embodiment but a modification). As a result, the beam selection number I is 8
It became.

After convergence (after updating the load coefficient 1000 times)
FIGS. 5 to 7 show the directivity characteristics at the time of reception and the transmission directivity characteristics at the time when the transmission weight coefficient matrix W _TX is obtained by _Equation 16 using the reception weight coefficient at that time. In both transmission and reception, it can be seen that the main beam is directed in the desired wave direction and the zero point is directed in the interference wave direction (however, the zero point in the -25 ° direction is slightly shifted). The depth of the zero point is the same for transmission and reception. This has been confirmed in simulations under other conditions.

The method of the conventional example (prior art document 3) was also tried. However, since the receiving-side multibeam is an orthogonal multibeam by DFT, the element spacing is d = (0.5c).
/ F _RX . The number of multi-beams is 17. FIG. 8 shows the directional characteristics of the 14 ° direction beam in which the power of the output signal becomes maximum during reception among the transmission and reception multi-beams. Although the main beam direction is the same, the transmit directivity becomes a compressed form of the receive directivity.
It can be seen that there is no agreement except at 70 °. The main beam width is smaller than that in FIG. The beam selection method is the same as in the simulation of the present embodiment. The number of multi-beams was 17, the same as the number of elements, and 4 beams were selected from them. After convergence (1000
The following shows the directional characteristics at the time of reception (after updating the weighting factor) and the directional characteristics at the time of transmission using the receiving weighting factor at that time. The directional characteristics at the time of transmission deviate from the directional characteristics at the time of reception in any of the interference wave directions. This is probably because the transmission / reception directional characteristics in FIG. 8 do not match the interference wave direction.
However, even in the conventional method, an interference wave direction zero point may be formed in both transmission and reception. When a trial was performed with the desired wave 25 °, the interference wave 35 ° and −50 ° (two waves), a zero point was formed in the direction of the interference wave even during transmission. Looking at the directional characteristics of the 28-degree beam in which the power of the output signal at the time of reception among the transmitted and received multi-beams was maximized, the directional characteristics of transmission and reception in the interference wave direction were close.

The differences from Japanese Patent Application Laid-Open No. 6-196921 are as follows. Although the configuration of this embodiment is similar to that of the conventional example, the reception weight coefficient matrix R calculated based on the virtual broadband DBF circuit 9 of FIG. And the method of calculating the transmission weight coefficient W _TX are different. The method of the present embodiment is characterized in that the reception and transmission weighting factors are calculated based on a virtual broadband DBF circuit 9 having a frequency-independent beam pattern.

As described above in detail, according to the present embodiment, when the frequency differs between reception and transmission as in a frequency division multiplexing system in mobile communication, the transmission and reception array antennas are shared and linearly spaced at equal intervals. On the premise that the array antennas arranged are used, the present inventor has proposed a transmission / reception adaptive beam forming method using a broadband DBF circuit having a frequency-independent beam pattern proposed in JP-A-9-261008 for weight factor calculation. The algorithm is disclosed. By computer simulation, it was confirmed that the main beam was directed in the desired wave direction and the zero point was directed in the interference wave direction in both transmission and reception, and that the depth of the zero point was equivalent in transmission and reception. That is, the embodiment according to the present invention has the following unique effects. (A) Even when the transmission and reception frequencies are different, at the time of transmission, the beam can be controlled such that the main beam and the zero point are directed in the same direction as the desired signal direction and the interference signal direction at the time of reception. (B) The circuit configuration is simpler than the conventional example.

<Supplementary Explanation> Lastly, a supplementary explanation will be given on a method of calculating the weighting factors am _{, n, s} of the virtual FIR type digital filter 11-mn shown in FIG. 3 which is executed in the filter coefficient calculator 30 of FIG. explain. First, the signal incident angle θ
To the antenna elements 1-1 to 1-N as shown in FIG.
Is defined as the angle between the perpendicular of the juxtaposed lines and the direction of incidence of the desired signal. The number of normalized time frequencies F ₁ follows by using the carrier frequency f _c0 and the sampling frequency f _s0 when processing signals with DBF circuit 9 of the frequency f ₁ and 2 of the incident signal is a radio frequency signal 19 Defined by Further, using the incident angle θ, the speed of light c, and the frequency f _{1 of the} incident signal, a non-normalized spatial frequency f ₂ is defined as in Expression 20,
The normalized spatial frequency F ₂ is defined by normalizing the denormalized spatial frequency f ₂ with the reciprocal of the element interval d of the antenna elements 1-1 to 1-N as represented by Expression 21. Here, the reciprocal 1 / d of the element interval d of the antenna element is called a spatial sampling frequency. That is, the normalized time frequency F ₁
Is the frequency obtained by normalizing the difference between the frequency f ₁ and the carrier frequency _{fc 0 at} a certain time by the sampling frequency f _s0 , and the normalized spatial frequency F ₂ is the antenna element 1-n at a certain time considering the incident angle θ. Is the spatial frequency normalized on the juxtaposed line of.

[0042]

[Number 19] _{F 1 = (f 1 -f c0} ) / f s0

[Equation 20] f ₂ = (f ₁ sin θ) / c

Equation 21] _{F 2 = df 2 = (df} 1 sinθ) / c = {(dsi
_{nθ) / c} (f s0} F 1 + f c0)

The beam number m of the DBF circuit 9 shown in FIG.
The transfer characteristic H _m (f ₁ , θ) can be expressed by the following equation 22, and the transfer characteristic H _m (f ₁ , θ) is defined by the normalized time frequency F ₁ and the normalized time frequency F ₁ defined as described above. It can be converted to a frequency response G _m (F ₁ , F ₂ ) shown in Equation 23 when viewed as a two-dimensional digital filter using the generalized spatial frequency F ₂ . That is, the frequency response G _m (F ₁ , F ₂ )
Is 2 and the normalized time frequencies F ₁ and the normalized spatial frequency F ₂
It is a frequency response on a dimensional frequency plane.

(Equation 22)

(Equation 23)

Here, in _Equation 22, T _s0 is a sampling interval in the DBF circuit 9 and is the reciprocal of the sampling frequency f _s0 in the DBF circuit 9, that is, T _s0 = 1 /
It is represented by f _s0 . In Equations 22 and 23, K is the FIR digital filter 11-mn as described above.
Is the tap length. Equations (22) and (23) above are equations that hold when the element spacing d is assumed to be constant, and there is a restriction that this method can be applied only to a linear array where the element spacing d is constant. However, when the antenna elements 1-1 to 1-N are arranged at regular intervals in the vertical and horizontal directions in a grid pattern on a plane, it can be dealt with by expanding Expression 22 and Expression 23. That is, the present invention can be applied to a case where the digital filter is three-dimensional (two-dimensional space-time).

This DBF circuit 9 has a desired characteristic on a frequency-angle plane for forming a broadband beam in a desired direction.
By converting the normalized time frequency F ₁ and the normalized spatial frequency F ₂ plane into desired characteristics on a two-dimensional frequency plane and approximating the characteristics of the two-dimensional digital filter to the desired characteristics on the two-dimensional frequency plane It has been realized. In the above-described processing, the present inventors have proposed a broadband DBF circuit 9 for processing a radio wave for transmitting information using a carrier wave, as shown in FIG.
Desired characteristics in the frequency-angle plane indicated by 0 (a) are:
On the two-dimensional frequency plane, it has been found that it is represented by a desired passband shown in FIG. Here, in order to form a broadband multi-beam and then select a beam to form an adaptive broadband beam or a broadband null, the phase characteristics of a circuit forming each beam are linear with respect to frequency, and All the slopes must be uniform, but the method described below can satisfy that requirement.

First, the digital in-phase / quadrature signal x _n (k) input to the DBF circuit 9 was examined in detail, and as a result, the digital in-phase / quadrature signal x _n (k) corresponding to the signal incident at the incident angle θ was obtained. It has been found that the two-dimensional spectrum relating to time and space at the time of input to the DBF circuit 9 is on a straight line represented by the following equation on a two-dimensional frequency plane. FIG. 11 is a diagram schematically showing the two-dimensional spectrum Sp.

Equation 24] F2 = {(dsinθ) / c } (f s0 F1 + f c0) That is, the straight line represented by the number 24, located in the rectangle in the width direction of the center of the desired pass band shown in FIG. 10 (b) Is a straight line. Accordingly, the desired passband of the rectangle shown in FIG. 10 (b), can be represented by the inequality shown in the following equation, a first normalized normalized time frequencies F ₁ at a sampling frequency f _s0
And a two-dimensional frequency plane formed by a second axis of the normalized spatial frequency F ₂ normalized by the reciprocal of the element spacing, having a center on the second axis and having a predetermined width. And extends from the negative normalized time frequency to the positive normalized time frequency.

[0047]

Equation 25] {(dsinθ) / c} ( f s0 F 1 + f c0) -ε ≦ F 2 ≦
{(dsinθ) / c} ( f s0 F 1 + f c0) + ε where the epsilon in the number 25 is a predetermined small number corresponding to the width of the desired pass band of rectangles shown in Figure 10 (b). That is, the amplitude characteristic D in the desired pass band of the rectangular 10 shown by the number _{25 (b) (F 1,} F 2) is = 1, the desired passband in the external amplitude characteristic D (F _1, F ₂ ) = 0, if the DBF circuit 9 which is a two-dimensional digital filter capable of approximately realizing the amplitude characteristic of the incident angle θ
Can form a DBF circuit 9 that forms a broadband beam in the direction of. Here, in the first embodiment, F ₁ =
Even in the vicinity of ± 0.5, the amplitude characteristic D (F ₁ , F ₂ ) = 0
I made it.

Next, an amplitude characteristic having an amplitude in the above-described rectangular desired pass band, that is, FIG.
Amplitude characteristic D (F ₁ , F ₂ ) = 1 in the desired pass band of
Outside the desired passband, the amplitude characteristic D (F ₁ ,
A two-dimensional digital filter capable of approximately realizing the amplitude characteristic of F ₂ ) = 0 was studied.

In this embodiment, a one-dimensional zero-phase FIR type low-pass digital filter (hereinafter referred to as 1
It is called a dimensional original filter. ) Is transformed into a predetermined straight line corresponding to the beam forming direction on the two-dimensional frequency plane,
It is characterized in that the two-dimensional frequency characteristics of the DBF circuit 9 are approximated to desired characteristics to calculate the load coefficients am _{, n, s} .

(Equation 26) In Equation 26, _Mc is an odd number of a predetermined impulse response length of the one-dimensional prototype filter, and p (k) is an impulse response. 1
Although the dimensional prototype filter has a linear phase characteristic, here, for simplicity, only a zero-phase part excluding a linear phase component that does not affect the amplitude characteristic is considered. T _k (x) is the first k-th order
Kind Chebyshev polynomial, T ₀ (x) = 1, T
₁ (x) = x, T _k (x) = 2 × T _k−1 (x) −T
_k−2 (x); k = 2, 3,..., T _k (cos x) = co
s (k x).

First, the variable conversion applied to the one-dimensional zero-phase FIR type low-pass digital filter will be described, and then the calculation method of the weighting factors am _{, n, s} of the DBF circuit 9 will be described. The variable conversion in the present embodiment is cos (2ππ) appearing in Equation 26 of the frequency response of the one-dimensional prototype filter.
F) is replaced by the appropriate four functions cos (2πF ₁ ), sin (2πF ₁ ), c
This is performed by replacing the function S (F ₁ , F ₂ ) represented by os (2πF ₂ ) and sin (2πF ₂ ). Here, the function S (F ₁ ,
F ₂ ) is a function approximating the amplitude characteristic D _T (F ₁ , F ₂ ) shown in the following equation, and will be described later in detail.

Equation 27] _{D T (F 1, F 2} ) = cos [2πF 2 / {(f s0 / f c0) F 1
+1} -2πF _2shift ]

Here, F _2shift in _Expression 27 is _{given by Expression} 2
The shift amount represented by 8 is determined by the incident angle θ corresponding to the direction in which the beam is to be formed. In the present embodiment, p (k) in Expression 26 is the impulse response of the one-dimensional original filter. This equation 27 is an arbitrary one-dimensional frequency F of the one-dimensional original filter (actually, some frequencies are excluded).
Is mapped to a straight line on a two-dimensional frequency plane represented by the following equation.

## _EQU28 ## F _2shift = (df _c0 / c) sin θ

Equation 29] _{F = F 2 / {(f} s0 / f c0) F 1 +1} -F 2shift

For example, the one-dimensional frequency F = 0 of the one-dimensional original filter is represented by a straight line F ₂ = F _2shift {(f
_s0 / f _c0 ) F ₁ +1}. Therefore, if this mapping is applied to a one-dimensional FIR type narrow band low-pass digital filter, the pass band becomes a straight line F ₂ = F on a two-dimensional frequency plane.
It becomes _{2shift {(f s0 / f c0} ) F 1 +1} to appear in the vicinity. In other words, in order to form a beam in the direction of the incident angle θ, the pass band on the two-dimensional frequency plane is F ₂ = (df _c sin θ /
c) from _{{(f s0 / f c0)} F 1 +1} appear in the vicinity of the shift amount F
_2shift is determined in accordance with _Expression 28, and the _mapping of _Expression 27 or _{Expression 29} may be performed on the one-dimensional FIR type narrow band low-pass digital filter instead of the variable conversion method used in the above embodiment. Become. FIG. 12 shows an example when mapping is performed using Expression 29. In this example, the distance between the antenna element _{d = 0.45λ c (λ c =} c / f c0), assuming a beam forming direction theta = 40 °, the one-dimensional frequency F +0.5 -0.5 Are divided at intervals of 0.1, and are shown for each one-dimensional frequency F.

If the above-described mapping is ideally performed, each equal-amplitude line of the amplitude characteristic of the two-dimensional digital filter obtained by such a variable conversion follows the straight line of Expression 24 in the above-described embodiment. On the other hand, the straight line represented by Expression 24 corresponds to a straight line parallel to the frequency axis at the angle θ on the frequency-angle plane. Therefore, the fact that the equal-amplitude line of the amplitude characteristic of the two-dimensional digital filter on the two-dimensional frequency plane follows Expression 24 means that the equal-amplitude line is parallel to the frequency axis on the corresponding frequency-angle plane. This is D
This means that the directional characteristics of the BF circuit 9 do not depend on the frequency. That is, the directivity of the DBF circuit 9 as a two-dimensional digital filter obtained by such a variable conversion depends only on the incident angle θ and does not depend on the frequency.
In other words, in the present embodiment, variable conversion is performed on the one-dimensional FIR type narrow-band low-pass digital filter so that the directivity of the DBF circuit 9 as a two-dimensional digital filter does not depend on frequency.

Next, the amplitude characteristic D _T (F ₁ , F ₂ ) of Expression 27 is obtained.
A method of obtaining a function S (F ₁ , F ₂ ) that approximates the following will be described. This approximation is based on a window function method of terminating the impulse response to the frequency characteristic represented by Expression 27 with a two-dimensional window function (for example, “Nishikawa et al.,“ Design of Fan Filter by Fourier Transform Method ”, IEICE 6th. Proc. Of Digital Signal Processing Symposium, Lecture No. B3-
2, pp. 287-292, 1991 ", which is called the Fourier series method. ) Is used. The specific procedure is shown in the following procedures SS1 to SS4.

(Procedure SS1) The normalized time frequency F ₁ and the normalized spatial frequency F ₂ are set to be −0.5 ≦ F ₁ <0.5, −
Sampling is performed at appropriate intervals in the range of 0.5 ≦ F ₂ <0.5, and each sampled normalized time frequency F ₁
And amplitude characteristics D for each normalized spatial frequency F ₂
Each frequency response value of _T (F ₁ , F ₂ ) is calculated. If the fast Fourier transform is used later, the number of points to be sampled, J
(Even number) is preferably set to a value that can be represented by a power of 2 such as 128 points. (Procedure SS2) The amplitude characteristic D _T (F ₁ ,
F ₂ 2-dimensional inverse discrete respective frequency response value) (Fast) Fourier transform obtains an impulse response (complex) for the frequency response value of the amplitude characteristic _{D T (F 1, F 2} ). This impulse response is called an ideal impulse response, and h _TI (m ₁ ,
m ₂ ). Here, (m ₁ , m ₂ ) = (0, 0) is set to be the center of the ideal impulse response h _TI (m ₁ , m ₂ ). Arguments m ₁ and m ₂ are -J / 2, (-J /
2) +1, ...,-1,0,1, ..., (J / 2) -1. (Procedure SS3) The ideal impulse response h _TI (m ₁ , m ₂ ) is cut off by the two-dimensional window function w _2D (m ₁ , m ₂ ). That is, the ideal impulse response h _TI (m ₁ , m ₂ ) and the two-dimensional window function w _2D
Take the product of (m ₁ , m ₂ ). Here, the two-dimensional window function w
_2D (m ₁ , m ₂ ) can be represented, for example, by the following equation.

[Equation 30] w _2D (m ₁ , m ₂ ) = w _1D (m ₁ ) w _1D (m ₂ )

[0056] In Equation 30, w _1D (m _i), to the truncation number of terms L _a is an odd number, an argument m _i is the time that satisfies the following equation, the center of the window function m _i = 0 _(i = 1, 2)
Is represented by an appropriate window function other than a rectangular window such as a Hamming window function. If the following expression is not satisfied, the function value is 0.
Is a window function.

-(L _a −1) / 2 ≦ m _i ≦ (L _a −1) / 2 The impulse response h _T ′ (m ₁ , m ₂ ) obtained by this operation is:
It is expressed by the following equation.

H ′ _T (m ₁ , m ₂ ) = h _TI (m ₁ , m ₂ ) w _2D (m ₁ , m ₂ ) where − (L _a −1) / 2 ≦ _mi ≦ ( L _a -1) / 2, i
= 1,2

(Procedure SS4) Scaling is performed. This is because step S is performed so that the characteristics of the DBF circuit 9 are not deteriorated.
This is an operation of setting the maximum value and the minimum value of the frequency response value of the impulse response h _T ′ (m ₁ , m ₂ ) obtained in S3 to 1 and −1, respectively. Therefore, the frequency response value of the impulse response h _T ′ (m ₁ , m ₂ ) obtained in (step SS3) is calculated, and the maximum value h _T ′ _max and the minimum value h _T ′ _min are obtained. Then, the scaling constants c ₁ and c ₂ represented by the following equations are calculated, and the two-dimensional impulse response after scaling represented by the following equation 34 or 35 is calculated using the scaling constants c ₁ and c _2. Find h _T (m ₁ , m ₂ ). In the two-dimensional impulse response h _T (m ₁ , m ₂ ), h _T (m ₁ , m ₂ ) = h _T * (−
m ₁ , −m ₂ ) and therefore its frequency response S
(F ₁ , F ₂ ) is a real number, and cos (2πF ₁ ), sin
(2πF ₁ ), cos (2πF ₂ ), sin (2πF ₂ )
Function. Here, * of h _T * (− m ₁ , −m ₂ )
Represents that it is a conjugate complex number.

C ₁ = 2 / (h ′ _Tmax −h ′ _Tmin ), c ₂ = c ₁
h ' _Tmax -1

When (m ₁ , m ₂ ) ≠ (0, 0), h _T (m ₁ ,
m ₂ ) = c ₁ h ′ _T (m ₁ , m ₂ )

When (m ₁ , m ₂ ) = (0, 0), h _T (m ₁ ,
m ₂ ) = c ₁ h ′ _T (0,0) −c ₂

Next, using the two-dimensional impulse response h _T (m ₁ , m ₂ ) obtained by the above procedure, the F
Weighting coefficient a of IR digital filter 11-mn
A method for calculating _{m, n, and s} will be described. In the following calculation method, the frequency response of the two-dimensional impulse response h _T (m ₁ , m ₂ ) is a function S (F ₁ , F ₂ ). The calculation method of the weighting factors am _{, n, s} of the FIR digital filter 11-mn in the DBF circuit 9 is as follows:
Consists of three.

<Step S1> First, a frequency response P (F) of a one-dimensional linear phase FIR type low-pass digital filter having an impulse response length of M _c (odd) is obtained. 1
The frequency response P (F) of the dimensional linear phase FIR type low-pass digital filter is the same as Equation 26 and is expressed by the following equation.
Here, in this embodiment, the value of the number of taps M _c in order to obtain good characteristics is set smaller than the number of elements N.

[Equation 36]

<Step S2> Next, the cos (2πF) on the right side in Expression 36 is converted into a function S (F ₁ , F ₂ ) expressed by the following equation:
Perform variable conversion to replace with. Therefore, the replaced frequency response G _m (F ₁ , F ₂ ) is represented by the following equation. Here, in the following equation, only the zero phase portion is shown. Similarly, only the zero-phase portion is shown in the equations described below.

(37) here,

(38)

<Step S3> Next, similarly,
Impulse response _{g m (m 1, m 2} ) of the frequency response G _m expressed by 7 (F _1, F ₂₎ is calculated, and the impulse response g _m (m _1,
m ₂₎ seek a _{m, n, s} from.

A _{m, n, s} = g _m (s− (K−1) / 2, −n + (N−1) / 2 + 1) where n = 1, 2,..., N; s = 0 , 1, ..., K-1
And K is the number of taps.

The iso-amplitude line of the amplitude characteristic on the frequency-angle plane of the DBF circuit 9 obtained by the above method appears approximately parallel to the frequency axis on the frequency-angle plane. Therefore, the directional characteristics can be approximately matched over the target frequency band. Further, the phase characteristic of the DBF circuit 9 in the present embodiment is linear, and the slope of the phase shift characteristic is determined by the value of the tap number K. Therefore, when forming a wideband multi-beam using this method, the values of the number of taps K in each single-beam DBF circuit need only be uniform. <End of supplementary explanation>

[0063]

As described above in detail, according to the method of controlling an array antenna according to the first aspect of the present invention, three or more odd-numbered N pieces are arranged at regular intervals on a straight line. A shared transmitting / receiving array antenna comprising antenna elements, receiving means for receiving the odd N received signals received by the odd N antenna elements of the transmitting / receiving array antenna, and the odd N received signals received by the receiving means. A receiving multi-beam forming circuit for forming a predetermined number M of multi-beam signals by multiplying a received signal by a predetermined receiving weight coefficient for each of the multi-beam signals; Selecting I multi-beam signals having higher power or higher power than a predetermined threshold power among the multi-beam signals. Selecting means for outputting, and multiplying each of the I multi-beam signals selected by the selecting means by a predetermined adaptive control weighting factor and adding the multiplication results to output a signal of the addition result as a reception signal Multiplying and adding means for dividing an input transmission signal into odd N transmission signals and multiplying the transmission signal by a predetermined transmission load coefficient to output odd N transmission signals; Based on the received signal that is output, using a predetermined adaptive control algorithm, such that the main beam of the array antenna is directed to the arrival direction of the desired wave and the level of the reception signal in the arrival direction of the interference wave is made zero, Coefficient calculating means for calculating adaptive control weighting coefficients respectively corresponding to the I multi-beam signals and setting the weights in the multiplying and adding means; Transmitting means for amplifying and transmitting the odd-N transmission signals to the odd-N antenna elements and transmitting the signals by transmitting the signals to the odd-N antenna elements. Utilizing the coefficients of a predetermined broadband digital multi-beam forming circuit having a frequency-independent beam pattern having a determined multi-beam forming direction, based on the receiving frequency and the transmitting frequency, the receiving multi-beam forming circuit A first step of calculating and setting a multi-beam forming weight coefficient with respect to a weight coefficient and a transmission frequency; and a multi-beam forming weight coefficient with respect to the receiving multi-beam forming circuit and the transmission frequency calculated by the first step. Based on the adaptive control load coefficient calculated by the coefficient calculating means, A second step of calculating and setting the transmission weight coefficient of the distribution and multiplication means so that both the transmission signal and the reception signal form a main beam in the direction of the desired wave and form a zero in the direction of arrival of the interference wave. Prepare. Therefore, the present invention has the following specific effects. (A) Even when the transmission and reception frequencies are different, at the time of transmission, the beam can be controlled such that the main beam and the zero point are directed in the same direction as the desired signal direction and the interference signal direction at the time of reception. (B) The circuit configuration is simpler than the conventional example.

Further, according to the array antenna control device of the second aspect of the present invention, three or more odd-numbered N antenna elements arranged at equal intervals at a predetermined interval on a straight line. A transmitting / receiving array antenna, receiving means for receiving odd N received signals received by the odd N antenna elements of the transmitting / receiving array antenna, and an odd N received signal received by the receiving means. A receiving multi-beam forming circuit for forming a predetermined M multi-beam signals by multiplying a predetermined receiving weight coefficient for each multi-beam signal; Selection of selecting and outputting I multi-beam signals having greater power or having power greater than a predetermined threshold power A multiplication and addition step of multiplying each of the I multi-beam signals selected by the selection means by a predetermined adaptive control weighting factor and adding the multiplication results to output a signal of the addition result as a reception signal Means for dividing an input transmission signal into odd N transmission signals and multiplying the transmission signal by a predetermined transmission weight coefficient to output odd N transmission signals; and output from the multiplication and addition means. Using the predetermined adaptive control algorithm based on the received signal, the main beam of the array antenna is directed to the arrival direction of the desired wave and the level of the reception signal in the arrival direction of the interference wave is reduced to zero. Coefficient calculating means for calculating adaptive control load coefficients respectively corresponding to the multi-beam signals and setting the multiplying and adding means in the multiplying and adding means; Transmitting means for amplifying a signal and outputting the amplified signal to the odd number N antenna elements, wherein the transmitting and receiving frequencies are different from each other. Utilizing coefficients of a predetermined broadband digital multi-beam forming circuit having a frequency-independent beam pattern having a forming direction, based on a receiving frequency and a transmitting frequency, a weight coefficient and a transmitting frequency of the receiving multi-beam forming circuit are determined. First calculating means for calculating and setting a multi-beam forming load coefficient for
The transmission signal and the reception signal are calculated based on the multi-beam forming load coefficient for the receiving multi-beam forming circuit and the transmission frequency calculated by the first calculating means, and the adaptive control weighting coefficient calculated by the coefficient calculating means. A second calculating means for calculating and setting the transmission weight coefficient of the distribution multiplying means so as to form a main beam in a direction of a desired wave and to form a zero point in a direction of arrival of the interference wave together with the signal. Therefore, the present invention has the following specific effects. (A) Even when the transmission and reception frequencies are different, at the time of transmission, the beam can be controlled such that the main beam and the zero point are directed in the same direction as the desired signal direction and the interference signal direction at the time of reception. (B) The circuit configuration is simpler than the conventional example.

[0065]

[0066]

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a configuration of a wireless transmission / reception system including an array antenna control device according to an embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration of a virtual DBF circuit 9 used when calculating a filter coefficient in a filter coefficient calculator 30 of FIG. 1;

3 is a virtual FIR digital filter 1 of FIG.
FIG. 2 is a block diagram illustrating a configuration of 1-mn.

FIG. 4 is a graph showing directivity characteristics in a 20 ° direction among transmission / reception multi-beams of the wireless transmission / reception system of the embodiment of FIG. 1;

FIG. 5 is a graph showing transmission / reception directional characteristics of the wireless transmission / reception system of the embodiment of FIG. 1;

6 is an enlarged view around an angle of 35 ° in FIG. 5;

FIG. 7 is an enlarged view around the angle of −25 ° in FIG. 5;

FIG. 8 is a graph showing a directional characteristic in a 14 ° direction among transmission / reception multi-beams of a conventional radio transmission / reception system.

FIG. 9 is a graph showing transmission directivity characteristics of a conventional wireless transmission / reception system.

10A is a graph illustrating a beam in a desired direction on a frequency-angle plane for explaining the operation principle of the virtual DBF circuit 9 in FIG. 2; FIG.
7 is a graph showing a desired passband when a beam in a desired direction in a frequency-angle plane of FIG.

FIG. 11 shows a two-dimensional spectrum S on a two-dimensional frequency plane.
It is the figure which showed p typically.

FIG. 12 is a diagram illustrating an example in which equation (6) is a one-dimensional frequency F
Is a graph when expressed on a two-dimensional frequency plane using as a parameter.

[Explanation of symbols]

1-1 to 1-N: antenna element, 2-1 to 2-N: transmitting / receiving device, 3: in-phase distributor, 4-1 to 4-N: multiplier, 5: digital beam forming circuit (DBF circuit), 6: Beam selector, 7-1 to 7-I: Multiplier, 8: Adder, 9: Virtual DBF circuit, 10-1 to 10-M: Adder, 11-1-1 to 11-M-N ... FIR digital filter, 20 ... circulator, 21 ... receiver, 22 ... A / D converter, 23 ... IQ signal converter, 25 ... D / A converter, 26 ... transmitter, 30 ... filter coefficient calculator, 31 Multi-beam load coefficient calculator 32 Transmit weight coefficient calculator 91-1 to 91- (K-1) Delayer 92-1 to 92-K Multiplier 93-1 to 93- ( K-1): Adder, 100: Array antenna.

──────────────────────────────────────────────────の Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H01Q 3/26-3/42 H01Q 25/00-25/04

Claims (2)

(57) [Claims] 1. A shared transmission / reception array antenna consisting of three or more odd-numbered N antenna elements arranged on a straight line at predetermined intervals at equal intervals, and reception by an odd-numbered N antenna elements of the transmission / reception array antenna Receiving means for receiving the received odd N received signals, and multiplying the odd N received signals received by the receiving means by a predetermined reception weighting coefficient for each multi-beam signal to obtain a predetermined M Receiving multi-beam forming circuit for forming a plurality of multi-beam signals, and having a larger power among the M multi-beam signals formed by the receiving multi-beam forming circuit, or having a power higher than a predetermined threshold power. Selecting means for selecting and outputting the I multi-beam signals having a large power; and selecting the I multi-beam signals selected by the selecting means, Multiplying and adding means for multiplying each by a predetermined adaptive control weighting factor and adding the multiplication result to output a signal of the addition result as a reception signal; and distributing an input transmission signal to an odd number N transmission signals. Dividing and multiplying by a predetermined transmission weight coefficient to output an odd number N of transmission signals; and a reception signal output from the multiplication and addition means,
Using a predetermined adaptive control algorithm, the main beam of the array antenna is directed to the arrival direction of the desired wave and the level of the received signal in the arrival direction of the interference wave is set to zero.
Coefficient calculating means for calculating adaptive control load coefficients respectively corresponding to the plurality of multi-beam signals and setting the multiplied and added means in the multiplying and adding means; A transmitting means for transmitting by outputting to N antenna elements, the method for controlling an array antenna for a wireless communication system having different transmission / reception frequencies, wherein a frequency-independent signal having a predetermined multi-beam forming direction is provided. Utilizing the coefficients of a predetermined broadband digital multi-beam forming circuit having a neutral beam pattern,
A first step of calculating and setting a weighting factor of the receiving multi-beam forming circuit and a multi-beam forming weighting factor for the transmitting frequency based on the transmitting frequency; and a receiving multi-beam forming calculated by the first step. Based on the multi-beam forming weight coefficient for the circuit weight coefficient and the transmission frequency, and the adaptive control weight coefficient calculated by the coefficient calculating means, both the transmission signal and the reception signal form a main beam in a desired wave direction; A second step of calculating and setting a transmission weight coefficient of the distribution and multiplication means so as to form a zero point in the direction of arrival of the interference wave. 2. A shared transmission / reception array antenna composed of three or more odd-numbered N antenna elements arranged on a straight line at predetermined intervals at equal intervals, and reception by an odd-numbered N antenna elements of the transmission / reception array antenna. Receiving means for receiving the received odd N received signals, and multiplying the odd N received signals received by the receiving means by a predetermined reception weighting coefficient for each multi-beam signal to obtain a predetermined M Receiving multi-beam forming circuit for forming a plurality of multi-beam signals, and having a larger power among the M multi-beam signals formed by the receiving multi-beam forming circuit, or having a power higher than a predetermined threshold power. Selecting means for selecting and outputting the I multi-beam signals having a large power; and selecting the I multi-beam signals selected by the selecting means, Multiplying and adding means for multiplying each by a predetermined adaptive control weighting factor and adding the multiplication result to output a signal of the addition result as a reception signal; and distributing an input transmission signal to an odd number N transmission signals. Dividing and multiplying by a predetermined transmission weight coefficient to output an odd number N of transmission signals; and a reception signal output from the multiplication and addition means,
Using a predetermined adaptive control algorithm, the main beam of the array antenna is directed to the arrival direction of the desired wave and the level of the received signal in the arrival direction of the interference wave is set to zero.
Coefficient calculating means for calculating adaptive control load coefficients respectively corresponding to the plurality of multi-beam signals and setting the multiplied and added means in the multiplying and adding means; A control unit for an array antenna for a wireless communication system having transmission and reception frequencies different from each other, comprising a transmission means for transmitting by outputting to N antenna elements, wherein a frequency-independent signal having a predetermined multi-beam forming direction is provided. Utilizing the coefficients of a predetermined broadband digital multi-beam forming circuit having a neutral beam pattern,
First calculating means for calculating and setting a weighting factor of the receiving multi-beam forming circuit and a multi-beam forming weighting factor for the transmitting frequency based on the transmitting frequency; and a receiving multi-factor calculated by the first calculating means. Based on the beam forming circuit load coefficient and the multi-beam forming load coefficient for the transmission frequency, and the adaptive control load coefficient calculated by the coefficient calculating means, both the transmission signal and the reception signal form a main beam in a desired wave direction. And a second calculating means for calculating and setting the transmission load coefficient of the distribution multiplying means so as to form a zero point in the direction of arrival of the interference wave.