JPH09182429A - Resonance type forward converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To realize the zero-voltage switching element without excessively enlarging the voltage/current in switching by connecting the series circuit of a diode having the carrier-life time equal to or higher than a switching period and a voltage clamping capacitor to a switching element in parallel. SOLUTION: A diode 13 has the carrier-life time equal to or more than the switching period of a switching element 3. A control circuit 12 imparts the control signal to the switching element 3 after the reverse conduction of the diode 13 is finished or the voltage of the switching element 3 becomes zero volt, further detects the DC output voltage, current or power between DC output terminals 10 and 10' and imparts the control signal for setting these values at the specified values to the switching element 3. The control signal imparts any of on-time control, off-time control, turn-on-time control or current- made control wherein main current value is present, or their combinational control.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance type forward converter that switches a switching element in a state where an applied voltage is substantially zero.

[0002]

2. Description of the Related Art As a conventional method, for example, Japanese Patent Laid-Open No.
There is a structure disclosed in No. 290216.
The resonant converter disclosed in this publication has a structure as shown in FIG. In the figure, 1 is a DC power supply, 2 is a transformer having a primary winding 2A and a secondary winding 2B of the illustrated polarities respectively, 3 is a switching element composed of MOSFET or bipolar transistor, and 4 is parallel to the switching element 3. A connected resonance capacitor, Lr is an inductance corresponding to the sum of the inductance of the wiring that provides a part or all of the resonance inductance and the leakage inductance of the transformer 2, and 5 is a saturable element having the characteristics shown in FIG. The switching element 3 is an inductor and gives a large inductance to the circuit during the resonance period of a small current flowing from the positive electrode of the DC power supply 1 through the primary winding 2A of the transformer 2 and the saturable inductor 5 when the switching element 3 is off. When it is turned on, it is magnetically saturated and its inductance decreases sharply. .

A rectifying diode 6 is connected in series to the secondary winding 2 B of the transformer 2, and a freewheeling diode 7 is connected across the secondary winding and the rectifying diode 6. Further, an output filter including a smoothing inductor 8 and a smoothing capacitor 9 is connected between the freewheeling diode 7 and the output terminals 10 and 10 '. Furthermore, a resonance capacitor 11 is connected in parallel to the rectifying diode 6, and the control circuit 12 gives a control signal to the switching element 3 to maintain the DC output voltage between the output terminals 10 and 10 'at a set voltage. .

Although a detailed description of this resonance type converter is omitted here, there is a feature that zero voltage switching can be realized in a wide current range. The typical operation waveform is shown in FIG.
Shown in

[0005]

However, in such a conventional resonant converter, when a desired resonant operation is requested over a wide range of load current, the voltage applied across the switching element becomes excessive. Therefore, it becomes necessary to use a high breakdown voltage element as a switching element, which inevitably increases the cost.

The present invention switches between a small resonance capacitance and a large voltage clamp capacitance by using a diode having a carrier lifetime equal to or longer than the switching period, thereby obtaining an optimum resonance operation and over a wide range of load currents. An object is to realize zero voltage switching operation without increasing the voltage / current of a switching element and to improve power efficiency.

[0007]

In order to solve such a problem, in the first invention, a DC power supply, a primary winding connected in series to the DC power supply, and the primary winding magnetically A transformer having a coupled secondary winding and a primary winding of the transformer are connected in series, and have a predetermined inductance in a low power region and magnetic saturation for a predetermined voltage integration application. A saturable inductor that exhibits a small inductance,
A switching element connected in series with the saturable inductor; a resonance capacitor connected in parallel with the switching element; and a rectifying diode connected in series with one terminal of the secondary winding of the transformer. A freewheeling diode connected across the rectifying diode and the other terminal of the secondary winding of the transformer and an output filter, and a control signal is supplied to the switching element. In a resonant forward converter having a control circuit for controlling conduction, a diode having a carrier lifetime longer than or equal to a switching period of the switching element.
There is provided a resonant forward converter in which a series circuit of a capacitor and a capacitor is equivalently connected in parallel to the switching element.

In order to solve such a problem, in the second invention, the resonance capacitor is an output capacitance of the switching element, and the resonance type forward converter according to claim 1. I will provide a.

In order to solve such a problem, the third invention is characterized in that the resonance capacitance is composed of an output capacitance of the switching element and a capacitance of a resonance capacitor connected in parallel with the switching element. A resonant forward converter according to claim 1 is provided.

In order to solve such a problem, in the fourth invention, the saturable inductor includes a first winding wound around a core of rectangular hysteresis and a linear winding connected in parallel to the first winding. The resonance type forward converter according to claim 1, comprising an inductor.

In order to solve such a problem, in the fifth invention, the saturable inductor is a first winding wound around a core of rectangular hysteresis and a first winding magnetically coupled to the first winding. The resonance type forward converter according to claim 1, wherein the resonance type forward converter comprises two windings and a linear inductor connected in parallel to the second winding.

In order to solve such a problem, in the sixth invention, the saturable inductor includes a first winding wound around a core of rectangular hysteresis and a first winding magnetically coupled to the first winding. 2. The resonance type forward converter according to claim 1, characterized in that it comprises two windings and a third winding connected in parallel to the second winding and not magnetically coupled to the first winding. -Provide data.

In order to solve such a problem, in the seventh invention, of the energy stored in the capacitor connected in series to the diode, the transformer is in reverse conduction of the diode. 2. A discharge circuit for discharging energy that could not be returned to the DC power supply through the junction is connected across the junction between the diode and the capacitor and one end of the DC power supply. A resonant forward converter as described is provided.

In order to solve such a problem, an eighth aspect of the present invention provides a resonance type forward converter according to claim 7, wherein the discharge circuit exhibits a variable impedance.

In order to solve such a problem, in the ninth invention, the control circuit detects that the reverse current of the diode becomes zero, or the zero voltage of the switching element, or 3. A resonance type forward converter according to claim 1, wherein a turn-on signal is output to said switching element after detecting the lowest voltage and ending reverse conduction of said diode. .

[0016]

BEST MODE FOR CARRYING OUT THE INVENTION One embodiment of the present invention will be described with reference to FIG. 1. In the figure, the same symbols as those shown in FIG. 8 indicate corresponding members. In FIG. 1, the diode 13 has a carrier lifetime equal to or longer than the switching period of the switching element 3. A diode with a long carrier lifetime has the property of essentially holding reverse conduction for a long time compared to a short one, but if the same amount of carriers as the accumulated carriers is poured from the reverse direction, the diode's reverse blocking capability Will recover. The present invention is a resonant forward converter utilizing this knowledge. Reference numeral 14 is a voltage clamping capacitor for limiting the voltage applied across the switching element 3.

The control circuit 12 gives a control signal to the switching element 3 after the diode 13 finishes the reverse conduction or when the voltage of the switching element 3 becomes zero volt, and further, between the DC output terminals 10 and 10 ′. Of the DC output voltage, the current, or the electric power is detected, and a control signal that makes them a predetermined value is given to the switching element 3. The control signal provides any one of on-time control, off-time control, turn-on time point control, on-time control by current mode control with a main current value interposed, or a combination of these.

In order to deepen the above understanding, a typical operation mode for keeping the output voltage at a constant voltage according to the present invention will be described with reference to FIG. In the following description, the saturable inductor 5 is described as being saturated at a predetermined current value. However, it is assumed that its constituent means magnetically saturates in accordance with a voltage integral value obtained by integrating the applied voltage with time. Be accurate. However, since there is a current value corresponding to this voltage integrated value in one steady state operation explanation,
By treating this current value as an apparent saturation current value, the same treatment as described below can be performed.

In FIG. 2, V4 is the voltage across the switching element 3, that is, the waveform of the voltage across the resonance capacitor 4, and I1 and I2 are the primary winding 2A and secondary winding 2B of the transformer 2, respectively. Waveform of flowing current, V2 is transformer 2
2 is a waveform of the voltage across the secondary winding 2B.

Period 1 (t 0 <t ≦ t 1) The switching element 3 is in the ON state, and the saturable inductor 5 is in the magnetic saturation state. At this time, a current equal to the sum of the current obtained by converting the current of the inductor 8 on the secondary side into the primary side and the exciting current of the transformer 2 flows through the switching element 3.

Period 2 (t1 <t ≦ t2) When the switching element 3 is turned off at time t1, the current flowing through the switching element 3 until then flows into the resonance capacitor 4. As a result, the resonance capacitor 4 is charged, and the voltage V4 across the terminals thereof rises rapidly and reaches the same voltage value as the voltage V1 of the DC power supply 1, and the voltage applied to the winding of the transformer 2 Becomes zero. During this period, the voltage V4 of the resonance capacitor 4 increases almost linearly. This is because the inductor 8 usually has a sufficiently large value for DC smoothing, and its current is almost constant during this period.
This is because the exciting current of (1) changes little during this period, so that the charging current of the resonance capacitor 4 becomes substantially constant. When the winding voltage of the transformer 2 becomes zero, the freewheeling diode 7, which has been reverse-biased by that voltage, starts conducting, and the freewheeling diode 7 and the rectifying diode 7 start conducting. The secondary winding 2B of the transformer 2 is short-circuited with the switch 6. This time is defined as t2.

Period 3 (t2 <t ≦ t3) Since the secondary winding 2B of the transformer 2 is short-circuited at time t2,
The exciting current of the transformer 2 is kept substantially constant during this period.
As for the charging current flowing into the resonance capacitor 4, the load current begins to shift to the freewheeling diode 7 as the freewheeling diode 7 starts conducting.
Although it starts to decrease, the inductance Lr corresponding to the sum of the wiring inductance and the leakage inductance of the transformer 2 and the saturation inductance Ls of the saturable inductor 5
Since s (equal to or less than Lr) exists, it does not become zero immediately. Due to the resonance determined by the sum of these inductances and the resonance capacitor 4, the voltage V4 of the resonance capacitor 4 continues to increase, and when the voltage V5 of the voltage clamping capacitor 14 is reached, the diode 13 starts to conduct in the forward direction. This time is defined as t3.

Period 4 (t3 <t ≦ t4) The forward conduction of the diode 13 at time t3 causes the inductance Lr corresponding to the sum of the wiring inductance and the leakage inductance of the transformer 2 and the saturable inductor 5. Due to the resonance determined by the sum of the saturation inductance Lss and the sum of the resonance capacitor 4 and the voltage clamp capacitor 14, the voltage of the resonance capacitor 4 and the voltage clamp capacitor 14 increases, the charging current decreases, and the diode 13 A charge is stored in the junction due to the resonance current flowing. Here, since the voltage waveforms of the resonance capacitor 4 and the voltage clamping capacitor 14 are substantially determined by the exciting inductance Lm of the transformer 2 and the capacitance of the voltage clamping capacitor 14, the voltage clamping capacitor 14 has time dependency. The value should be chosen large enough to be small. That is, in order to reduce the change in the voltage of the switching element 3 during this period, the voltage clamping capacitor 14 needs to have a sufficiently large capacitance. When the charging currents of the resonance capacitor 4 and the voltage clamping capacitor 14 decrease to the value of the exciting current of the transformer 2, the rectifying diode 6 is turned off and the short-circuited state of the secondary winding 2B of the transformer 2 is released. To be done. This time is t 4.

Period 5 (t4 <t ≦ t5) The rectifying diode 6 is turned off, and the secondary winding 2B of the transformer 2 is turned on.
When the short-circuited state is released, the excitation inductance Lm of the transformer 2 and the saturation inductance L of the saturable inductor 5
The ss, the resonance capacitor 4, the voltage clamping capacitor 14, the secondary resonance capacitor 11, and the inductance Lr form a resonance circuit. According to this resonance, the voltages of the resonance capacitor 4, the voltage clamping capacitor 14 and the secondary resonance capacitor 11 change.

Next, when the current of the saturable inductor 5 decreases to the magnetic saturation current value, the saturable inductor 5
Goes out of the magnetic saturation state, the inductance of the saturable inductor 5 changes to the unsaturated inductance Lsn, and becomes a large value which is the same level as the exciting inductance of the transformer 2. The time when the saturable inductor 5 comes out of the magnetically saturated state is t5.

Period 6 (t5 <t ≦ t6) When the saturable inductor 5 comes out of the magnetic saturation state at the time t5, the exciting inductance Lm of the transformer 2, the unsaturated inductance Lsn of the saturable inductor 5, the resonance capacitor 4, The voltage clamp capacitor 14, the secondary side resonance capacitor 11, the wiring inductance and the leakage inductance Lr of the transformer 2 form a resonance circuit and resonate. According to this resonance, the resonance capacitors 4 and 11 and the voltage clamping capacitor 14 are charged and the voltage rises.

The resonance energy of the exciting inductance Lm of the transformer 2, the unsaturated inductance Lsn of the saturable inductor 5, and the inductance Lr are all transmitted to the resonance capacitor 4, the voltage clamping capacitor 14, and the secondary side resonance capacitor 11. Then, the diode 13 becomes
In the forward conduction, the electric charge stored in the junction makes the conduction reverse, and the exciting inductance L of the transformer 2
m, the unsaturated inductance Ls of the saturable inductor 5
n, resonance capacitor 4, voltage clamp capacitor 1
4, the resonance capacitor 11 on the secondary side, the wiring inductance, and the leakage inductance Lr of the transformer 2 form a resonance circuit. According to this resonance, the resonance capacitors 4 and 1
1 and the capacitor 14 for voltage clamping are discharged and the voltage rises.

Here, the ratio of the power accumulated in the voltage clamping capacitor 14 due to the forward conduction of the diode 13 and the power discharged due to the reverse conduction of the diode 13 is defined as the power recovery rate, and the present invention will be described with reference to FIG. The characteristics of the diode 13 necessary for the above will be described. FIG. 3 shows power recovery rate characteristics with respect to (carrier lifetime of diode 13 / switching cycle). According to FIG. 3, when a diode having a carrier lifetime smaller than the switching cycle is used as the diode 13, the current recovery rate becomes extremely poor, and the uncollected power of the voltage clamping capacitor 14 with respect to the output power becomes extremely large.

This is the inductance corresponding to the sum of the exciting inductance Lm of the transformer 2, the unsaturated inductance Lsn of the saturable inductor 5, the wiring inductance and the leakage inductance of the transformer 2 due to the forward conduction of the diode 13. This means that the resonance energy transmitted from Lr to the voltage clamping capacitor 14 can be recovered to the DC power source 1 by the reverse conduction of the diode 13, and is applied to both ends of the voltage clamping capacitor 14 and the switching element 3. The voltage becomes high and the discharge circuit 15 causes the voltage clamping capacitor 14
When the above energy is discharged, the loss becomes large. Therefore, in order to efficiently realize the energy recovery from the voltage clamping capacitor 14 to the DC power supply 1 by the reverse conduction of the diode 13, it is necessary to use a diode having a carrier lifetime longer than the switching cycle.

When the resonance capacitor 11 in parallel with the rectifying diode 6 is charged and discharged to zero, the rectifying diode 6 becomes conductive, and the secondary winding 2B of the transformer 2 becomes a rectifying diode. A short circuit is made between the node 6 and the freewheeling diode 7. This time is defined as t6.

Period 7 (t6 <t ≦ t7) When the secondary winding of the transformer 2 is short-circuited at time t6, the unsaturated inductance Lsn of the saturable inductor 5, the resonance capacitor 4, the voltage clamping capacitor 14, And the inductance Lr forms a resonance circuit. According to this resonance, the resonance capacitor 4 and the voltage clamp capacitor 14
Is further discharged. By the time t6, the resonance capacitor 4 and the voltage clamping capacitor 14 are discharged by the energy stored in the unsaturated inductance of the saturable inductor 5 as a current in the discharging direction of the resonance capacitor 4 and the voltage clamping capacitor 14. to continue. At the diode 13, the time t7 when the electric charge stored in the junction becomes zero.
The reverse conduction is finished with.

Period 8 (t7 <t ≦ t8) When reverse conduction of the diode 13 ends at time t7,
The unsaturated inductance Lsn of the saturable inductor 5, the resonance capacitor 4, and the inductance Lr corresponding to the sum of the wiring inductance and the leakage inductance of the transformer 2 form a resonance circuit. By time t6,
The resonance capacitor 4 continues to discharge due to the energy stored in the unsaturated inductance of the saturable inductor 5 as a current in the discharging direction of the resonance capacitor 4.

When the voltage of the resonance capacitor 4 becomes zero, the body drain diode of the switching element 3 (when the switching element is not a FET, a diode connected in antiparallel thereto) becomes conductive. This time is defined as t8.

Period 9 (t8 <t ≦ t9) When the body diode of the switching element 3 or the diode connected in antiparallel thereto is conducting, the switching element 3 is turned on, the applied voltage is Turn-on switching can be realized with zero. In this operation mode, since the rectifying diode 6 is already turned on at time t5 and the winding of the transformer 2 is short-circuited, the saturable inductor 5 bears most of the DC power supply voltage V1,
The current increases linearly in the forward direction, and when it reaches the saturation current value, the saturable inductor 5 reaches magnetic saturation.
This time is set to t9. During this time, no power is supplied to the output, but since the secondary winding 2B of the transformer 2 is short-circuited, there is no change in the magnetic flux of the transformer 2.

Period 10 (t9 <t ≦ t10) When the saturable inductor 5 is magnetically saturated at the time t9, the entire voltage V1 of the DC power supply 1 corresponds to the sum of the wiring inductance and the leakage inductance of the transformer 2. The inductance Lr and the saturation inductance Lss of the saturable inductor 5 are shared. Since the sum of the inductances is small, the currents of the switching element 3 and the rectifying diode 6 increase rapidly, and when the current of the rectifying diode 6 becomes equal to the current of the inductor 8 at time t10, the freewheeling diode is turned on. -Do 7 is reverse biased and turns off. When the freewheeling diode 7 is turned off, the transformer 2
A voltage converted from the voltage V1 of the DC power supply 1 by the number of turns appears in the next winding 2B, and power is supplied from the DC power supply 1 to the secondary side through the switching element 3, the transformer 2, and the rectifying diode 6.

Thereafter, the operation returns to the operation in period 1 and the same operation as described above is repeated. The waveform of each part is as shown in FIG. Regarding the above operation, there are some periods when the operation is slightly different such as the order of each period due to the difference in the constants of the circuit components used, but the carrier life equal to or more than the switching period, which is the characteristic of this resonant forward converter, The basic operation of the diode 13 having a time is the same, and therefore will be omitted.

When the carrier lifetime of the diode 13 is sufficiently long with respect to the switching cycle, as shown in FIG. 3, the exciting inductance Lm of the transformer 2, the unsaturated inductance Lsn of the saturable inductor 5, the wiring inductance and the transformer 2 are used. The energy transferred from the inductance Lr corresponding to the sum of the leakage inductance to the voltage clamping capacitor 14 is almost 100% DC through the exciting inductance Lm of the transformer 2, the unsaturated inductance Lsn of the saturable inductor 5 and the inductance Lr. Since the power is recovered to the power supply 1, if the diode having a large leak current is used as the diode 13, the discharge circuit 15 becomes unnecessary.

Further, when the carrier life time of the diode 13 and the switching cycle are approximately the same, even if the unrecovered energy of the voltage clamping capacitor 14 is discharged, the loss of the discharge circuit 15 is very small and the conversion efficiency is not deteriorated. In addition, the operation as described above can be obtained.

An embodiment of the discharge circuit 15 to be added when the current recovery rate of the diode 13 is low and the uncollected amount of the resonance energy accumulated in the voltage clamping capacitor 14 is large is shown in FIGS. Shown in (3). Terminals 15-a and 15-b in FIG. 4 are terminals 15-a and 15-a of the discharge circuit 15 of FIG.
15-b respectively. In FIG. 4 (1), the terminal 15
A resistor 17 is connected between −a and 15−b, and the resistor 17 consumes an unrecovered portion of the resonance energy that was not recovered during the recovery operation of the diode 13. Since this discharge circuit positively uses a diode having a carrier lifetime equal to or longer than the switching cycle as the diode 13, the unrecovered portion of the resonance energy is very small, and the operation as described above is performed without deteriorating the conversion efficiency. Can be obtained.

In FIG. 4B, the collector of the PNP transistor 21 is connected to the terminal 15-a, the resistor 18 is connected to the terminal 15-b, and the emitter of the transistor 21 is connected to the other end of the resistor 18. A Zener diode 20 is connected between the collector and the base of the transistor 21 in the orientation shown in the drawing, and a resistor 19 is connected between the base and the emitter. The circuit shown in Fig. 4 (2) is the collector of transistor 21
A current flows from the terminals 15-a to 15-b so that the voltage between the emitters becomes the voltage of the Zener diode 20, and the voltage between these terminals is (voltage of the Zener diode 20) + (voltage drop of the resistor 18). ). When applied to the circuit of FIG. 1, it operates so that the voltage of the voltage clamping capacitor 14 becomes (voltage of the Zener diode 20) + (voltage drop of the resistor 18). Therefore, the peak voltage of the switching element 3 is [(Zener diode 2
0 voltage) + (voltage drop of the resistor 18)] + (voltage Ei of the DC power supply 1).

The circuit of FIG. 4 (3) is obtained by changing the Zener diode 20 of the circuit of FIG. 4 (2) into a control circuit 16, and the other configuration and operation are the same as those of FIG. 4 (2). .
The control circuit 16 has a structure capable of controlling the voltage of the transistor 21. When applied to the circuit of FIG. 1, it becomes possible to control the peak value of the voltage of the switching element 3 according to changes in the current of the load circuit 10 and the voltage Ei of the DC power supply 1, and the range of the input voltage and the load current. The peak voltage of the switching element 3 can be minimized even when the voltage is wide.

As described above, by using a diode having a carrier lifetime equal to or longer than the switching period as the diode 13, a desired resonance operation can be obtained without increasing the switching element voltage, which is a feature of the present invention. You can

When a diode having a carrier lifetime shorter than the switching cycle is used as the diode 13, the resonance energy stored in the transformer 2 during the ON period of the switching element 3 is used for voltage clamping due to the forward conduction of the diode 13. When it is transmitted to the capacitor 14, the electrons and holes are immediately coupled at the junction of the diode 13 and most of the charges are lost, and the reverse characteristic is immediately restored. Resonance energy recovery from the clamping capacitor 14 to the transformer 2 is very small. Therefore, most of the resonance energy transmitted to the voltage clamping capacitor 14 needs to be discharged by the discharge circuit 15, which increases the power loss of the discharge circuit 15 and increases the current capacity of the components forming the discharge circuit 15. Inevitable.

When a diode having a carrier lifetime shorter than the switching cycle is used as the diode 13 and the discharge energy of the discharge circuit 15 is reduced, the voltage clamping capacitor 14 and the switching element 3 must have a high breakdown voltage. The problem also arises.

As described above, in this embodiment, both the switching element 13 and the rectifying diode 6 can be turned on / off at zero voltage without increasing the voltage of the resonance capacitor 4 as in the conventional case. Further, the resonance capacitor 4 can be omitted regardless of the conversion frequency, and the resonance capacitance can be satisfied only by the output capacitance including the junction capacitor of the switching element 3.

Therefore, the resonance current of this circuit is as small as the exciting current of the transformer of the conventional non-resonant forward converter, so that the switching element, the winding current of the transformer, and the rectifying diode. -The current of the switching device is the same as that of the conventional non-resonant forward converter, and there is almost no increase in the circuit current for realizing zero-voltage switching over a wide range of load currents. It is possible to make a high-efficiency converter at high frequency without making it higher than the circuit.

Next, another embodiment of the present invention will be described with reference to FIG. 5. The same symbols as those shown in FIG. 1 indicate the corresponding members, and in this embodiment, in the embodiment shown in FIG. The resonance capacitor 11 on the secondary side is omitted,
The rectifying diode 6 does not turn off at zero voltage when turning off. The typical operation waveform is shown in FIG. In this embodiment, the junction capacitance of the rectifying diode 6 is small, and if the switching element 3 is switched to zero voltage, it can be operated at a sufficiently high conversion frequency. 1 differs from the embodiment of FIG. 1 in that the secondary side circuit does not resonate, but the main operation of switching the primary side switching element to zero voltage without increasing the voltage of the resonance capacitor, which is a feature of the present invention. 1 is almost the same as that of the embodiment shown in FIG.

[0048]

As described above, in the present invention, the series circuit of the diode 13 and the voltage clamping capacitor 14 having the carrier lifetime equal to or longer than the switching period is used as the switching element on the primary side of the conventional resonant converter. By connecting in parallel, while achieving the desired zero-voltage switching operation, it is possible to use a switching element with a current that is substantially the same as that of the conventional non-resonant type forward converter and that is considerably lower than that of the conventional circuit. It is possible to achieve a very large practical effect such as reducing power loss.

[Brief description of the drawings]

FIG. 1 is a diagram showing a first embodiment of a resonance type forward converter according to the present invention.

FIG. 2 is a diagram showing a waveform of each part for explaining the first embodiment.

FIG. 3 is a diagram showing a current recovery rate with respect to a recovery time of a diode having a recovery time capable of flowing a reverse current of the same amount as a forward current used in the present invention.

FIG. 4 is a diagram showing an embodiment of a discharge circuit used in the present invention.

FIG. 5 is a diagram showing a second embodiment of the resonance type forward converter according to the present invention.

FIG. 6 is a diagram showing a waveform of each part of the second embodiment.

FIG. 7 is a diagram showing characteristics of a saturable inductor used in the present invention.

FIG. 8 is a diagram showing an example of a conventional resonant converter.

FIG. 9 is a diagram showing a waveform of each part for explaining a conventional resonant converter.

[Explanation of symbols]

1 ... DC power supply 2 ... Transformer having primary winding 2A and secondary winding 2B 3 ... Switching element 4 ... Resonance capacitor 5 ... Saturable Inductor 6 ・ ・ ・ ・ Rectifying diode 7 ・ ・ ・ ・ ・ ・ Freewheeling diode 8 ・ ・ ・ ・ Smoothing inductor 9 ・ ・ ・ ・ Smoothing capacitor 10, 10 '・ ・ ・ DC output terminal 11 ・ ・ ・ ・ ・ ・ Resonance Capacitors 11-a, 11-b ... Discharge circuit terminal 12 ... Control circuit 13 ... Diode 14 ... Voltage clamp capacitor 15 ... Discharge circuit 16 ... Control circuit 17 ... Resistor 18 ... Resistor 19 ... Resistor 20 ... Zener diode 21 ... Transistor

Claims (9)

[Claims] 1. A transformer having a DC power supply, a primary winding connected in series to the DC power supply, and a secondary winding magnetically coupled to the primary winding; A saturable inductor that is connected in series with the next winding and that has a predetermined inductance in the low power region and exhibits magnetic saturation when applied to a predetermined voltage integration to become a small inductance, and is connected in series to the saturable inductor. A switching element, a resonance capacitor connected in parallel with the switching element, a rectifying diode connected in series with one terminal of the secondary winding of the transformer, the rectifying diode, and the rectifying diode. A freewheeling diode connected across the other terminal of the secondary winding of the transformer and an output filter, and a control circuit for applying a control signal to the switching element to control its conduction. Resonant type forward equipped In a diode converter, a resonance circuit characterized in that a series circuit of a diode having a carrier lifetime equal to or longer than the switching period of the switching element and a voltage clamping capacitor is equivalently connected in parallel to the switching element. Type forward converter. 2. The resonance type forward converter according to claim 1, wherein the resonance capacitor is an output capacitance of the switching element. 3. The resonance type forward converter according to claim 1, wherein the resonance capacitance comprises an output capacitance of the switching element and a capacitance of a resonance capacitor connected in parallel with the switching element. . 4. The saturable inductor comprises a first winding wound on a core of rectangular hysteresis and a linear inductor connected in parallel with the first winding. The described resonance type forward converter. 5. The saturable inductor comprises a first winding wound around a core of rectangular hysteresis, a second winding magnetically coupled to the first winding, and a second winding connected to the second winding. The resonance type forward converter according to claim 1, wherein the resonance type forward converter comprises a linear inductor connected in parallel. 6. The saturable inductor is parallel to a first winding wound around a core of rectangular hysteresis, a second winding magnetically coupled to the first winding, and the second winding. The resonance type forward converter according to claim 1, wherein the resonance type forward converter is constituted by a third winding which is connected and is not magnetically coupled to the first winding. 7. A discharge circuit for discharging, of the energy stored in the voltage clamping capacitor, energy that could not be returned to a DC power source through the transformer during reverse conduction of the diode, 2. The resonance type forward converter according to claim 1, wherein the diode and the voltage clamping capacitor are connected across the junction and one end of the DC power source. 8. The resonant forwarder of claim 7, wherein the discharge circuit exhibits a variable impedance.
DO converter. 9. The control circuit detects that the reverse current of the diode becomes zero, or detects the zero voltage or the minimum voltage of the switching element, and detects the reverse direction of the diode. The resonance type forward converter according to claim 1, wherein a turn-on signal is output to the switching element after the conduction is completed.