JPH06165510A - Inverter - Google Patents

Abstract

PURPOSE:To suppress unneeded resonance current and then to allow needed resonance to flow as needed by connecting the primary coil winding of a current transformer between the load connection point of a diode of a first diode series body and a load and then connecting the secondary coil winding of the current transformer to both terminals of a capacitor for auxiliary resonance. CONSTITUTION:A second diode series connection body where diodes 11 and 12 are connected in series is connected to a DC power supply 1 in opposite direction and a capacitor 13 for auxiliary resonance is connected between the connection points of each diode of the first and second diode series connection bodies. Further, a load is taken out of the connection point of diodes 71 and 72 of the first diode series connection body. However, the primary coil winding of a current transformer 10 is inserted into an area in reference to the load and then is connected to both terminals of the auxiliary resonance capacitor 13 so that the secondary coil winding of the current transformer 10 is in the polarity shown in figure, thus suppressing the peak value of resonance current to be small and hence reducing the current of the diode 71, a self arc- extinguishing type switching element 21, and a resonance reactor 9.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter devised so as to sufficiently suppress switching loss which occurs when a self-turn-off switching element is turned on and off.

[0002]

2. Description of the Related Art A typical basic circuit of a conventional inverter is shown in FIG. Reference numeral 1 is a DC power supply, and 21 and 22 are self-extinguishing type switching elements, which are connected in series and connected to the DC power supply 1. Diodes 31 and 32 are connected in anti-parallel to the self-extinguishing switching elements 21 and 22, respectively, and a series connection body of snubber diodes 41 and 42 and snubber capacitors 51 and 52 is connected in parallel. Snubber resistors 61 and 62 are connected in parallel to 41 and 42, respectively. The load is taken from the series connection point of the self-extinguishing type switching elements 21 and 22. The self-extinguishing switching elements 21 and 22 alternately turn on and off to supply AC power to the load.

In this circuit, when the load current is flowing to the self-extinguishing type switching element 21, when the element is turned off, the load current is commutated to the series circuit of the snubber diode 41 and the snubber capacitor 51, and the snubber capacitor 51. By this action, no voltage is immediately applied to the self-extinguishing type switching element 21, and the switching loss can be made extremely small. When the load current charges the snubber capacitor 51 to the DC power supply voltage, the load current starts to flow in the diode 32, and the self-extinguishing switching element 21 turns off after a predetermined time from turning off the self-extinguishing switching element 21.
No current flows when 22 is turned on. In the PWM controlled inverter, the self-extinguishing switching element 22
Is frequently turned on again and the self-extinguishing switching element 21 is turned on again.However, when the self-extinguishing switching element 21 is turned on, the charge accumulated in the diode 32 in which the current flows is Until it is discharged, the self-extinguishing type switching element 21 and the diode are in a state close to a short circuit.
A large current flows through 32. Immediately after the diode 32 reversely recovers, a large current continues to flow in the snubber diode 42 and the snubber capacitor 52 in a state close to a short circuit. These large currents cause large losses in the self-extinguishing switching element 21 and the diode 32.

At this time, a self-extinguishing type switching element
When 21 is turned on, the electric charge charged to the voltage E of the DC power supply 1 in the snubber capacitor 41 is discharged via the snubber resistor 61. This causes a loss in the snubber resistor 61 and the self-extinguishing type switching element 21. If the capacitance of the snubber capacitor is C _{s and the} operating frequency is f, this loss P becomes P = (1/2) C _s E ² × f, which increases in proportion to the operating frequency, and at high frequencies (1 kHz or more) It becomes a big problem.

As described above, in the conventional inverter, the high frequency operation is difficult due to the switching loss of the self-turn-off switching element and the loss due to the consumption of the electrostatic energy accumulated in the snubber capacitor. Various proposals have been made to solve these problems. For example, various methods have been proposed in which a part or all of the electrostatic energy stored in the snubber capacitor is regenerated to a DC power source via a power conversion circuit, but all of the circuits are complicated and are in practical use. Almost never.

[0006] In addition, "IEEE T"
RASACTION ON POWER ELECTRONICS "Vol.7 No.2 The paper" A Rugged Soft "described in p385-p392 of April 1992 issue.
"Com-mutated PWM Inverter for AC Drive" uses the minimum required number of two switching elements to compose one phase of the inverter, and also enables operation with switching loss suppressed as much as possible both on and off. An inverter circuit has been introduced.This circuit is shown in Fig. 4. The same reference numerals as in Fig. 3 denote parts having the same function. In this circuit, a snubber circuit connected to the self-extinguishing switching elements 21 and 22. Is only the snubber capacitors 51 and 52, which is a so-called lossless snubber circuit, which is simpler than that in Fig. 3. Further, the capacitors 71 and 72 in which the resonant capacitors 81 and 82 are connected in parallel are connected in series to the power supply. 1 in the reverse direction, connect the resonance reactor 9 between the series connection point of the self-extinguishing switching elements 21, 22 and the series connection point of the diodes 71, 72, and connect the diodes 71, 72 in series. So that take out the load from the connection point.

The operation of this circuit will be briefly described. When the self-extinguishing switching element 22 is in the ON state and the load current i ₀ is flowing in the direction shown by the arrow in FIG. i ₀ all flows from the diode 72, and the current due to the residual magnetic energy of the resonance reactor 9 flows in the direction opposite to the arrow of i _L shown in FIG. 7, and the self-extinguishing switching element 22 → diode 72 → resonance reactor 9 route (Refer to the operation to be described later for such a state). In this state, when the self-extinguishing type switching element 22 is turned off by the gate signal under the PWM control, the return current (-i _L )
Is commutated to the snubber capacitors 51 and 52, and no voltage is immediately applied to the self-extinguishing type switching element 22, that is, so-called zero voltage switching (ZVS), and turn-off loss can be extremely small.
The current -i _L is discharged and charged by the snubber capacitors 51 and 52 and the resonance reactor 9 by the resonance of the snubber capacitors 51 and 52, and the resonance is stopped when the respective voltages become zero and the DC power supply voltage E. At this time, there is residual magnetic energy in the resonance reactor 9, and the current due to this flows through the route of diode 31 → DC power supply 1 → diode 72 → resonance reactor 9 and is regenerated to the DC power supply 1. If the self-extinguishing type switching element 21 is turned on while the current is flowing through the diode 31, the current does not immediately flow out, that is, so-called zero current switching (ZCS), and no turn-on loss occurs.

Next, when the electrostatic energy of the resonance reactor 9 is released and the current becomes zero, the DC power supply 1 → self-extinguishing type switching element 21 → resonance reactor 9 contrary to the conventional case.
In → diode 72 → DC power source 1 route, for the DC power supply voltage E is applied to the resonant reactor 9, current i _L is gradually increased in the direction of the arrow in FIG. 4, it becomes equal to the load current i ₀ diode 72 It becomes non-conductive. From this moment, the resonance operation by the resonance reactor 9 and the resonance capacitors 81, 82 starts, the resonance capacitors 81, 82 are discharged and charged respectively, and become 0 and the DC power supply voltage E, respectively, and the resonance operation is stopped.
Thereafter, due to the residual magnetic energy of the resonant reactor 9, a return current flows in the route of the resonant reactor 9 → diode 71 → self-extinguishing type switching element 21. That is, the load current i ₀ and the resonance current having the peak value I _r flow in the self-extinguishing type switching element 21 and the resonance reactor 9 in a superimposed manner. I _r = E (2C _r / L) ^1/2 where C _r is the capacitance of the resonant capacitors 81 and 82, and L is the inductance of the resonant reactor 9.

Next, when the self-extinguishing type switching element 21 is turned off by the PWM signal, the current flowing in the resonance reactor 9 is commutated to the snubber capacitors 51 and 52, and the resonance operation between the capacitors and the resonance reactor starts. The self-extinguishing type switching device 21 can be turned off by the snubber capacitor 51 by ZVS described above without the occurrence of switching off loss. The resonance operation is stopped when the voltages of the snubber capacitors 51 and 52 become the DC power supply voltage E and zero. It was flowing in the resonance reactor 9 (load current +
A large current (resonance current) is regenerated to the DC power supply 1 through the route of the diode 32 → resonance reactor 9 → diode 71 → DC power supply 1. Here, if the self-extinguishing type switching element 22 is turned on while the current is flowing through the diode 32, Z
It becomes CS and no turn-on loss occurs. When the current i _L of the resonance reactor 9 decreases due to the release of the residual magnetic energy of the resonance reactor 9 and becomes equal to the load current I ₀ , the diode 71 becomes non-conducting, and the resonance reactor 9 and the resonance capacitors 81, 82 are connected. Resonance starts again.
Resonance continues until the resonant capacitors 81 and 82 are at the DC power supply voltage E and zero, respectively.

At this time, if the circuit constant is determined so that the peak value I _r of the resonance current becomes larger than the load current I ₀ ,
The current i _L of the resonant reactor 9 can be negative and can flow in the direction opposite to the arrow shown in FIG. That is, _assuming that the inductance of the resonance reactor 9 is L and the capacitance of the resonance capacitors 81 and 82 is C _r , i _L = I _0MAX −E (2C _r / L) ^1/2 <0 ∴ (2C _r / L) ^1/2 > I _0MAX / E Here, I _0MAX is selected as a constant so as to satisfy the condition of the expected maximum value of the load current.

As a result, the current due to the residual magnetic energy of the resonant reactor 9 after resonance recirculates through the route of the self-extinguishing type switching element 22 → diode 72 → resonant reactor 9 and becomes the first state of this explanation. Is repeated. The same operation is performed when the load current has a polarity opposite to that described above, and thus the description thereof will be omitted.

The above description is based on the current i of the resonance reactor 9.
FIG. 5 shows a time chart in which _L and the states of the self-extinguishing type switching elements 21 and 22 are associated with each other. In FIG. 5, the resonance operation between the resonance reactor 9 and the snubber capacitors 51 and 52 is omitted because the period is short. As shown in FIG. 5, the peak value I _{r of the} resonance current occurring during the period T ₅ is _set to be larger than the maximum value I _{0MAX of the} load current, and the resonance reactor current i _L is set to cross the zero line. For this reason, the peak value of the resonance current occurring in the period T ₂ is also I _r, and a large current (I _0MAX + I _r ) flows through the self-arc-extinguishing type switching element 21. In the case of FIG. 5, it is necessary to flow more than twice the load current I _0MAX into the self-extinguishing type switching element 21 (2.5 times or more is optimum in the above-mentioned paper).

As described above, this conventional example realizes an inverter in which the switching loss of the self-arc-extinguishing type switching element is suppressed and the loss due to the snubber resistance is eliminated. A large current flows during steady conduction, and the conduction loss of the self-extinguishing type switching element due to this large current increases, which hinders the realization of a large capacity inverter. Also, the load current i ₀
It is also a drawback of this circuit that a large resonance current flows even when is small.

[0014]

In the above-mentioned conventional inverter, it is possible to suppress the generation of switching loss when the self-arc-extinguishing switching element is switched on / off. There is a problem that the steady conduction loss increases. The present invention has been made in order to solve the above problems, and suppresses unnecessary resonance current (T ₂ period in FIG. 5) as much as possible and requires necessary resonance current (T ₅ period in FIG. 5). The purpose is to let as much as possible.

[0015]

A second diode series body in which two sets of diodes are connected in series is connected to a DC power supply 1 in the reverse direction, and the respective diodes of the first and second diode series bodies are connected in series. An auxiliary resonance capacitor is connected between the connection points, and a primary winding of the current transformer is connected between the series connection point of the diodes of the first diode series body and the load, and the secondary winding of the current transformer is connected. The winding is connected to both ends of the auxiliary resonance capacitor.

[0016]

In the case of the load current polarity shown in FIG. 5, the period T ₂
At resonance, the capacitance of the resonance capacitors connected in parallel to the diodes of the first diode series connection body is equivalently reduced, and the resonance current peak value [E (2C _r
/ L) ^1/2 ] is kept small, the capacitance of the resonance capacitor is equivalently increased at the time of resonance in the period T ₅ , and the peak value of the resonance current is increased, thereby eliminating the drawback of the method shown in FIG. it can.

[0017]

1 is a circuit diagram showing an inverter according to an embodiment of the present invention. The difference from the conventional example is that diode 1
A second diode series connection body in which 1, 12 are connected in series is connected in the reverse direction to the DC power source 1, and an auxiliary resonance capacitor 13 is provided between the connection points of the diodes of the first and second diode series connection bodies. Is the point that is connected. Further, the load is taken out from the connection point between the diodes 71 and 72 of the first diode series connection body, but the primary winding of the current transformer 10 is inserted between the load and the secondary winding of the current transformer 10. The point is that the line has the polarity shown in FIG. 1 and is connected to both ends of the auxiliary resonance capacitor 13.

The operation of the circuit shown in FIG. 1 will be described with reference to the time chart of FIG. At time t ₀ , the self-extinguishing switching element 22 is turned off, and before the polarity of the current i _L of the resonant reactor 9 becomes positive, the self-extinguishing switching element 21 is turned on to add the current i _{L of the} resonant reactor to positive. Although the operation of increasing to the side is exactly the same as the conventional example, the current i _L becomes equal to the load current i ₀ at time t ₁ , and the resonance between the resonance reactor 9 and the resonance capacitors 81 and 82 starts. Since the capacitance C _r of the resonance capacitors 81 and 82 is set to be much smaller than that in the case of FIG. 4, the resonance current [I _r ′ = E (2C _r /
L) ^1/2 ] and does not grow. At this time, the auxiliary resonance capacitor 13 conducts the diode 72 until resonance starts,
Moreover, since the secondary winding current i _c of the commutation transformer 10 does not flow to the auxiliary resonance capacitor 13 due to the action of the diode 12, the voltage of the auxiliary resonance capacitor 13 is zero volt.

Therefore, the auxiliary resonance capacitor 13 has no influence on the resonance operation starting from time t ₂ . When the resonance current reaches the peak, the current due to the residual magnetic energy of the resonance reactor 9 flows back through the path of the diode 71 → self-extinguishing type switching element 21 → resonance reactor 9,
The load current flows through the self-extinguishing switching element 21 → resonance reactor 9 → the primary winding of the current transformer 10. When resonance starts, the diode 72 becomes non-conducting and the resonance capacitor
82 is charged. At the same time, the secondary current i _c of the current transformer 10 flows into the auxiliary resonance capacitor 13, but the voltage increase of the auxiliary resonance capacitor 13 is slower than the voltage increase of the resonance capacitor 82. Has almost no effect.

The charging voltage of the auxiliary resonance capacitor 13 is substantially determined by the magnitude of the load current and the conduction time of the diode 71, but due to the action of the diodes 71, 72 and 11, 12,
It does not exceed the DC power supply voltage E. Period T ₂ as described above
Since the peak value of the resonance current that occurs in
The currents of the diode 71, the self-extinguishing type switching element 21, and the resonant reactor 9 can be greatly reduced as compared with the method of FIG.

Further, time t ₃ self-extinguishing type switching element
21 is turned off, the current i _L of the resonance reactor 9 becomes equal to the load current i ₀ at time t ₄ , and resonance starts again. At this time, the auxiliary resonance capacitor 13 is charged to the polarity shown in FIG. 1 ( In the following description, it is assumed that the resonance power is charged to the DC power supply voltage E. However, even if the load is small and the DC power supply voltage E is not charged, the resonance operation is not greatly affected. greatly influence the resonant peak current I _r "is I _r" = E _{[(2C r + C a) /} L ] ^1/2, compared with the method of FIG. 4 _{[(2C r + C a) /} 2
C _r ] ^1/2 . Where C _A is the auxiliary resonant capacitor
The capacitance is 13, and the resonance current can be significantly increased by increasing C _A , and the resonance reactor current i _L can be set to the negative side.

As a result, the current in the resonant reactor 9 is reversed in polarity as in the conventional example, and the self-extinguishing type switching element 22
A return current can be supplied to the device, and the device can be operated without impairing the conventional features.

The self-extinguishing type switching element in each of the above-mentioned embodiments has the same effect as long as it is any self-extinguishing type switching element.

[0024]

As described above, according to the present invention, the number of self-extinguishing switching elements constitutes an inverter and the switching loss generated when the self-extinguishing switching elements are turned on / off is suppressed. It is possible to provide a very excellent inverter that has a small amount of loss even if it is operated at a high frequency because the current that is slightly higher than the load current only needs to flow through the self-turn-off switching element.

[Brief description of drawings]

FIG. 1 is a circuit diagram of an embodiment of an inverter according to the present invention.

FIG. 2 is a time chart for explaining the operation of the embodiment.

FIG. 3 is a circuit diagram showing a conventional typical inverter.

FIG. 4 is a circuit diagram of a conventional inverter that minimizes switching loss.

5 is a time chart for explaining the operation of the embodiment of FIG.

[Explanation of symbols]

 1 DC power supply 9 Resonance reactor 10 Current transformer 11, 12 Diode 13 Auxiliary resonance capacitor 21, 22 Self-extinguishing type switching element 31, 32 Diode 41, 42 Snubber diode 51, 52 Snubber capacitor 61, 62 Snubber resistance 71, 72 Diode 81, 82 Resonant capacitor

Claims (1)

[Claims] 1. A diode in which two sets of self-extinguishing type switching elements in which a diode and a snubber capacitor are respectively connected in parallel are connected in series to a DC power source in the forward direction and a resonance capacitor is connected in parallel. An inverter circuit in which two sets of first diode series bodies connected in series are connected in the opposite direction, and a resonance reactor is connected between the series connection point of the self-turn-off switching element and the series connection point of the diode. , 2nd series connection of two diodes for DC power supply
Connecting the diode series bodies in the opposite direction, and connecting an auxiliary resonance capacitor between the series connection points of the diodes of the first and second diode series bodies, and connecting the diodes of the first diode series body in series. An inverter characterized in that a primary winding of a current transformer is connected between a connection point and a load, and secondary windings of the current transformer are connected to both ends of the auxiliary resonance capacitor.