JPH0847179A - Charger - Google Patents

Abstract

PURPOSE:To provide a charger including a rectifier for preventing a G-pulse voltage on the generator side from flowing to a battery side when the current from a generator is dropped abruptly. CONSTITUTION:A p-type substrate region just under a gate electrode of high-side switches 31 to 33 is grounded. When a channel of the high-side switches 31 to 33 is turned off at the generation of a G-pulse, and the G-pulse transmitted through a second parasitic diode D2 and the channel of high-side switches 31 to 33 to a load 9 can be prevented from being outputted. At the same time, when the low-side switches 34 to 36 are turned on, armature windings 11 to 13 are short-circuited through the channel resistance of the low-side switches 34 to 36 so that the G-pulse voltage can be reduced.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a charging device having a rectifier using a MOS transistor. INDUSTRIAL APPLICABILITY The charging device of the present invention is applied to, for example, a charging device that charges a battery by rectifying an AC power generation voltage of an alternator (alternator) for a vehicle.

[0002]

2. Description of the Related Art Japanese Unexamined Patent Application Publication No. 4-138030 has disclosed in
As shown in, a three-phase full-wave rectifier using a MOS power transistor is proposed. This three-phase full-wave rectifier 3 has M
High-side switches 31 to 3 including OS transistors
3 and low-side switches 34 to 36 are individually connected in series, and three sets of phase inverter circuits 37 to 39 are connected in parallel, and a pair of DC power terminals are individually connected to the high-level end and the low-level end of the battery 7. , Each phase inverter circuit 37-3
The connection points 41 to 43 of the nine switches, that is, the AC input terminals are individually connected to the phase output terminals of the armature windings 11 to 13 of the AC generator 1. Reference numeral 2 is a field coil, 4 is a field current control transistor, 5 is a flywheel diode, and 6 is a controller.

Also, in this prior art embodiment, the NMO
Low-side switches 34 to 36 composed of S transistors
The main electrode (shown as S) on the battery low-potential terminal side is connected to a P-type region (referred to as a substrate region) immediately below the channel to apply a potential to this substrate region, and similarly, high-side switches 31 to 33 composed of NMOS transistors are connected. Main electrode (shown as S) on the stator coil side of P
It is connected to a mold region (referred to as a substrate region) and a potential is applied to this substrate region.

Further, Japanese Patent Laid-Open No. 4-2 filed by the present applicant.
Japanese Patent No. 10739 proposes a three-phase full-wave rectifier using a MOS power transistor in order to realize a simplified structure and a brushless generator in a generator having a field pole composed of a permanent magnet as shown in FIG. are doing.

[0005]

In the conventional diode rectification type three-phase full-wave rectifier or the MOS transistor type three-phase full-wave rectifier 3, the charging line 8 of the battery 7 is disconnected, or the overcurrent of the vehicle load 9 is exceeded. When the protective fuse 10 is blown or the output current of the three-phase full-wave rectifier is suddenly reduced, the transistor 4 is turned off in order to reduce the field current, but the field coil 2 has a flywheel diode. In order to continue refluxing for 5 for a predetermined time,
Also, since a large amount of pulse voltage (hereinafter referred to as G pulse voltage) is generated because the magnetic energy accumulated in the magnetic circuit in which the armature windings 11 to 13 are interlinked, is generated as an induced voltage in the armature windings 11 to 13. It is applied to the vehicle load 9 through the channels of the MOS transistors 31 to 36 or the parasitic diode.

Therefore, conventionally, there has been a problem that an important vehicle load must be provided with a surge countermeasure, that is, a G pulse voltage blocking countermeasure, respectively. The present invention has been made in view of the above circumstances, and the generated current suddenly decreases. It is an object of the present invention to provide a charging device equipped with a rectifier capable of preventing the G pulse voltage generated on the generator side from being sent to the battery side.

Further, when the MOS transistor type three-phase full-wave rectifier proposed by Japanese Patent Laid-Open No. 4-138030 is applied to a magnet-excited AC generator, the following problems occur. That is, each M that constitutes the high side switches 31 to 33.
Since the OS transistor has a parasitic (junction) diode having an N-type region on the battery side, when the phase voltage exceeds the battery voltage and the parasitic diode is forward biased, the battery is charged through this diode, and as a result, Even if the high-side switches 31 to 33 are intermittently controlled, the battery voltage cannot be controlled. In the field coil excitation type AC generator, since the field current can be controlled, such a problem does not occur.

Further, the MOS transistor type three-phase full-wave rectifier proposed by Japanese Patent Laid-Open No. 4-210739 also has the following problems. That is, the P-type well region of the N-channel MOS transistor that constitutes the high-side switch of this three-phase full-wave rectifier is short-circuited to the N-type region (one of the main electrode pair) on the battery side, so that the high-side switch is eventually turned on. A parasitic (junction) diode having a P-type region on the battery side exists in parallel with each of the N-channel MOS transistors that are configured, and when the phase voltage exceeds the battery voltage, the parasitic diode is reverse-biased. When the phase voltage is less than the battery voltage, this parasitic diode is forward biased, which is equivalently equivalent to the case where the high side switch is turned on. Therefore, it is necessary to connect the backflow prevention diode in series with the high-side switch and the low-side switch, which causes problems such as complication of the configuration and increase of power loss.

The present invention has been made in view of the above problems, and has the above-mentioned power loss and configuration when a MOS transistor type three-phase full-wave rectifier is applied to a magnet excitation type AC generator that does not perform excitation current control. Another object of the present invention is to provide a charging device that can avoid the problem of complication of the above.

[0010]

A first aspect of the charging device of the present invention.
In the above configuration, a required number of phase inverter circuits in which high-side switches and low-side switches composed of N-channel MOS transistors are connected in series are connected in parallel, and a pair of DC output terminals are individually connected to both ends of a battery and a load. A rectifier whose connection point is connected to each phase output terminal of the armature winding of the alternator, and a control unit that controls the gate potential of each switch to connect and disconnect each switch. In the charging device provided, a substrate region formed of a P-type region immediately below the gate electrode of the high side switch is connected to the low end of the battery.

According to a second structure of the present invention, in addition to the above-mentioned first structure, the control section detects the phase voltage at each phase output end of the AC generator, and any one of the phase voltages is detected. It is characterized by having a surge voltage cutoff mode for cutting off at least the high side switch group when the voltage exceeds a predetermined voltage value. A third configuration of the present invention is characterized in that, in the above second configuration, the control unit continues the surge voltage cutoff mode for a predetermined time.

According to a fourth structure of the present invention, in addition to the second structure, the control section maintains the surge voltage cutoff mode while any one of the phase voltages exceeds a predetermined voltage value. It is characterized by being. Fifth of the present invention
In the second configuration, the control unit may further include the phase voltage applied to an N-type armature winding side region that forms a main electrode on a connection point side of the high side switches. It is characterized in that the high side switch having a voltage higher than the voltage of the battery is turned on.

According to a sixth aspect of the present invention, in addition to the above-mentioned second configuration, the control section applies a voltage to an N-type armature winding side region forming a main electrode on a connection point side among the low side switches. It is characterized in that the low-side switch whose ON phase voltage is lower than the voltage of the battery is turned on. In a seventh configuration according to the present invention, in addition to the second configuration, the first parasitic diode between the substrate region and the N-type battery side region forming the battery-side main electrode of the high-side switch further comprises: It is characterized by having a withstand voltage exceeding the maximum rated voltage value of the battery.

An eighth configuration of the present invention is the same as the second configuration, further including a first region between the N-type armature winding side region forming the main electrode on the connection point side of the high side switch and the substrate region. 2 Parasitic diode is a predetermined G that exceeds the maximum generated voltage.
It is characterized by having a breakdown voltage exceeding a pulse voltage value. A ninth configuration of the present invention is the second configuration, further including a third parasitic diode between the N-type armature winding side region forming the main electrode on the connection point side of the low side switch and the substrate region. It is characterized by having a withstand voltage exceeding a predetermined G pulse voltage value exceeding the maximum power generation voltage.

A tenth structure of the present invention is the same as the seventh structure, wherein the MO forming the high side switch is further added.
One of the both N-type regions forming the main electrode of the S transistor has an N ⁻ type withstand voltage layer, and the N ⁻ type withstand voltage layer is formed on the first parasitic diode side. According to an eleventh structure of the present invention, in addition to the eighth structure, one of both N-type regions forming a main electrode of the MOS transistor forming the high side switch has an N ⁻ type withstand voltage layer, The N ⁻ type breakdown voltage layer is formed on the side of the second parasitic diode.

A twelfth structure of the present invention is the same as the ninth structure, further including the MO switch forming the low side switch.
One of the both N-type regions forming the main electrode of the S transistor has an N ⁻ type withstand voltage layer, and the N ⁻ type withstand voltage layer is formed on the third parasitic diode side. According to a thirteenth configuration of the present invention, in addition to the second configuration, the high-side switch and the low-side switch forming the phase inverter circuit of the same phase are integrated on the same chip having an N-type region as a substrate. The substrate region is connected to the phase output terminal of the AC generator.

According to a fourteenth structure of the present invention, in addition to the above-mentioned second structure, the control section turns off all the high side switches and turns on the low side switch when the surge voltage cutoff mode is executed. It is characterized by being. A fifteenth configuration of the present invention is configured such that a required number of phase inverter circuits each including a high-side switch and a low-side switch made of N-channel MOS transistors connected in series are connected in parallel, and a pair of DC output terminals are provided at both ends of a battery and a load. A rectifier whose connection point is connected to each phase output terminal of the armature winding of the alternator, and a control for connecting and disconnecting each switch by controlling the gate potential of each switch. A substrate region formed of a P-type region immediately below the gate electrode of the high-side switch is connected to the N-type region on the armature winding side to be given a potential, and a P-type region directly below the gate electrode of the low-side switch. In a charging device in which a substrate region formed by a region is connected to an N-type region on the lower terminal side of the battery to apply a potential, the control unit includes the armature winding. When the generated voltage exceeds a predetermined voltage value, one of the low-side switches and all the high-side switches is turned on, and the other group is turned off to implement a surge voltage cutoff mode. I am trying.

In a sixteenth structure of the present invention, in addition to the fourteenth or fifteenth structure, all the high side switches or all the low side switches in which the control section is turned on at the same time when the surge voltage cutoff mode is executed. The average gate voltage is determined based on the maximum allowable current value of each switch. A seventeenth configuration of the present invention is the same as the first configuration, wherein the AC generator is a magnet-excited AC generator, and the controller controls the phase voltage of each phase output terminal of the AC generator and the The voltage of the battery is detected, and the phase voltage applied to the N-type armature winding side region forming the main electrode on the connection point side among the high side switches is higher than the voltage of the battery, and While the battery voltage is below the specified voltage value,
It is characterized in that the high side switch is turned on.

In an eighteenth configuration of the present invention, in addition to the above seventeenth configuration, the control section turns off the high side switch while the voltage of the battery exceeds a predetermined voltage value, and at least the high side switch. The low side switch connected to the same phase end as the side switch is turned on. A nineteenth structure of the present invention is the same as the seventeenth structure, wherein the high side switch and the low side switch forming the phase inverter circuit of the same phase are integrated on the same chip having an N type region as a substrate. The substrate region is connected to the phase output terminal of the AC generator.

The twentieth structure of the present invention is an N-channel M
A required number of phase inverter circuits in which a high-side switch and a low-side switch composed of OS transistors are connected in series are connected in parallel, a pair of DC output terminals are connected to both ends of a battery and a load, and a connection point of both switches is connected. A gate of the high-side switch, comprising a rectifier that is individually connected to each phase output terminal of the armature winding of the AC generator, and a control unit that controls the gate potential of each switch to connect and disconnect each switch. A substrate region consisting of a P-type region directly under the electrode is connected to an N-type region on the side of the armature winding and a potential is applied, and a substrate region consisting of a P-type region directly under the gate electrode of the low-side switch is a low terminal of the battery. N on the side
In a charging device that is connected to the mold region and is given a potential,
The AC generator comprises a magnet excitation type AC generator, the control unit detects a phase voltage at each phase output end of the AC generator and a voltage of the battery, and any one of the phase voltages is Turning on the high-side switch while the voltage of the battery is higher than the voltage of the battery and the voltage of the battery is less than a predetermined first voltage, and further turning on all the low-side switches while the high-side switch is off. Is characterized by.

[0021]

According to the first structure of the present invention,
Since the substrate region consisting of the P-type region immediately below the gate electrode of the high-side switch is connected to the low end of the battery, the N-type armature winding side region and the substrate region (main electrode on the connection point side of the high-side switch) and the substrate region ( The second parasitic diode formed by the PN junction with the semiconductor region immediately below the gate electrode) has a G pulse voltage generated in the armature winding, which will be described later.
Since it is prevented from being applied to the load through the parasitic diode, it is not necessary to take measures to protect each load from the G pulse voltage. The first parasitic diode is a parasitic diode formed by a PN junction between an N-type battery side region that forms a battery side main electrode of the high side switch and a substrate region (semiconductor region immediately below the gate electrode). .

According to the second structure of the present invention, in the first structure, at least the high side switch group is provided when at least one of the phase voltages at the phase output terminals of the AC generator exceeds a predetermined voltage value. Since the G pulse voltage is blocked, the G pulse voltage is prevented from being applied to the load through the channel region formed in the high-side switch during the ON operation, so that it is not necessary to take measures to protect each load from the G pulse voltage.

According to the third structure of the present invention, in the second structure, the high-side switch group is shut off for a predetermined time sufficient to decrease the surge voltage from the time when the surge voltage is detected. The residual surge voltage is not applied to the load when the side switch is turned on again.
According to the fourth configuration of the present invention, in the second configuration, the high-side switch group is turned off while any of the phase voltages at the phase output terminals of the alternator exceeds a predetermined voltage value. , The G pulse voltage can be surely blocked regardless of the G pulse voltage duration.

According to the fifth configuration of the present invention, in the second configuration, the high side switch is turned on when the phase voltage applied to the N-type armature winding side region is higher than the battery voltage. If not, there is no backflow of current from the battery to the armature winding. In the sixth configuration of the present invention, further, in the second configuration, the low-side switch in which the phase voltage applied to the N-type armature winding side region is lower than the battery voltage is turned on, and is turned off otherwise. No reverse current flow from the battery to the armature winding.

In a seventh configuration of the present invention, in addition to the second configuration, the first parasitic diode between the N-type battery side region of the high side switch and the substrate region exceeds the maximum rated voltage value of the battery. Since it has a withstand voltage, even if the generated voltage is 0 V, the first parasitic diode does not break down due to the battery voltage. According to the eighth configuration of the present invention, in the second configuration, the second parasitic diode between the N-type armature winding side region of the high side switch and the substrate region further has a predetermined G pulse voltage value. Since the withstand voltage exceeds, it is possible to prevent the surge voltage up to the G pulse voltage value from being applied to the load.

According to a ninth configuration of the present invention, in the second configuration, the third parasitic diode between the N-type armature winding side region of the low side switch and the substrate region further has a predetermined G pulse. Since it has a breakdown voltage exceeding the voltage value, this G
It is possible to prevent a surge voltage up to the pulse voltage value from being applied to the load. According to the tenth configuration of the present invention, since the N ⁻ type withstand voltage layer of the high-side switch is further formed on the first parasitic diode side in the seventh configuration, the pressure resistance of the first parasitic diode is increased. be able to.

According to the eleventh structure of the present invention, the eighth
Further, in the above configuration, since the N ⁻ type withstand voltage layer of the high side switch is formed on the second parasitic diode side, it is possible to increase the withstand voltage of the second parasitic diode that must withstand the G pulse voltage. According to the twelfth configuration of the present invention,
In the ninth configuration, the low side switch N
^{Since the −} type breakdown voltage layer is formed on the third parasitic diode side,
The withstand pressure of the third parasitic diode, which has to withstand the G pulse voltage, can be increased.

According to a thirteenth structure of the present invention, the second
Further, in the configuration, since the pair of high-side switch and low-side switch forming the phase inverter circuit of the same phase are integrated on the same chip, the structure and wiring can be simplified, and the miniaturization can be realized. According to a fourteenth configuration of the present invention, in the above second configuration, all high-side switches are turned off when the surge voltage cutoff mode is executed.
Turn on the low side switch. By doing so, the armature winding of each phase is short-circuited through the on-resistance of each low-side switch, so that surge power can be consumed without boosting the B voltage of the battery, and the G pulse voltage The increase can be prevented.

According to the fifteenth configuration of the present invention, in the conventional three-phase full-wave rectifier composed of N-channel MOS transistors (see FIG. 7), the generated voltage of the armature winding is a predetermined voltage value (predetermined G When the pulse voltage value) is exceeded, one of the low-side switches and all the high-side switches is turned on, and the other group is turned off. A part of (surge current) can be consumed (shunted) to be deenergized, and the G pulse voltage (surge voltage) applied to the load through the parasitic diode (first parasitic diode) of the high-side switch can be reduced accordingly. Can be weakened.

According to a sixteenth structure of the present invention, the above first
In the fourth or fifteenth configuration, further, the average gate voltage of all the high-side switches or all the low-side switches that are turned on at the same time when the surge voltage cutoff mode is performed, and their channel currents are equal to or less than the maximum allowable current value of each switch. Since it is controlled as described above, the switches that are turned on at the same time are not thermally destroyed.

According to the seventeenth structure of the present invention, the first
In the above configuration, even if a phase voltage (generated voltage of the armature coil) higher than the battery voltage is generated, the potential of the P-type well region is low, so that a parasitic between the P-type well region and the N-type region on the battery side. The diode is not forward biased and the battery will not charge uncontrolled. Therefore, the battery voltage can be controlled even in the magnet excitation type AC generator by the intermittent control of the MOS transistor that constitutes the high side switch.

According to an eighteenth structure of the present invention, the above first
In the configuration of 7, the high side switch is turned off while the battery voltage exceeds a predetermined voltage value, and
Turn on the low-side switch connected to the same phase end as the high-side switch. By doing so, the armature coils of each phase are short-circuited through the on-resistance of each low-side switch, and the surplus output current of the alternator is consumed by these low-side switch groups. Since the high voltage generated at the phase terminals during the period in which is turned off is suppressed, the breakdown of the MOS transistor is suppressed, and the reliability of the high side switch and the low side switch forming the phase inverter circuit is improved. Further, it is possible to suppress an increase in surge voltage due to the energy accumulated in the inductance of the armature coil when the output current is cut off.

According to a nineteenth structure of the present invention, the above first
In the configuration of No. 7, the high side switch and the low side switch forming the phase inverter circuit of the same phase are N
Since they are integrated on the same chip using the mold region as a substrate, the number of elements and wiring can be saved. According to the twentieth configuration of the present invention, in the conventional three-phase full-wave rectifier composed of N-channel MOS transistors, while the generated voltage of the armature winding is higher than the battery voltage and higher than the predetermined first voltage value. , The high side switch is turned off and all the low side switches are turned on, so the armature coils of each phase are short-circuited through the on resistance of each low side switch, and the surplus output current of the alternator is consumed by the low side switch etc. Therefore, the charging current to the battery can be prevented, the battery voltage can be adjusted, and the surge voltage when the switch is turned off can be weakened.

According to the above-mentioned first to sixteenth configurations of the present invention, in the case where the field current is normally controlled by the switching transistor, the switching transistor for controlling the field current or its exciting current control circuit system. Even if an abnormality occurs in the field current control, the battery voltage can be controlled well by controlling the MOS transistor of the three-phase full-wave rectifier, and overcharging of the battery can be prevented. it can.

[0035]

【Example】

(Embodiment 1) An embodiment of the charging device of the present invention used in a vehicle AC generator will be described below with reference to FIG. 1 is a three-phase AC generator for vehicles, and its armature windings 11 to 1
3 are connected to the AC input terminals (connection points described later) 41 to 43 of the three-phase full-wave rectifier 3 and the pair of DC output terminals of the three-phase full-wave rectifier 3 are connected to the battery 7. Connected to both ends of. A load 9 is connected to both ends of the battery 7 via fuses 10.

Reference numeral 2 is a field coil whose one end is connected to the high end of the battery 7, and the other end is connected to the collector of a field current control transistor whose emitter is grounded. Reference numeral 5 is a flywheel diode, 6 is a controller, and the controller 6 is a high-level terminal voltage (B
The field current is controlled by intermittently controlling the transistor 4 in accordance with the level of the voltage).
The generated voltage of 3 is controlled. The controller 6
Controls the transistors 31 to 36 of the three-phase full-wave rectifier 3 intermittently to full-wave rectify the generated voltage.

Next, the three-phase full-wave rectifier 3 will be described. The three-phase full-wave rectifier 3 includes three sets of high-side switches 31 to 33 and low-side switches 34 to 36, each of which is composed of a power NMOS transistor using Si or SiC having a higher breakdown voltage than Si, and which are individually connected in series. Phase inverter circuits 37 to 39 are connected in parallel, and a pair of DC output terminals are individually connected to the high level end and the low level end of the battery 7, and each switch 3 of each phase inverter circuit 37 to 39 is connected.
The connection points 1 to 36, that is, the AC input terminals 41 to 43 are individually connected to the phase output terminals of the armature windings 11 to 13 of the AC generator 1.

In addition, the main electrodes (displayed as drain electrodes D) of the low-side switches 34 to 36 on the battery low-potential terminal E side are located directly below the gate electrodes in the P-type substrate region (even if the P-type substrate is P.
(Although it may be a mold well region), a potential is applied to this substrate region, and a P-type substrate region immediately below the gate electrodes of the high side switches 31 to 33 is connected to the battery low potential terminal E to apply a potential.

Therefore, in this embodiment, the first parasitic diode D1, which is a junction between the main electrode (displayed as the source electrode S) of the high side switches 31 to 33 on the battery 7 side and the P-type substrate region, Side switch 31
No. 33 to a main electrode (displayed as a drain electrode D) on the side of the stator coils 11 to 13 and the above-mentioned P-type substrate region, a first parasitic diode D2 and the stator coils 11 to 11 of the low side switches 34 to 36. A third parasitic diode D3, which is a junction between the 13-side main electrode (displayed as the source electrode S) and the P-type substrate region, is parasitically formed.

Here, the first parasitic diode D1 has a withstand voltage exceeding the maximum rated voltage value of the battery, and even if the potential of the connection point 41 is 0 V (ground potential) or the forward voltage drop of the PN junction thereof. It does not break down even at low potentials. Further, the second parasitic diode D2 and the third parasitic diode D3 have a breakdown voltage exceeding a predetermined G pulse voltage value exceeding the maximum power generation voltage. The predetermined G pulse voltage value is the connection point 4 when the output current of the rectifier 3 becomes 0 from the predetermined current value at a predetermined time when a predetermined exciting current (field current) is supplied and at a predetermined rotation.
It is a surge voltage value induced in 1 to 43, for example, 300 V here and the maximum voltage of the battery 7 is 14 V.

Further, when the G pulse voltage is applied to the connection point 41, the breakdown voltage of the gate insulating film of the transistors 31 to 36 between the N-type regions on the connection points 41 to 43 side and the gate electrode is secured, and the high voltage is ensured. Transistors 31 to 31 between the N-type regions on the battery side of the side switches 31 to 33 and the gate electrodes
The breakdown voltage of the gate insulating film of 36 is also secured. As is well known, ensuring the breakdown voltage of each of these parts is achieved by transistors 31 to 36.
It is realized by selecting the impurity concentration of each part and the gate oxide film thickness.

Particularly in this embodiment, in order to improve the withstand voltage of the second parasitic diode D2 and the third parasitic diode D3,
The transistors 31 to 36 have a DMOS structure (see FIG. 5) or a vertical power MOS structure (see FIG. 6), and N
^The -type withstand voltage layer is formed on the second parasitic diode D2 and the third parasitic diode D2.
The parasitic diode D3 is arranged to facilitate securing the breakdown voltage of the second parasitic diode D2 and the third parasitic diode D3 against the G pulse voltage and the breakdown voltage between the N-type region on the connection points 41 to 43 side and the gate electrode. . In this case, since the substrates of the high-side switch and the low-side switch forming the same-phase inverter circuit have the same potential, a one-chip configuration can be adopted, and a total of three chips can form a rectifier.

Controller (control section in the present invention) 6
Has a built-in microcomputer, controls the exciting current so that the B voltage becomes a predetermined level, and switches each switch based on the phase and magnitude of the generated voltage at each phase output end for three-phase full-wave rectification. 31 to 36 are intermittently controlled. Hereinafter, the control operation of the controller 6 that executes the intermittent control method of the switches 31 to 36 in this embodiment will be described more specifically.

First, the potentials (each phase voltage) V1 to V3 of the connection points (each phase output end) 41 to 43 are detected, and each phase voltage V1 is detected.
~ V3 is compared with the battery voltage and the ground potential, and the high side switch to which a phase voltage higher than the battery voltage (B voltage) is applied and the low side switch of the other phase inverter circuit are turned on. In this way, three-phase full-wave rectification can be implemented.

Alternatively, the phase voltages V1 to V3 are compared with the battery voltage and the ground potential, the phase voltages V1 to V3 are higher than the battery voltage and the ground battery voltage (B voltage), and the highest phase voltage is applied. The high side switch and the low side switch to which a phase voltage lower than the ground potential E is applied are turned on. With this configuration, different high-side switches do not turn on at the same time to cause short-circuit current, and different low-side switches do not turn on at the same time to cause short-circuit current.

Further, in the present embodiment, each phase voltage V1 to V3.
When any of the above exceeds the predetermined G pulse voltage value, the high side switches 31 to 33 are cut off, so that the current to the load 7 is suddenly reduced (for example, the fuse 10 is turned off) or the battery connection terminal B is disconnected, and the rectifier. The surge voltage generated when the output current of 3 drops sharply is high side switch 31-
Since the channel of 33 is completely extinguished and the second parasitic diode D2 completely blocks, the G pulse voltage is not applied to the load 9. The surge voltage cutoff mode can be maintained for a predetermined time, or can be maintained while one of the phase voltages exceeds a predetermined voltage value.

Further, all the low side switches 34 to 36 can be turned on when the surge voltage cutoff mode for turning off all the high side switches 31 to 33 is executed. By doing so, the armature windings 11 to 13 of the respective phases are short-circuited through the ON resistances of the low-side switches 34 to 36, and the surge power can be consumed without boosting the B voltage of the battery. It is possible to prevent the G pulse voltage from increasing.

The control operation of the controller 6 described above may be carried out by judging whether the input voltages V1 to V3 and the B voltage are relative to each other or a predetermined threshold value by the built-in microcomputer and based on the result of this judgment, a normal software control method. Can be implemented by FIG. 2 shows an example of the flowchart.
That is, in step 100, the input phase voltages V1 to V3 and the battery voltage VB are input, and then the input phase voltages V1 to V3 are input.
It is determined whether or not any of 3 is larger than a predetermined G pulse voltage value Vg (102), and if it is less than or equal to, step 104
In step 3, normal three-phase full-wave rectification is performed by the sequential opening / closing control of the switches 34 to 36, and if exceeding, step 10
At 6, the high-side switches 31 to 33 are turned off and the low-side switches 34 to 36 are turned on to reduce the surge voltage and prevent the surge voltage from being applied to the load 9. (Modification) In the above embodiment, in step 106, the on-resistance values of the low-side switches 34 to 36 are
It is preferable that those average channel currents are controlled to be equal to or less than the maximum allowable current value. That is, a PWM voltage or an average voltage having a predetermined duty ratio is applied to the gate electrodes of the low side switches 34 to 36 according to the maximum allowable current value. In this way, the low side switches 34 to 36 are not thermally destroyed. (Modification) In the above embodiment, it is determined that the G pulse voltage is generated by comparing the phase voltages V1 to V3 with a predetermined G pulse voltage value. However, the battery voltage VB is compared with the predetermined G pulse voltage value. It goes without saying that the determination may be made based on the battery voltage VB and the phase voltages V1 to V3.
Is compared with a predetermined G pulse voltage value and is naturally included in claims 2 and after of the present invention. (Modification) Next, an example of a circuit in which the operations of steps 102 and 106 are configured by hardware will be described with reference to FIG. However, FIG. 3 shows the phase voltage (X
Phase) only.

The phase voltage V1 is divided by the voltage dividing circuit 58 and input to the + input terminal of the comparator 51 and the − input terminal of the comparator 52, and the battery voltage VB is divided by the voltage dividing circuit 58 to be − of the comparator 51. It is input to the input terminal, and the + input terminal of the comparator 52 is grounded. As a result, the comparator 51 outputs a high level to the AND circuit 54 when the phase voltage V1 is higher than the battery voltage VB, and the comparator 52 turns on the low side switch 34 when the phase voltage V1 is less than 0V.

On the other hand, the battery voltage VB is input to the + input terminal of the comparator 53, which is divided by the voltage dividing circuit 58, and the minus input terminal thereof has a predetermined threshold voltage (corresponding to the predetermined G pulse voltage value). ) Vth is input. As a result, the comparator 53 outputs a high level potential to the grounded-emitter transistor 56 when the divided voltage of the battery voltage VB is larger than a predetermined G pulse voltage value Vth (when a surge voltage occurs), and the transistor 56 is turned on. Then, the capacitor C is discharged.

The collector voltage of the transistor 56 is input to the AND circuit 54. Therefore, the AND circuit 54 raises the low level potential to the booster circuit 5 during the generation of the surge voltage.
5 is output. The booster circuit 55 is, for example, a switching capacitor circuit, which boosts the battery voltage to a level at which the high-side switch 31 can be turned on. When the low-level potential is input from the AND circuit 54, the low-level is output to the high-side switch 31. As a result, the high side switch 31 is turned off.

When the surge voltage disappears, the comparator 5
3 outputs a low level, the transistor 56 is turned off, charging of the capacitor C is started, and after a predetermined time determined by the CR time constant, the AND circuit 54 causes the comparator 51 to operate.
Can be opened and closed. With this configuration, the high-side switch 31 continues to be shut off for a predetermined time (at least one cycle) even in a phase period in which the instantaneous value of each phase voltage that is an AC voltage is equal to or less than the predetermined voltage value Vth. When the phase voltage applied to the other high side switch 32 or 33 exceeds the predetermined voltage value Vth, the high side switch 31
Can be shut off. (Modification) Next, another circuit example in which the operations of steps 102 and 106 are configured by hardware will be described with reference to FIG.

In this embodiment, the comparator 53 of FIG.
A transistor 56 and an integrating circuit 59 are used to compare a peak value hold circuit including an operational amplifier 53, a diode Di and a capacitor C, and an output voltage of the peak value hold circuit with a threshold voltage (predetermined G pulse voltage value). It is replaced with a comparator 60 that operates. To explain this operation, the divided voltage of the phase voltage V1 is input to the negative input terminal of the comparator 60 while being held by the peak value hold circuit. As a result, the instantaneous maximum value (peak value) of the phase voltage V1 is the predetermined threshold voltage (corresponding to the predetermined G pulse voltage value) Vth.
When it exceeds 1 (when a surge voltage is generated), the comparator 60 turns off the AND circuit 54, whereby the high side switch 51 is cut off.

Here, in the reset circuit composed of the comparator 61 and the transistor 62, the battery voltage VB is divided by the voltage dividing circuit 58 and input to the minus input terminal of the comparator 61, and the plus input terminal thereof has a predetermined value. The threshold voltage (reset occurrence) Vth2 is input. As a result, when the divided voltage of the battery voltage VB becomes smaller than the predetermined reset voltage Vth2, the comparator 61 outputs a high level to the grounded-emitter transistor 62, the transistor 62 is turned on to discharge the capacitor C, and the peak hold circuit is provided. Give reset action.

As a result, the same effect as that of the circuit of FIG. 3 can be obtained. (Modification) FIG. 5 shows an example of a DMOS integrated circuit in which the high-side switch 31 and the low-side switch 34 which form the X-phase inverter circuit 37 are integrated on the same chip.
An example of the vertical MOS integrated circuit is shown in FIG.

In FIG. 5, an N-type breakdown voltage layer 105 is formed on the N ⁺ -type substrate 106 by epitaxial growth,
A P-type well region 103 is formed on the surface of the die breakdown layer 105, and an N ⁺ -type region 104 is further formed on the surface of the P-type well region 103. Then, the gate insulating film 109
And a gate electrode 110 made of doped polysilicon is formed thereon.

Therefore, in this embodiment, the first parasitic diode D1 of the high side switch 31 arranged on the left side in FIG. 5 is formed from the PN junction between the N ⁺ type region 104 and the P type well region 103. And the second parasitic diode D2 is N
⁺ Consists PN junction between the substrate 106 and the P-type well region 103, in FIG. 5, a third parasitic diode of the low-side switch 34 disposed on the right - de D3 is N ⁺ substrate 106 and P
It consists of a PN junction with the mold well region 103.

In other words, in this embodiment, the high-side switch 31 and the low-side switch 34 can be integrated on the same chip, and the second and third parasitic diodes D2 and D3 which require a large surge withstand voltage (G pulse withstand voltage). Has a low-concentration N-type epitaxial breakdown layer 105, so that it has an excellent feature that a large surge breakdown voltage can be secured.

FIG. 6 shows a vertical MOS structure instead of the DMOS structure shown in FIG. 5, and can achieve the same effects as those of FIG. Furthermore, the structure shown in FIGS. 5 and 6 also simplifies wiring. (Embodiment 2) Another embodiment of the present invention will be described with reference to FIG. In this embodiment, in the conventional MOS transistor type three-phase full-wave rectifier described in the section of the prior art, the phase voltages V1 to V3 exceed a predetermined voltage value (G pulse voltage value) as in the first embodiment. It is determined whether or not the high side switches 31 to 33 or the low side switches 34 to 36 are turned off and the other is turned on.

Here, the case where the high side switches 31 to 33 are turned off and the low side switches 34 to 36 are turned on will be described. Now, even if the phase voltage V1 becomes a G pulse (giant pulse) voltage, all the low-side switches 3
Since 4 to 36 are turned on, each phase output terminal is short-circuited through the channel resistance of the low side switches 34 to 36, and the current due to the G pulse voltage bypasses the load 9 and flows through the low side switches 34 to 36. , The G pulse voltage applied to the load 9 is reduced.

Also in this embodiment, by controlling the gate voltage of the low side switches 34 to 36, the channel current is adjusted to control the low side switch 34.
~ 36 can be protected from thermal destruction. (Embodiment 3) Another embodiment of the charging device of the present invention used in a magnet-excited AC generator forming an AC generator for a vehicle will be described below with reference to FIG.

In this embodiment, the vehicle three-phase AC generator 1 in Embodiment 1 (see FIG. 1) is replaced with a magnet excitation AC generator. The field coil 2 and the flyback diode shown in FIG. 1 are used. 5, the switching transistor 4 is omitted, and the rotating field pole 2a has a total of four poles. An embodiment of the charging device of the present invention used for a vehicle AC generator will be described below with reference to FIG.

Reference numeral 1 denotes a three-phase AC generator for a vehicle, and the output terminals (each phase output terminal) of the armature windings 11 to 13 are each AC input terminals of the three-phase full-wave rectifier 3 (connection points described later). ) 41-4
3 and the pair of DC output terminals of the three-phase full-wave rectifier 3 are connected to both ends of the battery 7. Reference numeral 6 denotes a controller. The controller 6 intermittently controls the transistors 31 to 36 in accordance with the level of the high-order terminal voltage (B voltage) of the battery 7 to perform full-wave rectification of the generated voltage and also to perform high-side switches 31 to 33. Is controlled intermittently to control the battery voltage to a predetermined level.

The three-phase full-wave rectifier 3 is the same as that of FIG. The second parasitic diode D2 and the third parasitic diode D3 have a withstand voltage exceeding the maximum generated voltage. For example, here, the maximum generated voltage is 300V and the maximum voltage of the battery 7 is 14V. Furthermore, when the maximum generated voltage is applied to the connection point 41, the breakdown voltage of the gate insulating film of the transistors 31 to 36 between the N-type regions on the connection points 41 to 43 side and the gate electrode is also secured, and the high side switch 31 ~ 33
The withstand voltage of the gate insulating films of the transistors 31 to 36 between the N-type region on the battery side and the gate electrode is also secured. As is well known, securing the breakdown voltage of each of these parts is realized by selecting the impurity concentration of each part of the transistors 31 to 36 and the gate oxide film thickness.

Particularly, in this embodiment, in order to improve the withstand voltage of the second parasitic diode D2 and the third parasitic diode D3,
The transistors 31 to 36 have a DMOS structure (see FIG. 5), and their N ⁻ type withstand voltage layers are arranged on the second parasitic diode D2 and the third parasitic diode D3, and the second parasitic diode D2 for the maximum generated voltage and The withstand voltage of the third parasitic diode D3 and the withstand voltage between the N-type region on the side of the connection points 41 to 43 and the gate electrode are easily secured. In this case, since the substrates of the high-side switch and the low-side switch that form the inverter circuit of the same phase have the same potential,
A one-chip configuration can be adopted, and a rectifier can be configured with a total of three chips.

Controller (control section in the present invention) 6
Incorporates a microcomputer to intermittently control each of the switches 31 to 36 on the basis of the phase and magnitude of the generated voltage at each phase output end for three-phase full-wave rectification, and to control the high side switches 31 to 33. By controlling the on time by PWM, B
The voltage is adjusted to a predetermined level. Hereinafter, the control operation of the controller 6 that executes the intermittent control method of the switches 31 to 36 in this embodiment will be described more specifically.

First, the potentials (each phase voltage) V1 to V3 at the connection points (each phase output end) 41 to 43 are detected, and each phase voltage V1 is detected.
~ V3 is compared with the battery voltage and the ground potential, and the high side switch to which a phase voltage higher than the battery voltage (B voltage) is applied and the low side switch of the other phase inverter circuit are turned on. In this way, three-phase full-wave rectification can be implemented.

Further, the high side switches 31 to 33
Is turned on only while the battery voltage (B voltage) is lower than a predetermined value, the battery voltage can be controlled.
Further, when the three-phase full-wave rectification is performed, the phase voltages V1 to V3 are compared with the battery voltage and the ground potential, and the phase voltages V1 to V1 are compared.
The high side switch to which V3 is higher than the battery voltage and the ground battery voltage (B voltage) and the highest phase voltage is applied, and the low side switch to which the phase voltage lower than the ground potential E is applied are turned on. With this configuration, different high-side switches do not turn on at the same time to cause short-circuit current, and different low-side switches do not turn on at the same time to cause short-circuit current.

Further, in this embodiment, when the battery voltage is higher than the predetermined value, the high side switches 31 to 33 are cut off, and the high phase voltage generated at that time is completely blocked by the second parasitic diode D2. Therefore, it becomes possible to adjust the battery voltage to a desired level. Further, when turning off all the high side switches 31 to 33, all the low side switches 34 to 36 can be turned on. By doing so, the armature windings 11 to 13 of the respective phases are short-circuited through the ON resistances of the low side switches 34 to 36. Therefore, the surplus output current of the generator 1 is consumed by the low-side switches 34 to 36 that are short-circuited, so that the high voltage generated at the phase terminals while the high-side switches 31 to 33 are off is suppressed and the MOS transistor 31 is suppressed.
A breakdown of ~ 36 can be prevented. Further, it is possible to reduce the surge voltage due to the energy accumulated in the inductance of the armature coils 11 to 13 when the output current is cut off.

The control operation of the controller 6 described above may be carried out by judging whether the input voltages V1 to V3 and the B voltage are relative to each other or a predetermined threshold value by the built-in microcomputer, and based on the result of this judgment, a normal software control method. Can be implemented by FIG. 9 shows an example of the flowchart.
That is, in step 100, the input phase voltages V1 to V3 and the battery voltage VB are input, and then the input phase voltages V1 to V3 are input.
It is determined whether any of 3 is larger than the battery voltage VB (102), and if it is larger than the battery voltage VB, it is determined whether the battery voltage VB is lower than a predetermined adjustment voltage Vs (103),
If it is low, the normal switch 34-
Three-phase full-wave rectification is performed by the sequential opening / closing control of 36, and if the battery voltage VB exceeds the adjustment voltage Vs, the high side switches 31 to 33 are turned off at step 106, and based on V1 to V3 and VB. Low side switch 3
4 to 36 are sequentially controlled to be opened and closed to reduce a high generated voltage, and the battery voltage is controlled to a predetermined value. (Modification) In the above embodiment, in step 106, the on-resistance values of the low-side switches 34 to 36 are
It is preferable that those average channel currents are controlled to be equal to or less than the maximum allowable current value. That is, a PWM voltage or an average voltage having a predetermined duty ratio is applied to the gate electrodes of the low side switches 34 to 36 according to the maximum allowable current value. In this way, the low side switches 34 to 36 are not thermally destroyed. (Modification) Next, an example of a circuit in which the operations of steps 102 and 106 are configured by hardware will be described with reference to FIG. However, FIG. 10 shows only the phase voltage (X phase) of the armature winding 11.

The phase voltage V1 is divided by the voltage dividing circuit 57 and input to the + input terminal of the comparator 51 and the − input terminal of the comparator 52, and the battery voltage VB is divided by the voltage dividing circuit 58 and-of the comparator 51. It is input to the input terminal, and the + input terminal of the comparator 52 is grounded. As a result, the comparator 51 outputs a high level to the AND circuit 54 when the phase voltage V1 is higher than the battery voltage VB, and the comparator 52 turns on the low side switch 34 when the phase voltage V1 is less than 0V.

On the other hand, the battery voltage VB is divided by the voltage dividing circuit 58 and input to the-input terminal of the comparator 53, and the + input terminal thereof has a predetermined threshold voltage (corresponding to the predetermined battery voltage value Vs). ) Vth is input. As a result,
The comparator 53 outputs a low level to the AND circuit 54 when the divided voltage of the battery voltage VB is higher than the predetermined battery voltage Vth.

Therefore, the AND circuit 54 outputs a low level potential to the booster circuit 55 while the battery voltage is higher than a predetermined battery voltage value. The booster circuit 55 is, for example, a switching capacitor circuit, which boosts the battery voltage to a level at which the high-side switch 31 can be turned on. When the low-level potential is input from the AND circuit 54, the low-level is output to the high-side switch 31. The high side switch 31
Turn off. (Modification) The three-phase full-wave rectifier 3 shown in FIG. 8 is also the DM of FIG.
Obviously, the OS integrated circuit structure can be adopted. (Embodiment 4) Another embodiment of the present invention will be described with reference to FIG.

In this embodiment, the AC generator shown in Embodiment 2 (see FIG. 7) is replaced with a magnet excitation type AC generator. That is, this embodiment determines whether or not the phase voltages V1 to V3 exceed the battery voltage value VB in the conventional MOS transistor type three-phase full-wave rectifier described in the section of the related art as in the first embodiment. However, in the above case, the high side switches 31 to 33 are turned on while the battery voltage VB is lower than the predetermined adjustment voltage Vs, and turned off when the battery voltage VB is higher than the predetermined adjustment voltage Vs.

Here, the case where the high side switches 31 to 33 are turned off and the low side switches 34 to 36 are turned on will be described. Now, even if the phase voltage V1 becomes higher than the battery voltage, all the low side switches 34
Is turned on, the output end of each phase is low side switch 34 ~
Will be shorted through the 36 channel resistance,
The battery 7 is not charged with a current due to a phase voltage higher than the battery voltage VB and flows through the low-side switches 34 to 36, so that the battery voltage VB can be controlled to a predetermined value.

Also in this embodiment, by controlling the gate voltage of the low side switches 34 to 36, the channel current of the low side switches 34 to 36 is adjusted to control the low side switch 34.
~ 36 can be protected from thermal destruction.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of a charging device of the present invention.

FIG. 2 is a flowchart showing the operation of the controller 6 of FIG.

3 is a circuit diagram showing an example in which the controller 6 of FIG. 1 is configured by hardware.

4 is a circuit diagram showing an example in which the controller 6 of FIG. 1 is configured by hardware.

5 is a sectional view showing an example of a phase inverter circuit 37 of FIG.

6A is a sectional view showing an example of the phase inverter circuit 37 of FIG. 1, and FIG. 6B is a plan view thereof.

FIG. 7 is a circuit diagram showing another embodiment of a conventional MOS transistor type three-phase full-wave rectifier and the charging device of the present invention.

FIG. 8 is a circuit diagram showing a charging device according to a third embodiment.

9 is a flowchart showing the operation of the controller 6 of FIG.

10 is a circuit diagram showing an example in which the controller 6 of FIG. 1 is configured by hardware.

FIG. 11 is a circuit diagram showing a charging device according to a fourth embodiment.

[Explanation of symbols]

1 is a three-phase AC motor, 3 is a rectifier, 6 is a controller (controller), 7 is a battery, 9 is a load, 11 to 13 are armature windings, 31 to 33 are high-side switches composed of MOS transistors, 34 to 36 is a low-side switch composed of a MOS transistor, 37 to 39 are phase inverter circuits, 41 to 43 are connection points of the phase inverter circuits 37 to 39 (AC input terminals of the rectifier 3), 103 is a P-type region (substrate region), and D1. the first parasitic diode - de, D2 the second parasitic diode - de, D3 the third parasitic diode - de, 106 N-type armature winding side region of the high-side switch 31 and low-side switch 34, the 105 N ^- Mold breakdown layer.

Claims (20)

[Claims] 1. A required number of phase-inverter circuits in which a high-side switch and a low-side switch composed of N-channel MOS transistors are connected in series are connected in parallel,
A rectifier having a pair of DC output terminals connected to both ends of the battery and the load, and a connection point of both switches individually connected to each phase output terminal of the armature winding of the AC generator, and a gate potential of each switch. And a control unit for connecting and disconnecting each of the switches to control the switch, wherein a substrate region formed of a P-type region immediately below the gate electrode of the high-side switch is connected to the low end of the battery. Charging device. 2. The control unit detects a phase voltage at each phase output terminal of the AC generator and shuts off at least the high side switch group when any one of the phase voltages exceeds a predetermined voltage value. The charging device according to claim 1, which is provided with a surge voltage cutoff mode. 3. The charging device according to claim 2, wherein the control unit keeps the surge voltage cutoff mode for a predetermined time. 4. The charging device according to claim 2, wherein the control unit maintains the surge voltage cutoff mode while any of the phase voltages exceeds a predetermined voltage value. 5. The control unit is configured such that, in each of the high-side switches, the phase voltage applied to an N-type armature winding side region forming a main electrode on a connection point side is higher than a voltage of the battery. The charging device according to claim 2, wherein a side switch is turned on. 6. The low-side switch, wherein the control section has the phase voltage applied to a region of an N-type armature winding forming a main electrode on a connection point side of the low-side switches lower than a voltage of the battery. The charging device according to claim 2, wherein the charging device is turned on. 7. The first parasitic diode between the N-type battery side region forming the battery-side main electrode of the high-side switch and the substrate region has a withstand voltage exceeding the maximum rated voltage value of the battery. Item 2. The charging device according to item 2. 8. The second parasitic diode between the N-type armature winding side region forming the main electrode on the connection point side of the high-side switch and the substrate region has a predetermined G exceeding a maximum generated voltage.
The charging device according to claim 2, which has a withstand voltage exceeding a pulse voltage value. 9. A third parasitic diode between the N-type armature winding side region forming the main electrode on the connection point side of the low side switch and the substrate region has a predetermined G exceeding a maximum generated voltage.
The charging device according to claim 2, which has a withstand voltage exceeding a pulse voltage value. 10. The MO forming the high side switch.
8. The charging device according to claim 7, wherein one of both N-type regions forming the main electrode of the S transistor has an N ⁻ type withstand voltage layer, and the N ⁻ type withstand voltage layer is formed on the first parasitic diode side. 11. The MO forming the high side switch.
9. The charging device according to claim 8, wherein one of both N-type regions forming the main electrode of the S transistor has an N ⁻ type withstand voltage layer, and the N ⁻ type withstand voltage layer is formed on the second parasitic diode side. 12. The MO forming the low side switch.
The charging device according to claim 9, wherein one of both N-type regions forming the main electrode of the S transistor has an N ⁻ type withstand voltage layer, and the N ⁻ type withstand voltage layer is formed on the third diode side. 13. The high-side switch and the low-side switch forming the phase inverter circuit of the same phase are integrated on the same chip having an N-type region as a substrate, and the N-type substrate region is at the phase output end of the AC generator. The charging device according to claim 2, which is connected. 14. The charging device according to claim 2, wherein the control unit turns off all the high side switches and turns on the low side switches when the surge voltage cutoff mode is executed. 15. A required number of phase inverter circuits in which a high-side switch and a low-side switch composed of N-channel MOS transistors are connected in series are connected in parallel, and a pair of DC output terminals are connected to both ends of a battery and a load, respectively. A rectifier in which a connection point of the both switches is individually connected to each phase output end of an armature winding of an alternator, and a control unit which controls a gate potential of each switch to connect and disconnect each switch A substrate region consisting of a P-type region directly below the gate electrode of the high-side switch is connected to the N-type region on the armature winding side to be given a potential, and a substrate region consisting of a P-type region directly below the gate electrode of the low-side switch. In a charging device in which a region is connected to an N-type region on the lower terminal side of the battery and a potential is applied to the battery, the controller controls the generated voltage of the armature winding. When a voltage exceeds a predetermined voltage value, a surge voltage cut-off mode is implemented in which one of the low-side switches and all the high-side switches is turned on and the other group is turned off. Charging device. 16. The control unit determines an average gate voltage of all the high-side switches or all the low-side switches that are turned on at the same time when the surge voltage cutoff mode is executed, based on a maximum allowable current value of each switch. The charging device according to claim 14, wherein the charging device is a battery. 17. The alternating current generator comprises a magnet excitation type alternating current generator, and the control unit detects a phase voltage at each phase output end of the alternating current generator and a voltage of the battery,
In each of the high-side switches, the phase voltage applied to the N-type armature winding side region forming the main electrode on the connection point side is higher than the voltage of the battery, and the voltage of the battery is less than a predetermined voltage value. The charging device according to claim 1, wherein the high side switch is turned on during the period. 18. The control unit turns off the high side switch while the voltage of the battery exceeds a predetermined voltage value, and turns on at least the low side switch connected to the same phase end as the high side switch. 18. The charging device according to claim 17, which is a charging device. 19. The high side switch and the low side switch forming the phase inverter circuit of the same phase are integrated on the same chip having an N type region as a substrate, and the N type substrate region is at a phase output end of the AC generator. Claim 17 connected
The charging device described. 20. A required number of phase inverter circuits in which a high-side switch and a low-side switch composed of N-channel MOS transistors are connected in series are connected in parallel, and a pair of DC output terminals are connected to both ends of a battery and a load. A rectifier in which a connection point of the both switches is individually connected to each phase output end of an armature winding of an alternator, and a control unit which controls a gate potential of each switch to connect and disconnect each switch A substrate region consisting of a P-type region directly below the gate electrode of the high-side switch is connected to the N-type region on the armature winding side to be given a potential, and a substrate region consisting of a P-type region directly below the gate electrode of the low-side switch. In the charging device in which the region is connected to the N-type region on the lower terminal side of the battery and a potential is applied, the AC generator is a magnet excitation type AC generator. The control unit detects the phase voltage of each phase output terminal of the AC generator and the voltage of the battery, and one of the phase voltages is higher than the voltage of the battery, and A charging device, wherein the high side switch is turned on while the voltage is less than a predetermined first voltage, and further, all the low side switches are turned on while the high side switch is turned off.