JPH08278362A - Radar device - Google Patents

Abstract

PURPOSE: To reduce measurement time and the number of parts, miniaturize a device, reduce the device cost, and to improve measurement precision by measuring the distance from an object and relative speed at the same time at a common circuit. CONSTITUTION: Relating to a radar device wherein electric wave whose band is diffused by PN code from a PN generator 12 is transmitted by a transmission part 13 and then the reflected wave from an object 10 on the basis of the electric wave is received by a receiver part 17 and then the correlation between the received signal and the PN code is detected for detecting the object, the received signal diffused into wide-band is converted into a low frequency band, in which measurement is easily performed, by a down converter 18, and signal is generated when correlation is established on the basis of delay of the PN code from a delay circuit 15. Then the signal is waveform-shaped for generating pulse signal, and then the object is detected and its relative speed and distance are measured by a calculation process part 11 on the basis of the pulse signal and delay time.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a radar device for detecting the presence of an object, measuring a relative velocity and measuring a distance to the object by using a spread spectrum technique, and more particularly to a radar device used for an automotive radar.

[0002]

Related Background Art Conventionally, in this type of radar device,
In the direct spread method, a method using a sliding correlation method in which a pseudo noise code (hereinafter referred to as “PN code”) sequence is sequentially delayed and a correlation with a reflected wave is confirmed is known. As an example of using this sliding correlation method, there is a method of preparing a plurality of receiving PN codes whose sliding amounts are changed in fine steps and sequentially confirming the correlation, as in JP-A-5-256936. It was

[0003]

However, Japanese Unexamined Patent Publication No.
In JP-A-256936, since it is necessary to prepare an independent correlation circuit / signal detection circuit for each of a plurality of reception PN codes having different sliding amounts, the circuit becomes complicated and the circuit scale becomes large. Therefore, there is a problem that the manufacturing cost becomes high. In addition, there is a problem that the moving direction of the object cannot be determined only by the information of the frequency displacement.

Further, some radar devices using the sliding correlation method obtain velocity information secondarily from displacements of distance information. However, since displacements of distances need to be observed for a predetermined time, the measurement time is long. There was a problem that Further, some radar devices using the sliding correlation method determine the chip and period length of the PN code based on the requirements of distance measurement accuracy and maximum measurement distance. However, if the chip length and one cycle length are determined only by these requirements, as shown in FIG. 10, for example, relative velocities v1, v2
Reflected waves a from a plurality of different objects (vehicles A and B),
When b is received by a vehicle C equipped with a radar device and, for example, a correlation with a reflected wave a is obtained, one reflected wave b acts as noise, which causes a problem that measurement performance deteriorates.

The present invention has been made in view of the above problems, and measures the distance to an object and the relative speed at the same time by a common circuit to reduce the measurement time and the number of parts, thereby reducing the size and size of the device. An object of the present invention is to provide a radar device that realizes cost reduction and improves measurement performance. Also,
Another object of the present invention is to provide a radar device that improves noise resistance.

Another object of the present invention is to provide a laser device capable of obtaining a certain range of detection accuracy and detection time.

[0007]

In order to achieve the above object, the present invention transmits a radio wave whose band is spread by a pseudo noise signal, receives a reflected wave from an object based on the radio wave, and receives the received signal. In the radar device for detecting the object by detecting the correlation between the pseudo noise signal and the pseudo noise signal, a despreading means including a receiving unit for despreading the received signal with the pseudo noise signal, and a frequency of the received signal of a low frequency Frequency conversion means including a down converter for converting into a signal, identification means including an arithmetic processing unit for identifying the presence of the object based on the frequency-converted signal, and operation for measuring the cycle of the frequency-converted signal A measuring unit including a processing unit; a detecting unit including an arithmetic processing unit that detects the velocity of the object according to the measured period; and the pseudo noise signal when detecting the correlation. A delay unit configured to delay a predetermined time, a distance measuring unit including a calculation processing unit that measures a distance to the object according to the delay time when the correlation is detected, and the frequency conversion unit. Provided is a radar device equipped with a passing means consisting of a narrow band filter for passing the signal in a narrow band, measuring the distance to the object from the delay time when the correlation is detected, and despreading the received signal. Is converted to a low frequency suitable for period measurement, and the relative velocity of the object is detected from the period change of the signal.

In the eleventh aspect, a radio wave whose band is spread by a pseudo noise signal is transmitted, a reflected wave from an object based on the radio wave is received, and a correlation between the received signal and the pseudo noise signal is detected. In a radar device that detects the presence of the object, measures the velocity, or measures the distance to the object, the shift range due to the Doppler effect of the frequency of the received signal in which the power is correlated and converted into low frequency is not correlated. Does not overlap with the shift range of the frequency of the as-spread signal,
And an oscillating means for setting and oscillating the sign of the pseudo noise signal according to the assumed relative velocity of the object so that the minimum value of the frequency shift range of the power-concentrated signal becomes larger than "0". A despreading means for despreading the received signal with the oscillated pseudo noise signal, and a passing means comprising a narrow band filter for passing the despread signal through a frequency shift range due to the Doppler effect of the frequency of the concentrated signal. Even if a plurality of reflected waves are received, the narrow band filter extracts only the frequency component in which the power is concentrated.

According to a twelfth aspect of the present invention, transmitting means for transmitting a radio wave whose band is spread by a pseudo noise signal, receiving means for receiving a reflected wave from an object based on the radio wave, and the received signal by the pseudo noise signal. Despreading means for despreading, and frequency conversion means for converting the frequency of the received signal into a low-frequency signal, and based on the frequency-converted signal, presence detection of the object, speed measurement, or even the object In a radar device that performs distance measurement of a phase-locked loop (hereinafter, referred to as “P”) that passes the frequency converted signal in a narrow band.
LL ". ) Circuit and a PLL circuit is used to reduce the mixing of noise in the pass band.

In the thirteenth and fourteenth aspects, the PLL circuit has a phase comparator, a loop filter, a voltage controlled oscillator (hereinafter referred to as "VCO"), and a feedback loop from the VCO to the phase comparator. Then, the frequency-converted signal is taken in, and the measuring means performs the presence detection of the object, the speed measurement, or the distance measurement to the object based on the output from the loop filter or the VCO.

According to the sixteenth and seventeenth aspects, the PLL circuit has a lock-in range or a lock-in range within an expected frequency displacement range of the signal which is correlated by the despreading means, the power is concentrated, and the frequency is converted by the frequency converting means. A lock range is set, a lock-in range of the PLL circuit or a lock-in range of the PLL circuit is set in the expected frequency displacement range of the spectrum that has been spread without being correlated by the despreading means and has been spread by the frequency converting means. The lock ranges are set so that they do not overlap, and even if a plurality of reflected waves are received, only the frequency component in which power is concentrated is extracted by the PLL circuit that functions as a narrow band filter.

In the eighteenth aspect of the present invention, the radar device comprises cycle length measuring means for measuring the length of one cycle and a plurality of cycles of the frequency-converted signal in a predetermined time, and the cycle length measuring means converts the frequency. According to the cycle of the signal, the number of the cycles to be measured is made variable to obtain a certain range of detection accuracy and detection time.

[0013]

DETAILED DESCRIPTION OF THE INVENTION A radar device according to the present invention will be described with reference to the drawings of FIGS. FIG. 1 is a block diagram showing a schematic configuration of an embodiment of a radar device according to the present invention. In the figure, the radar device according to the present embodiment includes an arithmetic processing unit 1 that performs various controls and arithmetic processing on an input signal.
1, a PN generator 12 connected to the arithmetic processing unit 11 to generate a PN code, a transmitting unit 13 for transmitting radio waves spread in a band by a direct spreading method, a transmitting antenna 14, an arithmetic processing unit 11 and a PN. A delay circuit 15 connected to the generator 12 for delaying and outputting a PN code for a predetermined time, a receiving antenna 16, a receiving unit 17 for receiving and despreading a reflected wave from the object 10, and down-converting a received signal. And a down converter 18 that operates. The transmitting antenna 14 and the receiving antenna 16 can be used for both transmission and reception by a circulator or other means.

The PN generator 12 includes an arithmetic processing unit 11
Under the control of, the random PN code having a predetermined length (cycle) is output to the transmission unit 13 and the delay circuit 15. In the transmitter 13, a high frequency signal spread over a wide band is generated by the input PN code, and the high frequency signal is transmitted as a radio wave via the transmitting antenna 14. Note that, for example, when the radar device of the present embodiment is used in a general automobile, the signal generated by the transmitter 13 is a millimeter wave band signal having an optimum frequency for radar, for example, 60.
It is a high frequency signal of GHz.

On the other hand, the delay circuit 15 delays the input PN signal by a predetermined time t1 and outputs it to the receiving unit 17 under the control of the arithmetic processing unit 11. The transmitted radio wave reaches the object 10 separated by a distance L, for example, and the object 1
It is reflected by 0 and is received by the receiving unit 17 as a received signal via the receiving antenna 16. In the present embodiment, the time required for the radio wave output from the transmitting antenna 14 to travel the round trip distance 2L to the object 10 before returning to the receiving antenna 16 is t2.

The received signal received by the receiving antenna 16 is, for example, in the case of the two reflected waves a and b in FIG. 10, where the spectrum of the PN code for transmission appears at both sides around 60 GHz. The spectrum is a composite of a and b (see FIG. 3A). Also in the following description, the spectra for the reflected waves a and b in FIG. 10 are shown in FIG. The spectrum shown in FIG. 3 has a frequency in the horizontal direction and power in the vertical direction.

As shown in FIG. 2, the receiver 17 includes an oscillator 171 for generating a signal of a predetermined frequency (for example, a frequency of 59 GHz band), and a mixing circuit (hereinafter, referred to as a mixing circuit) for mixing the radio wave and the signal from the oscillator 171. 172, and a mixer 1 for mixing the signal from the mixer 172 and the PN code from the delay circuit 15.
73 and a filter 174 that passes only the signal component of a predetermined narrow band among the signal components from the mixer 173.

In the mixer 172 of this embodiment, 60 GHz
The received signal in the band (see FIG. 3A) and the signal in the 59 GHz band are mixed to output a signal with an intermediate frequency in the 1 GHz band (see FIG. 3B) that is easy to handle. Further, in the mixer 173, the spread signal of the above 1 GHz band and the PN
The code is mixed and despread, and the reflected wave a,
Considering the case of b, if correlation is found here, 1
The signal having a sharp peak at the intermediate frequency in the GHz band cannot be correlated and has no sharp peak, and outputs a composite signal of a signal that is still diffused, and the filter 174 passes only the signal component in the 1 GHz band. It is output to the down converter 18.

The down converter 18 converts the signal in the intermediate frequency band input from the receiving unit 17 into a low frequency by dividing it into a plurality of stages by a plurality of mixers. That is, the down converter 18 of the present embodiment uses a predetermined frequency (for example, 0.93).
Oscillator 181 for generating a signal in the GHz band)
A first stage mixer 182 for mixing the signal from the receiving unit 17 and the signal from the oscillator 181; and the mixer 18
Among the signal components from 2, the filter 183 that passes only the signal component in the predetermined narrow band, the amplifier 184 that amplifies the signal component from the filter 183, and the signal of the predetermined frequency (for example, the frequency in the 70.025 MHz band) are generated. The oscillator 185, the signal from the amplifier 184 and the oscillator 185.
Second-stage mixer 186 for mixing signals from the
, A filter 187 that passes only a signal component in a predetermined narrow band among the signal components from the mixer 186, and diodes D1 and D2 that shape the waveform of the signal from the filter 187.
And resistors R1 and R2.

In the mixer 182 of this embodiment, the receiving section 17
1GHz band signal from the oscillator and 0.93 from the oscillator 181
The signal in the GHz band is mixed and the low-frequency composite signal in the 70 MHz band is output to the filter 183. The filter 183 receives 70 MHz of the input beat signal component.
It is composed of a bandpass filter (BPF) that passes only the z-band signal component and outputs it to the amplifier 184.

The amplifier 184 amplifies the input signal,
The mixer 186 produces a signal having the spectrum shown in FIG.
Is output to. The oscillator 185 is 70.025MH
z, that is, a signal having a frequency of 70 MHz + 25 kHz is generated and output to the mixer 186, but 70 MHz−
It can also be configured to generate a frequency of 25 kHz = 69.975 MHz.

In mixer 186, 7 from amplifier 184
Signal of 0MHz band and 70.025M from oscillator 185
The frequency signal of Hz is mixed, and a low-frequency signal of 25 KHz band (see FIG. 3E) is output to the filter 187. In this case, for an approaching object,
The output frequency is Doppler-shifted in the direction of lowering from the center frequency. Further, when the signal output from the oscillator 185 has a frequency in the 69.975 MHz band, the output frequency of an approaching object is Doppler-shifted in the direction of increasing from the center frequency.

Therefore, in the present embodiment, the frequency of the signal output from the oscillator 185 can be selected depending on whether it is desired to accurately detect an approaching object or to detect it quickly. That is, if the frequency from the oscillator 185 is set to 70.025 MHz, the output frequency becomes lower than the center frequency, one cycle becomes longer, and the count value in the arithmetic processing unit increases, so that the detection of the relative speed becomes high. Become. The frequency from the oscillator 185 is 69.97.
When the frequency is set to 5 MHz, the output frequency becomes higher than the center frequency, and one cycle becomes shorter, which speeds up object detection.

In this embodiment, the frequency f0 is 60G.
Assuming a relative velocity of ± 200 Km / h with a carrier of Hz, there is a Doppler shift of ± 22 KHz, so the center frequency is 25 KHz and the Doppler shift range is 47 KHz.
Set to 3KHz, etc. The filter 187 passes only the narrow-band signal component of the 25 kHz band among the input signal components and outputs it to the BPF or L that is output to the amplifier 188.
It is composed of PF. As a result, the signal components of the other frequency bands have their spectrum levels lowered as shown in FIG. 3 (f) and are output to the amplifier 188.

Incidentally, FIGS. 3 (e) and 3 (f) show 25 kHz.
It is a magnified view of the vicinity, where signals with concentrated power (25K
Hz) and a part of the frequency component of the signal that has been spread. In addition, in this embodiment, as shown in FIG.
As shown in (a), the frequency component fα in which the reflected wave from a certain object is despread and correlated and the power is concentrated, the spectra fβ and fγ of the reflected wave from another object, which is still diffused in a wide band. , Fδ and other spectra and the frequency shift due to the Doppler effect causes the spectra f to f
β, fγ, fδ and other spectra become noise components.

That is, in FIG. 4A, when the certain object has a relative velocity, the frequency of the frequency component fα is shifted by the Doppler effect. Δf1 is the shift range of the reflected wave from a certain object due to the Doppler effect. Further, when the other object has a relative velocity, the spectra fβ, fγ, fδ which remain diffused in the wide band and all other spectra are shifted by the Doppler effect.
Δf2 is the shift range of the reflected wave from another object due to the Doppler effect. Strictly speaking, the Doppler shift amount of each spectrum spread over a wide band is different, but it can be considered to be almost the same value.

When the two objects have different relative velocities, f
In order that α and fβ do not overlap, as is apparent from FIG. 4A, the repetition frequency fe of the PN code string is Δf1 + Δ.
It is enough if it is sufficiently larger than f2. That is, fe> 2
It suffices that Δf = Δf = (Doppler shift range by relative velocity in the measurement range of carrier frequency f0) / 2.

If this condition is satisfied, if this combined signal is passed through a filter in the band of ± Δf1 centering on fα when the relative velocity is 0, only fα will pass and the performance will be improved by the reflected waves from other objects. It is no longer restricted by. The above condition is satisfied when the center frequency after down conversion is higher than the band of the spread signal.

When the frequency after the down conversion is lower than the band of the spread signal (when the circuit configuration of the present invention is realized, the frequency is down converted to a low frequency so that the logic circuit can count it. This is true of the case), and as shown in FIG. 4B, the spectrum of the component whose frequency is lower than the center of the spread signal is DC (0
Since it is folded back from Hz) and overlaps with the spectrum of a high-frequency component, it is necessary to satisfy a stricter condition than the above condition.

That is, even if the spectra of the components of the diffused signal appearing adjacent to fα, for example, fγ and fδ are shifted from fα by the Doppler shift amount Δf3,
It is necessary to be sufficiently separated from the Doppler shift range Δf1 of α. In other words, fα + Δf1 <fδ-Δf3 fα-Δf1> fγ + Δf3 Further, when there is no frequency component lower than fα as shown in FIG. Need to do so.

In this embodiment, in order to satisfy this, the chip length is about 40 ns and the cycle length is 255.
3 (e) and 3 (f) using the code of the chip (10 μs).
The frequency component of the spread signal is 75 kHz, 1
It is set to appear at 25 kHz, 175 kHz, .... Even if the 25 kHz component is +22 kHz Doppler-shifted to 47 kHz and the 75 kHz component is −22 kHz Doppler-shifted to 53 kHz, they do not overlap each other.
By attenuating the z component or more, only the necessary component (25 kHz component) can be extracted.

There is a small amount of noise, etc., that oscillates the oscillation frequency component of the oscillator 181 into the mixer 186. Since such noise is input to the arithmetic processing unit 11 in the same manner as a signal having a correlated and concentrated power, An erroneous determination may occur in the arithmetic processing unit 11. Therefore, the amplifier 188 amplifies the input signal.
When set to the center, if the gain of the signal is large, noise due to the above-mentioned sneak occurs in the output, so the center of the output is set to a level lower than 0V. Therefore,
The amplifier 188, when correlation is obtained in the receiving unit 17,
As shown in FIG. 2, when a pulse containing positive and negative components is output to the diode D1 and there is no correlation,
The noise of the negative component is output to the diode D1.

The diode D1 removes the negative component of the input signal, and the Zener diode D2 removes the positive component of, for example, 5 V or more of the above signal to perform waveform shaping, and when the correlation is obtained, When a pulse signal of 0 to 5 V is output to the arithmetic processing unit 11 and there is no correlation,
The pulse signal will not be output. The arithmetic processing unit 11 recognizes that the correlation is obtained by the input of the pulse signal from the down converter 18, and detects the presence of the object. When the delay time is shifted and the correlation starts to be obtained, the pulse signal is output or disappears and is disturbed. Therefore, the arithmetic processing unit 11 measures the pulse signal for several cycles, By confirming that stable counting is possible, it is judged that the correlation has been obtained.

The arithmetic processing unit 11 also receives the received signal and PN.
The output of the PN code by the delay circuit 15 is controlled to be delayed by a predetermined time so that the codes can be correlated with each other. In the arithmetic processing unit 11 of the present embodiment, the delay time is set somewhat coarsely at the initial stage of control, and the correlation between the received signal and the PN code as shown in FIGS. 5A and 5B begins to be obtained. That is, when the pulse signal from the down converter 18 starts to rise, the delay time is finely set and the delay circuit 15 is controlled.

Further, the arithmetic processing section 11 measures the distance to the correlated object. The distance measuring method is as follows.
The beginning of correlation between the received signal and the PN code is shown in FIG.
(B), (c) the end of the correlation, that is, the delay time at the middle of the start of the pulse signal from the down converter 18 starts to disappear (the middle point of the correlation shown in FIG. 5 (d)) Is used as the distance information, and the distance to the object is measured according to the distance information.

The range of the delay time that can be correlated is P
Since the time is one chip length of N code or less, there is also a method of calculating the distance from the start (or end) of the correlation, for example, based on the time-corrected time of half chip length. Further, the arithmetic processing unit 11 also detects the relative speed of the correlated object. In the present embodiment, the operation processing unit 1 is performed by the down converter 18 without returning the received signal to the baseband.
The frequency is converted into a frequency that can be easily counted by a digital counter in 1 and the arithmetic processing unit 11 counts the period of the signal (pulse signal obtained by waveform-shaping the received signal) to obtain the relative speed. In the detection of the relative speed, the arithmetic processing unit 11 determines that the cycle has a center frequency (in the present embodiment, 2).
The moving direction and the relative speed of the object can be obtained by obtaining the shift amount from 5 kHz). That is, in this embodiment, the moving direction and the relative speed of the object are detected by obtaining the Doppler shift amount and the Doppler shift direction of the measured frequency with respect to the set center frequency.

In an actual circuit, the relative speed cannot be detected by measuring the cycle unless the signal input to the arithmetic processing unit for measuring the cycle is accurately down-converted. However, in reality, the deviation of the oscillation frequency,
Due to temperature characteristics, changes over time, etc., errors occur in the down converter. Therefore, a means for automatically correcting this error is required, but it can be easily performed by the following method.

For example, since the frequency difference between the oscillators of 70 MHz and 70.025 MHz determines the frequency when input to the arithmetic processing unit, these two frequencies are mixed,
A generation unit that generates the difference and a period measurement unit that measures the period of the frequency of the difference are provided. It should be noted that this cycle measuring means can be carried out by simply devising a part of means originally provided for signal detection.

This difference signal is the center frequency of the original signal,
That is, since it is the frequency when the relative speed is 0, if this is input to the arithmetic processing unit, the correction by the arithmetic can be easily performed. Of course, it is also possible to make one of the oscillators variable in frequency and control so that the difference in frequency is always constant. Therefore, in the present embodiment, the received signal spread in a wide band is not converted to the baseband, but the signal is generated when the signal is down-converted into the low frequency band that is easy to measure and the correlation is obtained, and the waveform is shaped. Since the pulse signal is generated, the distance to the object can be measured from the correlated delay time, and at the same time, the relative velocity of the object can be measured from the period of the pulse signal. As a result, in this embodiment, the measurement time can be shortened.

Further, in the present embodiment, even when a plurality of reflected waves are received, only the signal in which the power from the correlated object is concentrated can be taken out by the narrow band filter and the signal can be detected. Due to the diffused signal (noise) that cannot be removed, it is possible to reduce detection performance deterioration or restriction. In the present embodiment, the moving direction of the object and the shift amount of the converted frequency are related to each other according to the setting method of the frequency mixed with the received signal by the mixer. The moving direction of the object can be easily recognized by the shift amount of the frequency together with the speed.

Further, in this embodiment, an expensive integrator and A /
Since the circuit can be configured without using a device such as a D converter, the manufacturing cost can be reduced. In this embodiment, the received signal is despread and then down-converted. However, the present invention is not limited to this, and the received signal is converted to a low-frequency signal before despreading. It is also possible.

Further, in the delay circuit of the present embodiment, when the correlation is detected, the range of the delay time corresponding to the preset distance is shifted, but the present invention is not limited to this and, for example, is set in advance. It is also possible to fix the corresponding delay time of the distance, in which case an object at a specific distance can be detected. Further, the code setting condition of the PN code and the pass band setting condition of the narrow band filter shown in the present embodiment are not limited to the radar device having the configuration of the above embodiment, and may be, for example, P
The radio wave whose band is spread by the N signal is transmitted, the reflected wave from the object based on the radio wave is received, the correlation between the received signal and the PN signal is detected, and the correlated signal is converted to the baseband. It can also be used in a radar device that detects the presence of an object, measures the velocity, or measures the distance to the object. Even in this case, even if a plurality of reflected waves are received, only the signal in which the power is concentrated can be extracted by the narrow band filter and the signal can be detected, so that the detection performance is not deteriorated or restricted.

In the above embodiment, the narrow band filter is assumed to be a general band pass filter or low pass filter. Therefore, when some noise is mixed in the pass band of the filter, the measurement is erroneous. there is a possibility. Therefore, the present invention provides another embodiment of the radar device as shown in FIG. In the figure, the radar device of the present embodiment is similar to the device of FIG. 1 in that the arithmetic processing unit 11, the PN generator 12, the transmitting unit 13, the transmitting antenna 14, the delay circuit 15, the receiving antenna 16, the receiving unit 17, and the down unit 17. The converter 18 is provided, and in addition, a PLL circuit 19 that functions as a narrow band filter is provided.

The functions of the arithmetic processing unit 11, the PN generator 12, the transmitting unit 13, the transmitting antenna 14, the delay circuit 15, the receiving antenna 16 and the receiving unit 17 shown in FIG.
The description of the down converter 18 will be omitted because it is the same as that of each part of FIG.
85, mixers 182, 186, a filter 183 and an amplifier 184, and outputs a low frequency signal in the 25 kHz band shown in FIG. 3 (e) to the PLL circuit 19. Also in this case, similarly to the above-described embodiment, the frequency of the output signal is Doppler-shifted to the approaching object in the direction of decreasing from the center frequency, and the signal output from the oscillator 185 has a frequency of 69.975 MHz band. In Case of,
For an approaching object, the output frequency is Doppler-shifted in the direction of increasing from the center frequency.

Therefore, in the present embodiment, the frequency of the signal output from the oscillator 185 can be selected depending on whether it is desired to accurately detect an approaching object or to detect it quickly. Also in this embodiment, assuming a relative speed of ± 200 Km / h on a carrier whose frequency f0 is 60 GHz, there is a Doppler shift of ± 22 KHz, so the center frequency is set to 25 KHz and the Doppler shift range is set to 47 KHz to 3 KHz. To do.

As shown in FIG. 7, the PLL circuit 19 compares the input signal component with the signal component fed back from the VCO 193, which will be described later, and outputs a difference between the phase comparator 191 and the narrow band signal component. A loop filter 192 for controlling, a VCO 193 that outputs a signal component synchronized with the input signal component, and a feedback loop 194 from the VCO 193 to the phase comparator 191 are included, and 3 KHz to 47 KHz of the input signal component are included.
It is set to lock only the narrow band signal component of.

Also in this embodiment, when the center frequency after down conversion is higher than the band of the spread signal, FIG.
As is clear from the above, when fe> 2 · Δf, Δf =
(Doppler shift range due to relative velocity in measurement range of carrier frequency f0) / 2, and relative velocity 0
This composite signal is passed through a filter in the band of ± Δf1 centering on fα at the time of, and only fα is passed through, and the restriction on the performance is eliminated by the reflected waves from other objects.

When the frequency after the down conversion is lower than the band of the spread signal, the spectrum of the component whose frequency is lower than the center of the spread signal is as shown in FIG. 4 (b).
Since it folds back from DC (0 Hz) and overlaps with the spectrum of the high frequency component, the above fα + Δf1
The condition of <fδ-Δf3, fα-Δf1> fγ + Δf3 is satisfied.

Further, when the frequency component lower than fα is set as shown in FIG. 3E, the DC level is not reached even if the Doppler shift is performed in the lower fα direction. In addition, also in the present embodiment, in order to satisfy this, a code having a chip length of about 40 ns and a cycle length of 255 chips (10 μs) is used, and the frequency components of the signal spread as shown in FIG. Is 75kHz, 125kHz, 17
It is set to appear at 5 kHz, .... 25kHz component is + 22kHz Doppler shift to 47kHz, 7
5kHz component shifts by -22kHz Doppler and 53kHz
It is set so that they do not overlap each other even when the frequency becomes Hz, and the PLL circuit 1 having a lock-in range of 3 kHz to 47 kHz or having a lock range is input to the PLL circuit 1 based on a necessary frequency component.
9, the output signal frequency of the VCO 193,
Alternatively, the output voltage of the loop filter 192 can be taken out.

The arithmetic processing unit 11 recognizes that the PLL circuit 19 is in the locked state and the correlation is obtained, and detects the presence of the object. When the delay time is shifted and the correlation starts to be obtained, the PLL circuit 19 is locked or released from the locked state. Therefore, in the arithmetic processing unit 11, the PLL circuit 1 is operated for a predetermined time.
By confirming that the lock state of No. 9 continues, it is determined that the correlation is obtained.

The arithmetic processing section 11 also receives the received signal and PN.
The output of the PN code by the delay circuit 15 is controlled to be delayed by a predetermined time so that the codes can be correlated with each other. In the arithmetic processing unit 11 of the present embodiment, the delay time is set somewhat coarsely at the initial stage of control, and the correlation between the received signal and the PN code as shown in FIGS. 5A and 5B begins to be obtained. That is, when the pulse signal from the down converter 18 starts to rise, the delay time is finely set and the delay circuit 15 is controlled.

Further, the arithmetic processing section 11 measures the distance to the correlated object. The distance measuring method is as follows.
The beginning of correlation between the received signal and the PN code is shown in FIG.
The end of the correlation as shown in (b) and (c), that is, P
The delay time in the middle of the lock state of the LL circuit 19 from the beginning to the end (the midpoint of the range shown in FIG. 5D) is used as the distance information, and the distance to the object is measured according to the distance information. There is.

The range of the delay time which can be correlated is P
Since the time is one chip length of N code or less, there is also a method of calculating the distance from the start (or end) of the correlation, for example, based on the time-corrected time of half chip length. Further, the arithmetic processing unit 11 also detects the relative speed of the correlated object. In this embodiment, the received signal is not returned to the baseband, the downconverter 18 converts it to a low frequency, the converted signal is input to the PLL circuit 19, and the PLL circuit 19 is locked. VCO1
The arithmetic processing unit 11 counts the cycle of the output of 93, or the arithmetic processing unit 11 calculates the output voltage of the loop filter 192 as A /
The relative speed is obtained by D conversion. In the detection of the relative speed, the arithmetic processing unit 11 moves the object if the period or the voltage shifts from the period or the voltage corresponding to the center frequency (25 KHz in this embodiment) to which direction. Direction and relative velocity can be determined. That is, in this embodiment, the moving direction and the relative speed of the object are detected by obtaining the Doppler shift amount and the Doppler shift direction of the measured frequency with respect to the set center frequency.

In an actual circuit, the relative speed cannot be detected by measuring the period or voltage unless the signal input to the arithmetic processing unit for measuring the period or voltage is accurately down-converted. However, in reality, an error occurs in the down converter due to the deviation of the oscillation frequency, the temperature characteristic, the change over time, and the like. Therefore,
A means for automatically correcting this error is required, but it can be easily performed by the following method.

For example, since the frequency difference between the oscillators of 70 MHz and 70.025 MHz determines the frequency when input to the arithmetic processing unit, these two frequencies are mixed,
A generation unit that generates the difference and a period measurement unit that measures the period of the frequency of the difference are provided. It should be noted that this cycle measuring means can be carried out by simply devising a part of means originally provided for signal detection.

This difference signal is the center frequency of the original signal,
That is, since it is the frequency when the relative speed is 0, if this is input to the arithmetic processing unit, the correction by the arithmetic can be easily performed. Of course, it is also possible to make one of the oscillators variable in frequency and control so that the difference in frequency is always constant. Therefore, in the present embodiment, the received signal spread in a wide band is not converted into the base band, but is down-converted into a low frequency band that can be easily measured, and when the correlation is obtained, the PLL circuit 19 is locked and the VCO 193 or the loop is obtained. Since the signal of the filter 192 is taken out,
The distance to the object can be measured from the correlated delay time, and at the same time, the relative speed of the object can be measured from the cycle or voltage of the output signal of the PLL circuit 19.
As a result, in this embodiment, the measurement time can be shortened.

In the relative velocity measurement of the present invention, as shown in FIG. 8, the VCO 193 to the phase comparator 191 are used.
It is also possible to insert a frequency divider 195 in the feedback loop 194 to the VCO 19 for changing the input.
3 and the output change of the loop filter 192 become large, so that the measurement accuracy can be improved. Further, in the present embodiment, even if a plurality of reflected waves are received, the PLL circuit locks only on the signal in which the power from the correlated object is concentrated, and the output signal of the PLL circuit based on the signal is detected. Therefore, it is possible to reduce the deterioration of the detection performance or the restriction of the detection performance due to a signal (noise) in a diffused state that cannot be correlated.

Further, in the PLL circuit of the present embodiment, when signals of different frequencies are present in the lock range, the PLL circuit is characterized in that it locks to the one with larger power, so that the power is smaller than the signal required within the lock range. Stable detection is possible even if noise is mixed. Further, in this embodiment, the PL is used without using expensive integrators, A / D converters, and other devices.
Since the circuit can be configured by a general-purpose dedicated IC that configures the L circuit, the manufacturing cost can be reduced.

In the above embodiment, the arithmetic processing unit 11 measures the time of one cycle of the input signal to measure the relative speed of the object. Therefore, the detection accuracy and the detection time differ depending on the relative speed of the object. That is, the frequency from the oscillator 185 is set to 7
When the frequency is set to 0.025 MHz, the period of the signal input to the arithmetic processing unit 11 becomes longer for an approaching object, so that the detection time becomes longer, but the arithmetic processing unit 1
The count value at 1 increases, and the detection accuracy (resolution) increases. On the other hand, for a distant object, the cycle of the input signal becomes shorter and the detection time becomes faster, but the detection accuracy (resolution)
Will be lower. In this way, if the detection time and the detection accuracy (resolution) differ depending on the period of the signal to be detected, inconvenience may occur depending on the application of the radar device.

Therefore, the present invention will be described below with reference to another embodiment of a radar device for solving this problem. The radar device of this embodiment is similar to the device of FIG.
N generator 12, transmitter 13, transmission antenna 1
4, delay circuit 15, receiving antenna 16, receiving unit 17,
It is configured to include a down converter 18 and a PLL circuit 19.

The PN generator 12 in this embodiment,
The functions of the transmission unit 13, the transmission antenna 14, the delay circuit 15, the reception antenna 16, the reception unit 17, the down converter 18, and the PLL circuit 19 are the same as those of the respective units in FIG. As shown in FIG. 9, is composed of an edge detection circuit 11a, a timer circuit 11b, counter circuits 11c and 11d, a clock circuit 11e that outputs a clock signal, and a processing circuit 11f, and calculates the length of one cycle. There is.

The edge detection circuit 11a includes a PLL circuit 19
When the edge of the input signal from the PLL circuit 19 is detected, the pulse signal is output to the timer circuit 11b and the counter circuit 11c when the edge is detected. The direction of the edge detected by the edge detection circuit 11a may be rising or falling. Further, the edge detection circuit 11a is not impossible to function even when detecting both edges, but since a pulse is output every half cycle of the input signal and measurement is performed based on this, the duty ratio of the input signal is accurate. Not suitable because it must be 50%.

The timer circuit 11b is the edge detection circuit 11
a and the processing circuit 11f, the edge detection circuit 11
The pulse signal is input from a and the reset signal is input from the processing circuit 11f. Then, while the reset signal is valid (low level in this example), the timer circuit 11b is in a reset state, and its output is also fixed in an inactive state (low level in this example). Further, when the pulse signal is input once from the edge detection circuit 11a after the reset signal becomes invalid (high level in this example), the output of the timer circuit 11b becomes active (high level in this example). And time measurement is started. Timer circuit 11
The output of b is input to the counter circuits 11c and 11d as a signal instructing the counting operation. Then, the timer circuit 11b, after the preset predetermined time has elapsed,
When the pulse signal from the edge detection circuit 11a is input,
Return the output to the inactive state. The predetermined time is the lowest frequency of the input signal, that is, 3k in the above example.
Sufficient time to measure Hz, eg 1/3 kHz = 3
The time is set to 33 μs or more and not too long.

The counter circuit 11c is the edge detection circuit 1
1a, the timer circuit 11b and the processing circuit 11f are connected, and the respective output signals from the edge detection circuit 11a and the timer circuit 11b and the reset signal from the processing circuit 11f are input. The counter circuit 11d includes a timer circuit 11b,
It is connected to the processing circuit 11f and the clock circuit 11e, and the output signal is input from the timer circuit 11b and the reset signal is input from the processing circuit 11f and the clock circuit 11e.

In both counter circuits 11c and 11d, the count is cleared while the reset signal is valid, and the outputs thereof are fixed at a state showing "0". When the reset signal becomes invalid and the output of the timer circuit 11b is active, that is, when the pulse signal is input once from the edge detection circuit 11a, the counter circuit 11c causes the edge detection circuit 11c to operate.
The counter circuit 11d starts counting the pulse signal from a, and the counter circuit 11d starts counting the clock signal from the clock circuit 11e.

After that, the output of the timer circuit 11b is inactive, that is, after a lapse of a predetermined time, the edge detection circuit 11
When the pulse signal is input again from a, both counter circuits 11c and 11d stop counting, hold the count value up to that point, and output the count value to the processing circuit 11f. The processing circuit 11f is connected to the timer circuit 11b and the counter circuits 11c and 11d, receives the output signal from the timer circuit 11b, the respective count values from the counter circuits 11c and 11d, and the timer circuit 11b and the counter circuits 11c and 11d. The reset signal is output to.

The processing circuit 11f monitors the output signal from the timer circuit 11b. When the processing circuit 11f deactivates the reset signals to the timer circuit 11b and the counter circuits 11c and 11d, the output signal of the timer circuit 11b becomes active, and then returns to inactive, the counter circuit 11c detects an edge. The count value of the pulse and the count value of the clock signal are fetched from the counter circuit 11d. Then, the processing circuit 11
For f, the count value of the clock signal is divided by the count value of the edge detection pulse to calculate the length of one cycle.

Next, the processing circuit 11f can find the moving direction and the relative speed by finding how much the length of one cycle shifts from the center frequency. Therefore, in this embodiment, the length of one cycle and a plurality of cycles of the frequency-converted signal at a certain predetermined time is counted, and the length of one cycle is obtained from the number of the above cycles and the count value. The number of counting cycles can be made variable in accordance with the signal cycle, which makes it possible to obtain a certain range of detection accuracy (resolution) and detection time regardless of the cycle of the input signal.

[0069]

As described above, according to the present invention, a radio wave whose band is spread by a pseudo noise signal is transmitted, a reflected wave from an object based on the radio wave is received, and the received signal and the pseudo noise signal are received. In the radar device for detecting the correlation by detecting the correlation of the object, despreading means for despreading the received signal with the pseudo noise signal, frequency conversion means for converting the frequency of the received signal into a low frequency signal, Identification means for identifying the presence of the object based on the frequency-converted signal, measuring means for measuring the cycle of the frequency-converted signal, and detection for detecting the speed of the object according to the measured cycle Means, delay means for delaying the pseudo-noise signal for a predetermined time when detecting the correlation, and distance measuring means for measuring the distance to the object according to the delay time when the correlation is detected. Since a means for passing the frequency-converted signal in a narrow band is provided, the distance to the object and the relative speed are simultaneously measured by a common circuit, and the measurement time and the number of parts are reduced to reduce the size of the device. It is possible to realize higher efficiency and lower cost and improve the measurement accuracy.

In the eleventh aspect, a radio wave whose band is diffused by a pseudo noise signal is transmitted, a reflected wave from an object based on the radio wave is received, and a correlation between the received signal and the pseudo noise signal is detected. In a radar device that detects the presence of the object, measures the velocity, or measures the distance to the object, the received signal in which the power is correlated and the power is concentrated, and the frequency of the received signal in which the power is correlated and converted to a low frequency is converted. The shift range due to the Doppler effect does not overlap with the shift range of the frequency of the signal which is not correlated and is still diffused, and the minimum value of the shift range of the frequency of the power-concentrated signal is larger than "0". As described above, oscillating means for setting and oscillating the sign of the pseudo noise signal according to the assumed relative velocity of the object, and despreading means for despreading the received signal with the oscillated pseudo noise signal Because with the passage means comprising the despread signal from the narrow band filter that passes a frequency shift range of the Doppler effect of the frequency of the concentration signal, the detection performance is deteriorated, or it is no longer restricted.

According to a twelfth aspect of the present invention, transmitting means for transmitting a radio wave whose band is spread by a pseudo noise signal, receiving means for receiving a reflected wave from an object based on the radio wave, and the received signal by the pseudo noise signal. Despreading means for despreading, and frequency conversion means for converting the frequency of the received signal into a low-frequency signal, and based on the frequency-converted signal, presence detection of the object, speed measurement, or even the object In the radar device that performs the distance measurement, since the PLL circuit that allows the frequency-converted signal to pass in a narrow band is provided, the distance to the object and the relative speed are simultaneously measured by a common circuit, and the measurement time and the number of parts are reduced. It is possible to reduce the size of the apparatus and reduce the cost, improve the measurement accuracy, reduce the mixing of noise in the pass band, and improve the noise resistance.

In the thirteenth and fourteenth aspects, since the PLL circuit has the phase comparator, the loop filter, the VCO, and the feedback loop from the VCO to the phase comparator, the measuring means is the loop filter or the VCO. It is possible to easily detect the presence of an object, measure the speed of the object, or measure the distance to the object based on the output of. In claims 16 and 17, the PLL circuit has a lock-in range or a lock range in an expected frequency displacement range of a signal that is correlated by the despreading means, the power is concentrated, and the frequency is converted by the frequency converting means. The lock-in range or the lock range of the PLL circuit is set in the assumed frequency displacement range of the spectrum which is set and spread without correlation in the despreading means and which is frequency-converted by the frequency converting means. Since they are set so as not to overlap, even if a plurality of reflected waves are received, it is possible to extract only the frequency component in which the power is concentrated.

In the eighteenth aspect of the present invention, the radar device comprises cycle length measuring means for measuring the length of one cycle and a plurality of cycles of the frequency-converted signal in a predetermined time, and the cycle length measuring means converts the frequency. Since the number of cycles to be measured is variable according to the cycle of the input signal, it is possible to obtain detection accuracy and detection time within a certain range regardless of the cycle of the input signal.

[Brief description of drawings]

FIG. 1 is a block diagram showing a schematic configuration of an embodiment of a radar device according to the present invention.

FIG. 2 is a circuit diagram showing configurations of a receiving unit and a down converter shown in FIG.

FIG. 3 is a waveform diagram showing spectra of outputs of respective parts shown in FIG.

FIG. 4 is a spectrum waveform diagram for explaining a code setting condition of a PN code.

FIG. 5 is a waveform diagram for explaining the distance detection operation of the arithmetic processing unit.

FIG. 6 is a block diagram showing a schematic configuration of another embodiment of the radar device according to the present invention.

7 is a diagram illustrating a receiving unit, a down converter, and a P shown in FIG.
It is a circuit diagram showing a configuration of an LL circuit.

FIG. 8 is a block diagram showing the configuration of another embodiment of the PLL circuit.

9 is a block diagram showing the configuration of another embodiment of the arithmetic processing unit shown in FIG.

FIG. 10 is a schematic diagram for explaining the operation of a conventional radar device mounted on an automobile.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 10 Object (automobile) 11 Arithmetic processing section 11a Edge detection circuit 11b Timer circuit 11c, 11d Counter circuit 11e Clock circuit 11f Processing circuit 12 PN generator 13 Transmitting section 14, 16 Antenna 15 Delay circuit 17 Receiving section 18 Downconverter 19 PLL circuit 171 , 181, 185 Oscillator 172, 173, 182, 186 Mixer 183, 187 Filter 184, 188 Amplifier 191 Phase comparator 192 Loop filter 193 VCO D1, D2 Diode

Claims (18)

[Claims] 1. A radio wave whose band is spread by a pseudo noise signal is transmitted, a reflected wave from an object based on the radio wave is received, and a correlation between the received signal and the pseudo noise signal is detected to detect the object. In a radar device for detecting, a despreading unit that despreads a received signal with the pseudo noise signal, a frequency conversion unit that converts the frequency of the received signal into a low-frequency signal, and the frequency conversion unit based on the frequency-converted signal. A radar device comprising: an identification unit that identifies the presence of an object. 2. A radio wave whose band is spread by a pseudo noise signal is transmitted, a reflected wave from an object based on the radio wave is received, a correlation between the received signal and the pseudo noise signal is detected, and the object is detected. In a radar device for detecting, a despreading unit that despreads a received signal with the pseudo noise signal, a frequency conversion unit that converts the frequency of the received signal into a low-frequency signal, and a period of the frequency-converted signal is measured. A radar device comprising: a measuring unit for detecting the velocity of the object according to the measured period. 3. The radar device, when detecting the correlation, delays the pseudo noise signal by a predetermined time and a distance between the object and the object according to the delay time when the correlation is detected. 3. The radar device according to claim 1, further comprising a distance measuring unit that measures the distance. 4. The radar device according to claim 1, wherein the radar device includes a passage unit that passes the frequency-converted signal in a narrow band. 5. The detecting means detects a moving direction and a relative speed of the object according to a Doppler shift amount and a Doppler shift direction of the measured frequency with respect to a set center frequency. Claims 2 to 4
The radar device according to any one of 1. 6. The radar device according to claim 3, wherein the delay means shifts a range of a delay time corresponding to a preset distance when detecting the correlation. 7. The radar device according to claim 3, wherein the delay means fixes a corresponding delay time of a preset distance when detecting the correlation. 8. The radar device according to claim 3, wherein the distance measuring unit measures the distance to the object according to a delay time at an intermediate point between the start and the end of correlation. . 9. The radar device according to claim 3, wherein the distance measuring unit measures the distance to the object according to the delay time at the start point or the end point of the correlation. 10. The pseudo noise signal has a shift range of a frequency of a received signal, which is correlated and converted into a low frequency and in which power is concentrated, due to a Doppler effect, and a shift range of a frequency of a signal which is uncorrelated and remains diffused. It is set according to the assumed relative velocity of the object such that the minimum value of the frequency shift range of the signal in which the power is concentrated does not overlap with the range is larger than “0”, and the passing means is The radar device according to claim 4, wherein frequencies in a band from a maximum value to a minimum value of a frequency shift range of a signal in which power is concentrated are passed. 11. A radio wave whose band is spread by a pseudo noise signal is transmitted, a reflected wave from an object based on the radio wave is received, and a correlation between the received signal and the pseudo noise signal is detected to detect the object. In a radar device that performs presence detection, velocity measurement, or distance measurement to the object, the shift range due to the Doppler effect of the frequency of the received signal in which the correlation and low frequency conversion are concentrated and the power is concentrated and remains diffused. According to the relative velocity of the assumed object so that the minimum value of the frequency shift range of the signal in which the power is concentrated does not overlap with the shift range of the signal frequency of An oscillating means for setting and oscillating a sign of a noise signal; a despreading means for despreading a received signal with the oscillated pseudo-noise signal; and a despread signal of the power-concentrated signal. What is claimed is: 1. A radar device, comprising: a passing unit that passes frequencies in a maximum value band and a minimum value band in a frequency shift range. 12. A transmitting means for transmitting a radio wave whose band is spread by a pseudo noise signal, a receiving means for receiving a reflected wave from an object based on the radio wave, and a despreading of the received signal with the pseudo noise signal. Despreading means and frequency conversion means for converting the frequency of the received signal into a low-frequency signal, and based on the frequency-converted signal, presence detection of the object, speed measurement, or distance measurement to the object In the radar device for performing the above, a radar device including a phase-locked loop circuit that allows the frequency-converted signal to pass in a narrow band. 13. The phase locked loop circuit comprises:
Phase comparator, loop filter, voltage controlled oscillator,
A feedback loop from the voltage controlled oscillator to the phase comparator, wherein the measuring means performs presence detection of the object, speed measurement, or distance measurement to the object based on the output from the loop filter. The radar device according to claim 12, which is characterized in that. 14. The phase locked loop circuit comprises:
Phase comparator, loop filter, voltage controlled oscillator,
A feedback loop from the voltage controlled oscillator to the phase comparator, wherein the measuring means performs presence detection of the object, speed measurement, or distance measurement to the object based on the output from the voltage controlled oscillator. The radar device according to claim 12, wherein: 15. The phase locked loop circuit comprises:
15. The radar device according to claim 12, further comprising a frequency divider in a feedback loop from the voltage controlled oscillator to the phase comparator. 16. The phase-locked loop circuit comprises:
A lock-in range or a lock range is set in an expected frequency displacement range of a signal that is correlated by the despreading means, has a concentrated power, and is frequency-converted by the frequency converting means. Item 16. The radar device according to any one of items 12 to 15. 17. The phase locked loop circuit comprises:
Set so that the lock-in range or the lock range does not overlap with the expected frequency displacement range of the spectrum that has been frequency-converted by the frequency conversion means while being spread without being correlated by the despreading means. The radar device according to any one of claims 12 to 16, wherein: 18. The radar device comprises cycle length measuring means for measuring lengths of one cycle and a plurality of cycles of the frequency-converted signal in a predetermined time, and the cycle length measuring means comprises the frequency-converted signal. 18. The radar device according to claim 2, wherein the number of the cycles to be measured is made variable according to the cycle.