JPH0819253A - Single-element composite current resonance converter circuit - Google Patents

Abstract

PURPOSE:To reduce a switching loss by a method wherein a DC power supply is closed and opened with one switching device and, further, a resonance current is applied to the switching device. CONSTITUTION:An inductor L21 is connected in parallel to a series resonance circuit composed of an insulating transformer T21 and a capacitor C21 and a switching means which closes and openes a DC power supply supplied to the above-mentioned components is provided. When a switching device Q21 is turned off, the inductor L21, a primary coil L22 and the capacitor C21 make resonance to reduce the switching loss when the DC power supply is closed next.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a DC-DC conversion circuit used in power supply circuits of general electronic equipment or lighting equipment.

[0002]

2. Description of the Related Art As a conventional example of a switching power supply used as a power supply for electronic equipment, FIG. 5 shows a circuit diagram of a flyback converter circuit. In this figure, E is a DC voltage source, T ₁ is an insulating transformer, L ₁ and L ₂ are winding coils of the insulating transformer T ₁ , Q ₁ is a switching element,
1 is a differential amplifier circuit, 2 is a control circuit, D ₁ is a diode,
C ₁ is a capacitor.

The operation of this circuit will be briefly described. The switching element Q ₁ is turned on / off by a pulse signal or the like from the control circuit 2. Further, the circuit period on the switching element Q ₁ is, transformer T ₁ does not perform the power transferred to the secondary circuit of the transformer T ₁ by simply storing energy, transformer during the off of the switching element Q ₁ The electromagnetic energy of T ₁ is transmitted to the secondary side circuit as a back electromotive force. Then, an AC voltage generated in the secondary winding L ₂ of the transformer T ₁ is rectified by the diode D _1, is outputted as a DC voltage is smoothed by the capacitor C _1. Further, this output voltage is detected by the error amplification circuit 1, and by applying the detection signal to the control circuit 2, the control circuit 2 applies, for example, a pulse to the switching element Q ₁ so that the output voltage becomes constant. Controls the pulse width of the signal.

However, the above-described circuit is for turning on / off the current I ₁ flowing through the primary winding coil L ₁ of the transformer T ₁ at the switching element Q _1, the voltage V ₁ and current I according to the switching element Q _{1 1} has a waveform as shown in FIG. 9A, and the waveform of the current I ₁ flowing when the switching element Q ₁ is on has a sawtooth shape. Therefore, the voltages V ₁ to transient period the switching element to Q ₁ on / off operation switching element Q _1, the current I ₁ is generated, the switching losses may period applied simultaneously. In particular, when the switching element Q ₁ is turned off, the voltage V ₁ rapidly rises at the point where the current I ₁ is maximum, which causes a drawback that large switching loss and switching noise occur.

Therefore, a quasi-resonant flyback converter circuit is known as a circuit in which these problems are slightly improved, and its circuit diagram is shown in FIG. In this figure, T ₂ is an insulating transformer, L ₃ , L ₄ , and L ₅ are winding coils of the insulating transformer T ₂ , D ₂ is a diode, and 3 is ON.
It is a delay circuit and is otherwise the same as the flyback converter circuit of FIG. 5 described above.

The operation of this circuit is the same as that of the flyback converter circuit described above, but the switching element Q
When _1-on, to use the voltage of the secondary winding L ₅ of the transformer T _2, switching elements falling for Q ₁ of voltage becomes resonant, the voltage V ₁ and current I according to the switching element Q _{1 1} has the waveform shown in FIG. 9 (b). As a result, when the voltage V ₁ falls, the curve becomes a gentle curve, and the switching loss is slightly reduced as compared with the circuit described above. However, the loss in the rise of the voltage V ₁ , that is, in the section where the switching element Q ₁ is turned off, remains large.

Further, the above-mentioned flyback type circuits of FIGS. 5 and 6 do not utilize the leakage inductance of the insulating transformer, so that the structure of the insulating transformer becomes complicated and the influence of the leakage magnetic flux becomes large. Further, since the currents I ₁ and I ₂ flowing in the switching element Q ₁ are sawtooth and the peak value of the current is large, the field effect transistor (FE
When T) is used, power loss due to the internal resistance of the FET becomes a problem. Further, when a bipolar transistor is used, it is difficult to set drive conditions and the drive circuit becomes complicated.

In order to solve these problems, a partial voltage resonance type forward type converter circuit is known, and its circuit diagram is shown in FIG. In this circuit, T
₃ is an insulating transformer whose polarity is opposite to that of the flyback circuit transformer, and L ₆ , L ₇ , and L ₈ are transformers T _3.
Winding coil, 4 is a sub-control circuit, S is a switch, C ₃ is a capacitor, D ₂ and D ₃ are diodes, L ₉ is a choke coil, and the others are the same as the above-mentioned circuits.

To briefly explain the operation of this circuit, the switching element Q ₁ is connected to the control circuit 2 via the ON delay circuit 3.
It is turned on / off by an applied signal from. Then, during the ON period of the switching element Q ₁ , power is transmitted to the secondary coil L ₇ of the transformer T ₃ , is rectified by the diode D ₂ , and smoothed by the capacitor C ₁ via the choke coil L ₉ to generate a voltage. Output. Further, during the off period of the switching element Q ₁ , the diode D ₃ conducts and the coil L ₉
The energy remaining in is output as the output voltage.

[0010] waveforms of the voltage V ₁ and current I ₁ in the switching element to Q ₁ this circuit, as shown in (c) of FIG. 9, the rise of the voltage, and both falling in a resonance type. Therefore, the switching loss can be further reduced as compared with the above-mentioned quasi-resonant type flyback circuit, and the voltage applied to the switching element Q ₁ can be made substantially constant with respect to the input voltage of the DC voltage source E and the load fluctuation. However, this circuit has a drawback that an expensive switch S that can newly cope with a high breakdown voltage and a large current is required.

In order to solve these problems, a half-bridge type current resonance type converter circuit is known, and its circuit diagram is shown in FIG. In this circuit,
Q ₂ , Q ₃ are switching elements, C ₄ is a capacitor, T
₄ is a transformer, L ₁₀ , L ₁₁ and L ₁₂ are winding coils of the transformer T ₄ , D ₄ and D ₅ are diodes, and others are the same as those in the circuit described above.

The operation of this circuit is based on the switching element Q.
₂ and Q ₃ are turned on / off by an applied signal from the control circuit 2. And the secondary coil L _{11 of} the transformer T ₄ ,
Power is transmitted to L ₁₂ , rectified by diodes D ₄ and D ₅ , smoothed by smoothing capacitor C ₁ and output. The output voltage is detected by the error amplification circuit 1, and the switching elements Q ₂ and Q ₃ are controlled by the control circuit 2 so that the output voltage becomes constant.

The current I of the switching element Q ₂ of this circuit
The waveforms of ₁ and the voltage V ₁ are as shown in FIG. 9 (d), and the leakage inductance of the transformer T ₄ is used to make the current flowing through the switching element Q ₂ into a plurality of resonance waveforms, which further reduces the switching loss. Will be reduced. Further, since the voltage applied to the switching element Q ₂ can be kept substantially constant and small with respect to variations in the input voltage and the load, it is possible to cope with a relatively large output. Further, since the voltage applied to the switching element Q ₂ is the voltage of the DC voltage source E even at the maximum, a withstand voltage lower than that of the partial voltage resonance type converter circuit described above can be used.

[0014]

By the way, the above-mentioned half-bridge type current resonance converter circuit of FIG. 8 requires two switching elements, resulting in an increase in cost. Although not shown, the number of peripheral circuit parts is increased and the number of integrated circuits (ICs) used in the control circuit is small.

[0015]

The present invention has been made in view of the above-mentioned problems, and an inductor is connected in parallel to a series resonance circuit composed of an insulating transformer and a capacitor. And a switching means for connecting and disconnecting the DC power supply supplied to the series resonance circuit and the inductor,
The secondary side circuit voltage obtained thereby is rectified / smoothed and output.

The above-mentioned switching means is composed of a switching element, a control circuit, and an error amplifier circuit, and the switching element is a bipolar transistor, a field effect transistor, or an insulated gate bipolar transistor (Insulated Gate Bipolar Transistor).
). Further, the coupling of the insulation transformer is loosely coupled because the leakage inductance is utilized. For example, a coil is connected in series to the primary side winding coil of the insulation transformer to obtain a resonance inductance. Good. Further, the inductor is a choke coil having a single winding, but may be composed of a plurality of windings that are transformer-coupled.

[0017]

With the above configuration, the current flowing through the switching element is a current in which a plurality of resonance currents are superposed, and the switching loss can be reduced.

[0018]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 is a circuit diagram of an embodiment of a one-stone current resonance type converter circuit of the present invention. In this figure, E is a DC voltage source, L ₂₁ is an inductor, C ₂₁ is a capacitor, Q ₂₁
Is a switching element controlled by a pulse signal from the control circuit 22, T ₂₁ is a transformer, L ₂₂ and L ₂₃ are winding coils on the primary and secondary sides of the transformer T ₂₁ , and D ₂₁ is a rectifying diode. , C ₂₂ is a smoothing capacitor, 2
Reference numeral 1 is an error amplification circuit that detects a change in the output voltage, and reference numeral 22 is a control circuit that can change the pulse width, for example. Also, I
₂₁ is a current flowing through the inductor L ₂₁ , I ₂₂ is a capacitor C
₂₁ and the current flowing through the primary winding coil L ₂₂ , I ₂₃ is the current flowing through the switching element Q ₂₁ (I ₂₁ + I ₂₂ ), I ₂₄
Is a diode D connected to the secondary winding coil L _23
The currents flowing in ₂₁ are shown respectively.

The operation of the embodiment of the one-stone current composite resonance type converter circuit of the present invention will be briefly described. If for example, a pulse signal from the control circuit 22 to the switching element Q ₂₁ is applied, the switching element Q ₂₁ performs OFF / ON operation, intermittently current flows through the primary coil L ₂₂ of the transformer T _21. AC voltage generated in the secondary winding L ₂₃ of the transformer T ₂₁ By this operation, the diode D _21, and outputs a voltage V _o is rectified and smoothed by the capacitor C _22. The polarity of the winding coil of the transformer T ₂₁ at this time is configured so that an AC voltage is generated in the secondary coil L ₂₃ when the switching element Q ₂₁ is turned on. Further, the error amplifier circuit 21 detects the voltage of the output voltage V _o , and by applying the detection signal to the control circuit 22, the cycle applied to the switching element Q ₂₁ by the control circuit 22 is, for example, the pulse width of the pulse signal. and controls the output voltage V _o is set to be constant.

Next, the current flowing through the circuit of the present invention will be described with reference to the equivalent circuit diagram of FIG. 2 and the waveform diagram of the current flowing through each part shown in FIG. First, the current flowing through each part during the period when the switching element Q ₂₁ is on (the period of 0 ≦ t <t ₂ in FIG. 3) will be described. In this circuit,
The current I ₂₁ flowing through the inductor L ₂₁ is the power supply voltage V _E ,
Inductance L _21a of L _21, the elapsed time when the t, in the period of the switching element Q ₂₁ is turned on, (Equation 1)
Required by

[Equation 1] This current I ₂₁ becomes a sawtooth current that increases linearly until the switching element Q ₂₁ turns off with time t as shown in FIG.

Next, the current I ₂₄ flowing through the diode D ₂₁ on the secondary side becomes a current as shown in FIG. This current I ₂₄ is generated by the diode D ₂₁ becoming conductive due to the generation of a voltage in the secondary coil L ₂₃ based on the primary side resonance current of the transformer T ₂₁ . Then, when the output voltage V _o reaches a predetermined voltage, the diode D ₂₁ becomes non-conductive and the current I ₂₄ becomes zero.

Next, the current I ₂₂ flowing through the capacitor C ₂₁ and the primary coil L _22, becomes the resonance current of the capacitor C ₂₁ and the primary coil L _22. However, as described above, the polarity of the winding coil of the insulating transformer T ₂₁ is configured so that the voltage is generated in the secondary coil L ₂₃ when the switching element Q ₂₁ is turned on, so that the diode D ₂₁ Is 0 ≦ t
In the section of <t ₁ (hereinafter referred to as section A1), (d) of FIG.
A current I ₂₄ as shown in the figure flows. In this section A1, 2
The secondary coil L ₂₃ is short-circuited, and the current I ₂₂ flowing through the primary coil is the leakage inductance of the transformer T ₂₁ and the resonance current of the capacitor C ₂₁ which will be described below when the secondary coil L ₂₃ is short-circuited. Become.

The current I ₂₂ flowing through the capacitor C ₂₁ and the primary coil L ₂₂ in the section A1 will be described with reference to the equivalent circuit of the insulating transformer T ₂₁ shown in FIG. First, the circuit part of the capacitor C ₂₁ and the transformer T ₂₁ in FIG. 1 is shown in FIG. (However, the capacitance of the capacitor C ₂₁ C
_21a , the inductance of the primary coil of the transformer T ₂₁ is L
_22a, inductance L _23a of the secondary coil, the mutual inductance and M) transformer T ₂₁ of this circuit can be represented as an equivalent circuit shown in FIG. 2 (b). Furthermore, since the secondary side circuit is conducting and is in a short-circuited state, as shown in FIG.
_22a inductance -M) and (L _23a -M) are connected in series, can be replaced by inductance (L _23a -M circuit connected in parallel with the inductance) of (M).

Therefore, when the combined inductance of the transformer T _{21 in} the section A1 is L _24a ,

[Equation 2] It can be expressed as. In addition, the mutual inductance M of the transformer T ₂₁ is

(Equation 3) , The combined inductance L _24a is

[Equation 4] Can be shown as Accordingly, the current I ₂₂ flowing through the capacitor C ₂₁ and the primary coil L _22, in the section A1, the
As a resonance current of the series resonance circuit of the combined inductance L _24a and the capacitor C _21a as shown in FIG. 2D, a current as shown in FIG. 3B flows.

The time t ₁ is determined by the resonance frequency at this time,

(Equation 5) Indicated by. By substituting (Equation 4) into this equation,

(Equation 6) Can also be shown. That is, time t ₁ is coil L ₂₂
Inductance L _22a and capacitance C ₂₁ of the capacitor C _21
21a and the time determined by the coupling coefficient K of the transformer T ₂₁ .

[0026] Then, in the switching element Q ₂₁ is turned on, the section (hereinafter, referred to as sections _{A2) t 1 ≦ t <t} 2 shown in (d) of FIG. 3, the output voltage V _o becomes the predetermined voltage Since the current I ₂₄ flowing through the diode D ₂₁ becomes zero, the secondary coil L ₂₃ is opened. Therefore, in this section A2, the current I ₂₂ flowing through the capacitor C ₂₁
Is no longer affected by the leakage inductance of the secondary coil L ₂₃ , and becomes a resonance current between the capacitor C ₂₁ and the primary coil L ₂₂ . If the resonance frequency of the current in this section A2 is f ₂ , then f ₂ is

(Equation 7) Therefore, in this section A2, the current I ₂₂ shown in FIG. 3B flows until the switching element Q ₂₁ is turned off.

The current I flowing through the switching element Q _21
23 is a combined current of the currents I ₂₁ and I ₂₂ as shown in (c) of FIG. 3 while the switching element Q ₂₁ is on.

Next, the current flowing through each part in the section where the switching element Q ₂₁ is off (hereinafter referred to as section A3) will be described. In the switching element Q ₂₁ is turned off in the section A3, the current I ₂₃ of the switching element Q ₂₁ does not flow. Further, since no voltage is generated in the secondary coil L ₂₃ , the diode D ₂₁ is non-conductive and the current I ₂₄ does not flow. However, the inductor L ₂₁ has the above-mentioned switching element Q _21.
When is on, a voltage is generated by the electromagnetic energy accumulated and a current I ₂₁ flows. This current I ₂₁ is due to the fact that the switching element Q ₂₁ is off, so that the inductor L ₂₁ ,
The resonance current is generated by the secondary coil L ₂₂ and the capacitor C ₂₁ . If the resonance frequency of this current is f ₃ ,

(Equation 8) Therefore, in this section A3, the current I shown in FIG.
₂₁ flows, and a current I ₂₂ as shown in FIG. 3B flows in the primary coil L ₂₂ .

Therefore, in the embodiment of the present invention, the relationship between the voltage and the current applied to the switching element Q ₂₁ has a waveform as shown in FIG. 4, and the current flowing through the switching element Q ₂₁ by utilizing the leakage inductance of the transformer T _21. Is a superposition of a plurality of resonance waveforms. Therefore, it is possible to reduce the switching loss in the switching element Q _21.

[0030] Incidentally, the leakage inductance of the isolation transformer T ₂₁ is the insulating transformer T ₂₁ which bond is dense can also be formed by connecting a coil in series. Further, the inductor L ₂₁ can also be formed by a transformer coupling of a plurality of windings.

[0031]

As described above, since the single-stone current composite resonance type converter circuit of the present invention utilizes the leakage inductance of the transformer, an operating state with little influence of leakage magnetic flux can be easily realized with one switching element. This is very useful when it comes to downsizing and weight reduction. Further, since switching can be performed at almost zero crossings when the switching element is on, and switching is performed with a small current when it is off, there is an advantage that loss of the switching element and switching noise can be suppressed.

[Brief description of drawings]

FIG. 1 is a circuit diagram of an embodiment of the present invention.

FIG. 2 is a diagram showing an equivalent circuit of a capacitor and a transformer according to an embodiment of the present invention.

FIG. 3 is a diagram showing a waveform of a current flowing through each part in the embodiment of the present invention.

FIG. 4 is a diagram showing a relationship between voltage and current applied to a switching element according to an example of the present invention.

FIG. 5 is a circuit diagram of a conventional flyback converter circuit.

FIG. 6 is a circuit diagram of a conventional quasi-resonant flyback converter circuit.

FIG. 7 is a circuit diagram of a conventional partial voltage resonance type forward type converter circuit.

FIG. 8 is a circuit diagram of a conventional half bridge type current resonance type converter circuit.

FIG. 9 is a diagram showing a relationship between voltage and current applied to a switching element of a circuit of a conventional example.

[Explanation of symbols]

1, 21 Error amplification circuit 2, 22 Control circuit 3 ON delay circuit 4 Sub control circuit C ₍₁ , ₂₂₎ Smoothing capacitor C ₍₂ , ₃ , ₄ , ₄ , ₂₁₎ Capacitor C _21a Capacity D ₍₁ , ₂ , ₃ , ₃ , ₄₎ , ₅ , ₂₁₎ Diode E DC voltage source L ₍₁ , ₂ , ₃ , ₄ , ₅ , ₅ , ₆ , ₇ , ₈ , ₈ , ₁₀ , ₁₁ , ₁₂ , ₂₂ , ₂₃₎ Winding coil for transformer L ₍₉ , ₂₁₎ Inductor L _(22a , _22b) Inductance M Mutual inductance of transformer Q ₍₁ , ₂ , ₃ , ₂₁₎ Switching element S switch T ₍₁ , ₂ , ₃ , ₄ , ₄ , ₂₁₎ Isolation transformer

Claims (7)

[Claims] 1. A series resonance circuit comprising an insulating transformer and a capacitor, an inductor connected in parallel to the series resonance circuit, and switching means for connecting and disconnecting the series resonance circuit and the DC power supply supplied to the inductor. Prepare,
A one-stone current composite resonance type converter circuit characterized by rectifying an alternating voltage induced in the insulation transformer to output a predetermined voltage. 2. The switching device according to claim 1, wherein the switching means is controlled by a switching element, an error amplifier circuit for detecting a change in output voltage, and a control circuit.
Stone current compound resonance type converter circuit. 3. The single-pole current composite resonance type converter circuit according to claim 2, wherein the switching element is a bipolar transistor, a field effect transistor, or an insulated gate bipolar transistor. 4. The single-pole current composite resonance type converter circuit according to claim 1, wherein the series resonance circuit has coils connected in series. 5. The one-pole current composite resonance type converter circuit according to claim 1, wherein the isolation transformer has a coupling coefficient set so as to obtain a predetermined leakage inductance. 6. The one-stone current composite resonance type converter according to claim 1, wherein the inductor is a choke coil having a single winding or a transformer having a plurality of windings forming a transformer coupling. circuit. 7. The single-stone current composite resonance type converter circuit according to claim 2, wherein the switching element has a pulse width or a switching frequency controlled based on a change in output voltage.