JPH08149898A - Control method for induction motor - Google Patents

Abstract

PURPOSE: To make it possible to control position, speed, and torque with accuracy, by superimposing an AC current into an output voltage command value at an inverter, and estimating the amount of magnetic flux from an AC voltage and leakage inductance in windings of a motor. CONSTITUTION: A given amount of magnetic flux (ϕ) is generated in a motor 2 according to a voltage and a current of the motor 2. A core part, where the magnetic flux (ϕ) passes, is saturated with magnetism. Considering a tooth for storing a primary winding similarly, part of the tooth in the same direction of magnetic flux is saturated with magnetism. At the same time, leakage inductance changes under an influence of magnetic saturation at the tooth part. An AC voltage other than a fundamental wave factor is superimposed in a motor voltage. Then, the inductance of the winding is measured from a relation of a current passed through and the AC voltage, while the magnetic flux (ϕ) is calculated from a change in inductance. The output voltage and current of an inverter 1 are controlled according to the estimated magnetic flux (ϕ), and non-interference control between the torque and magnetic flux (ϕ) is carried out in the motor 2.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a device for controlling an induction motor by means of a power converter such as an inverter, and more particularly to a method for controlling an induction motor capable of highly accurate speed and torque control from an extremely low speed range.

[0002]

2. Description of the Related Art At present, a slip frequency control type vector control method, which is widely used for driving steel rolling mills, FA servo drives, etc., uses an inverter output according to the sum of a slip frequency command value and an actual rotation speed. Since the frequency is controlled, a speed sensor attached to an electric motor is indispensable, and the application is restricted accordingly. For this reason, some high-accuracy speed control methods that do not use speed sensors have been announced (for example, 1993 S. 9th Annual Meeting of the Institute of Electrical Engineers of Japan, Symposium S. 9 "Current status and problems of applying induction motor speed sensorless vector control"). However, both methods estimate the rotation speed based on the induced electromotive force associated with the rotation of the motor.Therefore, in the range where the rotation speed is close to zero and the electromotive force is small, the speed estimation accuracy is affected by the primary resistance drop. Is deteriorated, resulting in insufficient control accuracy of speed and torque.

Further, in the slip frequency control type vector control method (with speed sensor), when the secondary resistance value of the motor used for calculation of the slip frequency command value does not match the actual value,
Problems such as a change in the magnetic flux of the motor depending on the torque or a delay in torque control occur. This is well known as a problem of vector control caused by the variation of the secondary resistance value.

[0004]

In the speed control without a speed sensor, as a method of eliminating the influence of the primary resistance drop, a method of providing a search coil inside the motor or detecting the third harmonic component of the voltage and current of the motor is used. There is also a method of detecting the slot harmonic voltage of the electric motor. However, both methods are similar to the speed sensorless vector control method described above in that the electromotive force due to the change in the primary interlinkage magnetic flux due to the rotation of the electric motor is detected. Similarly, high-accuracy control is difficult because the S / N ratio with respect to noise (such as higher harmonic ripple from the inverter) contained in is reduced. Further, both methods have a drawback that the electric motor structure is specialized.

On the other hand, with respect to the problem of the secondary resistance variation in the vector control with speed sensor, a method of detecting the induced electromotive force of the motor and correcting the secondary resistance value for calculation based on the variation, and a method of modifying the internal resistance of the motor There is a method in which a thermometer is installed in the system, the secondary resistance value is estimated from the detected temperature value, and this is used as the secondary resistance value for calculation, but in the former case, in the range where the induced electromotive force near zero speed is small, Similar to the above, it is difficult to make an accurate correction due to the influence of the primary resistance drop, and in the latter, the electric motor structure is complicated.

An object of the present invention is to solve the above-mentioned problems and to provide a control method for an induction motor which enables highly accurate position, speed and torque control including near zero speed.

[0007]

In order to achieve the above object, an AC voltage is superposed on an output voltage command value of an inverter,
The motor current component flowing in response to this is detected, and the leakage inductance of the motor winding is measured from the superimposed AC voltage and the detected current. The magnitude of the magnetic flux is estimated from the inductance value based on the phenomenon that the inductance value changes depending on the positional relationship of the motor magnetic flux with respect to the windings and the magnitude of the magnetic flux, and the amplitude, phase, and phase of the inverter output voltage are estimated according to the estimated magnetic flux. At least one of the frequencies is controlled to control the excitation component and the torque component (corresponding to the secondary current) of the electric motor current.

[0008]

Function: A magnetic flux is generated inside the motor according to the voltage / current of the motor. Therefore, magnetic saturation occurs in the iron core portion through which the magnetic flux passes (high saturation). The same applies to the teeth portion in which the primary winding is housed, and the portion located in the magnetic flux direction has a high degree of saturation. The leakage inductance of the primary winding changes under the influence of the magnetic saturation of the teeth. Therefore, as described above, an AC voltage different from the fundamental wave component is superimposed on the motor voltage, the inductance of the winding is measured from the relationship between the current flowing thereby and the AC voltage, and the change in this inductance causes the magnetic flux to change. To estimate. The inverter output voltage / current is controlled according to the estimated magnetic flux, and non-interference control (vector control) of the torque and magnetic flux of the electric motor is performed.

In this case, the above-mentioned problem is solved because the vector control is reliably performed even in the low speed rotation range.

[0010]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS One embodiment of the present invention applied to a vector control system without a speed sensor will be described with reference to FIG. In the figure, 1 is an inverter that outputs a voltage proportional to the voltage command v ₁ *, 2 is an induction motor, and 3 is a voltage command v _1d based on the current commands i _1d *, i _1q * and the output frequency command ω ₁ *. *, V _1q * output voltage command calculator, 4 is v
A coordinate converter that calculates a three-phase voltage command v ₁ * from _1d *, v _1q *, 5 is a PWM signal generator that converts v ₁ * into a pulse width modulation signal, and PWM controls the inverter output voltage, and 6 is an electric motor A current detector for detecting a current, 7 is a current component detector for converting an electric motor current into a rotating magnetic field coordinate and detecting an exciting current i _1d and a torque current i _1q , and 8 is a torque current command i _1q * and i _1q.
A current regulator that outputs a frequency command ω ₁ * according to the difference between
9 is an output corresponding to the difference between the exciting current commands i _1d * and i _1d v _1d *
Is a current controller for adding ω ₁ * to output a phase reference signal θ *, 11 is a speed command circuit that outputs a speed command ω _r *, and 12 is a slip frequency ω _s Slip frequency calculator, 13 is ω _r * and velocity estimated value ω _r
A speed adjuster that outputs i _1q * according to the difference of ^ and performs speed control, 14 adds sine wave signals v _1d ″, v _1q ″ to v _1d *, v _1q *, and corresponding i _1d based on the components, the flux calculator for estimating a motor magnetic flux [Phi, 15 is the set value of the magnetic flux [Phi _2d
It is a compensation element used when correcting v _1q * or ω ₁ * from * and Φ.

Next, the operation of this control system will be described. For details of the operation of parts 1 to 13 except for part numbers 14 to 18, Okuyama, Fujimoto, et al., "Velocity / voltage sensorless vector control method for induction motors", The Institute of Electrical Engineers of Japan, D, 107, 2 , Pp. 191-198 (1987), so an outline will be given here.

The system is roughly divided into three parts. The first part is an output voltage controller, which is composed of a voltage command calculator 3, a coordinate converter 4, and a pulse width modulator 5. FIG. 2 shows a vector diagram of the motor voltage v ₁ and the current i ₁ . Here, the dq axes are Cartesian coordinates that rotate at the synchronous speed ω ₁ . v ₁ is the induced electromotive force e ₁ ′ and the leakage impedance drop (r ₁ i ₁ , ω ₁ (l ₁ + l ₂ ′) as shown in the figure.
It is represented by the sum of i ₁ ). Therefore, when controlling v ₁ , the command values v _1d * and v _1q * are calculated according to the equation 1.

[0013]

[Equation 1]

Further, in the coordinate converter 4, v _1d *,
A three-phase voltage command value v ₁ * is calculated from v _1q *. Since the respective phase signals differ only by 120 ° in phase from each other, if only the u-phase voltage command v _u * is shown, then the equation 2 is obtained.

[0015]

[Equation 2]

Further, in the pulse width modulator 5, v ₁ *
Is converted into a pulse width modulation signal, and the output voltage of the inverter 1 is controlled accordingly. In this way, the instantaneous value of the fundamental wave component of the inverter output voltage is controlled in proportion to v ₁ *, and the motor voltage v ₁ is controlled according to v _1d *, v _1q * and θ *. At this time, if the leakage impedance drop estimated value in Equation 1 matches the actual value, the actual value e ₁ ′ (vector) of the induced electromotive force follows the reference value given in Equation 1. Under this condition, the orientation of e ₁ ′ coincides with the q axis, but at this time, the phase reference θ * coincides with the rotation angle θ of the real magnetic flux vector (orthogonal to e ₁ ′) from the stator u phase axis, θ * is equivalent to the rotation angle θ of the magnetic flux.

The second part is a current controller, which is composed of a current detector 6, a current component detector 7, and two current regulators 8 and 9.

As described above, under the condition that the direction of e ₁ ′ coincides with the q-axis, the current component detector 7 calculates i _1d and i _{1q as} the exciting current i ₀ ′ and the torque current i, respectively. It matches ₂ '.

[0019]

(Equation 3)

[0020] Thus, the current regulator 9, modifying the v _1d * depending on the control deviation of _{i 1d, i 1d (i 0} ') is i _1d
Controlled to match *. Also, the motor magnetic flux amount Φ
_2d is controlled by i _1d *. On the other hand, the current regulator 8 controls ω ₁ * and the electromotive force reference value e ₁ ′ * (= ω ₁ * (M / L ₂ ) Φ _2d *) according to the control deviation of i _{1q. 1q} (i ₂ ′) is controlled so as to match i _1q *.
At this time, the motor generated torque τ _e is expressed by _Equation 4, and i _1q
Proportional to *

[0021]

[Equation 4]

The third part is a speed controller, which is composed of a speed command circuit 11, a slip frequency calculator 12, and a speed controller 13. The calculator 12 calculates the estimated value ω _s ^ of the slip frequency according to the equation 5.

[0023]

(Equation 5)

Next, ω _s ^ is subtracted from ω ₁ * to obtain an estimated speed value ω _r ^, and i _1q * is calculated in the speed controller 13 based on the difference between the speed command values ω _r * and ω _r ^. Is calculated.
Then, as described above, i _1q and torque τ _e are controlled according to i _1q *, and as a result, speed control is performed so that ω _r ^ matches ω _r *.

The above is the basic operation of the vector control without speed sensor. By the way, in the operating frequency range of 1 Hz or higher, speed control can be performed with sufficient accuracy in accordance with the above-described operation. However, in the low frequency range of 1 Hz or less, the rotational speed and torque control accuracy deteriorates.

It is considered that this problem is mainly caused by the fluctuation of the motor primary resistance r ₁ . That is, when r ₁ fluctuates due to the temperature change of the electric motor, the estimated value (r ₁ * i ₁ *) of the primary resistance drop used in Equation ₁ and the actual primary resistance drop (r ₁ i ₁ ) do not match. At this time, the actual value of the motor magnetic flux Φ becomes a value different from the set value Φ _2d *, and the direction of Φ does not match the d axis. Under the condition that the frequency is low and the electromotive force e ₁ ′ is minute, the proportion of the primary resistance drop in the voltage v ₁ increases, and this tendency becomes remarkable. In this way, in low frequency operation, the magnetic flux of the motor has a value different from the set value due to primary resistance fluctuation (estimation error of primary resistance drop). At this time, the vector control becomes incomplete, and the torque τ _e is no longer proportional to i _1q *. Also, the torque (i _1q )
Since the magnetic flux Φ fluctuates in relation to ω _s ^, an estimation error also occurs in ω _s ^ calculated according to Equation 5, and as a result, ω _s ^
_An error also occurs in _r ^. As a result, the speed and torque control accuracy deteriorates.

The above problems are common to vector control without a speed sensor, and various methods have been proposed as countermeasures as described above, but there is currently no drastic method. The present invention solves these problems by adding a magnetic flux calculator 14 and the like.

FIG. 3 is a block diagram of the calculation contents of the magnetic flux calculator 14. In the figure, 31 is a two-phase sine wave signal (sinω
t, cosωt) output signal generator, 32 is a signal (sin
ωt) and input (1 / √2) s according to modes 1, 2 and 3.
In ωt, (1 / √2) sin ωt, a switch circuit that outputs each signal of sin ωt, 33 corresponds to modes 1, 2, 3
(1 / √2) sinωt, − (1 / √2) sinωt, switch circuit that outputs each signal of 0, 34 and 35 are current i _1d and signal
Multipliers for multiplying sin ωt and cos ωt respectively, 36, 3
7 is an integrator for integrating the outputs of the multipliers 34, 35, 38 is an inductance calculator for measuring the inductance values L _σ1 , L _σ2 , L _σ3 in each mode based on the output values of the integrators 36, 37, and 39 is An arithmetic unit for calculating the magnetic flux of the motor based on each L _σ , 40 is √ according to modes 1, 2 and 3.
It is a switch circuit that outputs 2i _1d , √2i _1d , i _1d .

Next, the contents of calculation will be described. First, the basic principle of estimating the magnetic flux magnitude Φ will be described. FIG. 4 shows a model of the induction motor. Assuming that the magnetic flux Φ exists inside the motor in the direction shown in the figure, the iron core portion located in the Φ direction causes magnetic saturation (high saturation). The same applies to the teeth portion in which the primary winding is housed, and the portion located in the Φ direction has a high degree of saturation. The leakage inductance of the primary winding changes due to the influence of magnetic saturation of the teeth. For example, as shown in the drawing, the leakage inductance of the winding A located in the Φ direction is smaller than that of the winding B orthogonal to the Φ direction.

FIG. 5 shows the actual measurement result of the leakage inductance, and shows the change of the leakage inductance value of each winding with respect to the exciting current (magnetic flux amount). As shown in the figure, in the vicinity of the rated excitation current (3A), the positional relationship between the magnetic flux and the winding causes
It was confirmed by experiments that the inductance value changes significantly.

From this, it is possible to detect the inductance values at several positions by changing the position (phase) to be measured, and conversely estimate the magnitude of the magnetic flux or the position (direction) from the difference between these values. By controlling the output voltage / current of the inverter according to the estimated magnetic flux, vector control can be performed with high accuracy without being affected by the change in the primary resistance described above. The above is the basic principle of magnetic flux estimation.

Next, the principle of measuring the inductance L _σ used when estimating the magnetic flux will be described. Now, a sine wave voltage v (= sinωt) having a frequency different from the fundamental wave is applied to the electric motor, and the alternating current i flowing by this is observed. Under the condition that the angular frequency ω of v is sufficiently higher than the reciprocal of the secondary time constant T ₂ of the electric motor, the transfer function of the alternating current / applied voltage of the winding can be approximated by a first-order lag system, and therefore i is expressed by Equation 6.

[0033]

(Equation 6)

On the other hand, the detected i is Fourier-transformed with v as a reference to obtain a component synchronized with v and a 90 ° phase difference component,
Given that each is equal to the first term and the second term on the right side of Equation 6, L _σ is obtained from Equation 7, that is, v and i
L _σ can be measured based on

[0035]

(Equation 7)

Next, the basic principle of magnetic flux estimation and the operation of the calculator 14 will be described. Now, as shown in FIG.
The angle between the direction of the magnetic flux Φ and the direction of the magnetomotive force of the winding C to which the AC voltage v is applied is defined as φ. L _σ is π / 2,
Since it becomes the minimum at 3π / 2 and the maximum at 0 and π, L _σ changes as a function of 2φ. So L
_σ can be set as in Expression 8.

[0037]

Equation 8] _{L σ = L σm (1 +} acos2φ) ... ( 8) Here, L _.sigma.m: Mean value of L _σ, a: L variation range of _{sigma here, φ = φ 1 + π /} 4, and phi = An alternating voltage is sequentially applied to each of the windings of φ ₁ −π / 4, and L _σ is set as described above.
To measure. _Let each L _σ be L _σ1 and L _σ2 ,

[0038]

[Formula 9] L _σ1 = L _σm (1-asin2φ ₁ ) (Formula 9)

[0039]

[Formula 10] L _σ2 = L _σm (1 + asin2φ ₁ ) (Formula 10) From Formula 9 and Formula 10,

[0040]

[Equation 11]

That is, φ = φ ₁ + π / 4, and φ =
The average value L _σm of L _σ can be obtained by the two-point measurement of φ ₁ −π / 4. Since L _σm changes according to the magnitude of the magnetic flux, the magnitude of the magnetic flux can be conversely estimated from this value.

Further, by applying an AC voltage to the winding of φ = φ ₁ and measuring L _σ3 ,

[0043]

[Equation 12] L _σ3 = L _σm (1 + acos2φ ₁ ) (Equation 12) From Equations 9 and 12,

[0044]

(Equation 13)

[0045]

[Equation 14]

That is, φ = φ ₁ + π / 4, φ ₁ −π /
By measuring three points of 4, φ ₁ , φ ₁ and the change width a can be obtained. The position of the magnetic flux can be estimated by obtaining φ ₁ . Further, since the variation width a is considered to depend on the magnitude of the magnetic flux, Φ can be estimated.

Next, two methods for obtaining the magnitude Φ of the magnetic flux from the value of L _σm or a will be described. One is Φ
And L _{σ m} (or a) are made into a function, and necessary coefficients are obtained in advance before operation. For example, the relationship between Φ and L _{σ m} is approximated as in Expression 15, the necessary coefficient K _m is measured in advance, and during normal operation, an approximate expression such as Expression 15 is applied to the obtained L _{σ m.} Φ is calculated and obtained.

[0048]

(Equation 15)

Another method is to make detailed measurements by changing the conditions before operation, and make a table of the relationship between L _σm and Φ under each operating condition, and use this table to calculate Φ from the table during operation.
The value of is calculated. Although the number of data will increase, high precision can be expected. In order not to increase the number of data,
Φ may be obtained by combining a function such as Expression 15 and a table. When a has a strong dependence on Φ, Φ may be obtained by relating a function or table to a. Alternatively, both L _σm and a can be used.

The calculator 14 operates based on the above-mentioned estimation principle. The operation will be described below with reference to the vector diagrams of FIGS. 3 and 7. In order to perform vector control, it is ideal that the magnitude and direction of the magnetic flux Φ match the set value Φ _2d *, but here the angle difference φ ₁ is assumed assuming that they do not match. Hereinafter, each case of modes 1, 2, and 3 will be described in order.

[Mode 1] Sine wave signal ((1 / √2) sin
ωt) is v _1d ″ through the switch circuit 32, and v
₇ is added to _1d * and is added to _v1q * as _v1q ″ via the switch circuit 33. This state is shown in FIG.
Direction of mode 1 (4 with respect to d-axis)
This is equivalent to applying an alternating voltage v to a winding having a magnetomotive force direction of 5 °. At this time, an alternating magnetomotive force is generated in this direction, and an alternating current i flows. The phase of i does not change even when observed from the d axis, so i can be detected from i _1d . Therefore, by using the multipliers 34 and 35, the integrators 36 and 37, and the inductance calculator 38, i _1d and the signal (sin ω
Based on t, cos ωt), the calculation of Expression 7 is performed to obtain L _σ1 . L _σ1 is stored in the calculator 38.

[Mode 2] signal ((1 / √2) sinωt)
Via the switch circuit 32 as v _1d ″, and the polarity inversion signal (− (1 / √
2) sin ωt) is added to each voltage command value as v _1q ″. In this state, v is applied in the direction of mode 2 in FIG. 7, and the alternating current i flows in the same direction. similar to allows detection of i _1d, by carrying out the same operation as mode 1, .L L _{.sigma. @ 2} is required _{.sigma. @ 2} are similarly stored.

[Mode 3] The signal (sin ωt) is set to v _1d ″, and v _1q ″ = 0, and added to each voltage command value. In this state, v in the direction of Mode 3 (d-axis) in FIG.
Is applied, the alternating current i in the same direction (d-axis)
Flows. i can be detected as i _1d as it is, and L _σ3 is obtained and stored in the same manner as described above.

L _σ1 , measured as described above,
Based on L _σ2 and L _σ3 , the arithmetic unit 39 calculates
1 or the equations 13 and 14 are performed to estimate Φ or the magnetic flux position angle φ ₁ with respect to the d-axis.

In order to detect the position of the magnetic flux, the method of measuring L _σ by the three modes has been shown. However, when only the value L _σm related to the magnitude of the magnetic flux Φ is obtained, as shown in equation (11), It can be calculated by measuring only in two modes, mode 1 and mode 2.

Next, the operation of the entire vector control system without a speed sensor to which the present invention is applied will be described with reference to FIG. The basic operation of the system is as described above. In order to improve the accuracy of vector control, the magnetic flux calculator 14 is added in the present invention. The AC signals v _1d ″, v _1q ″ from the calculator 14 are constantly v _1d *, v _1d *, during operation.
It is added to v _1q *, so that the current i _1d becomes
Current components related to v _1d ″, v _1q ″ are included. Although i _1d originally includes a direct current component related to the fundamental wave component of the electric motor current, this effect is eliminated in the calculation of L _σ in equation 7. Therefore, Φ is measured regardless of the operating state of the electric motor.

The magnitude Φ of the magnetic flux is output from the calculator 14. Φ is input to the slip frequency calculator 12 and the compensation element 15. The slip frequency calculator 12 calculates the slip frequency using Φ. Therefore, the estimation accuracy of ω _s ^ is improved, and at the same time, the estimation accuracy of ω _r ^ is also improved as compared with the conventional calculation using the set value Φ _2d *. Further, v _1q * or ω ₁ * is corrected via the compensating element 15 in order to compensate the torque fluctuation of the electric motor due to the fluctuation of Φ. The output 16 of the compensation element 15 is ΔΦ (= Φ _2d * −Φ)
_Is directly added to v _1q * to correct the fluctuation of the primary resistance drop. Also, the output 17 is ω instead of v _1q *.
₁ * is corrected. In this case, v _1q * is controlled by the current regulator 8 so that i _1q = i _1q *, and Φ =
Ω ₁ * is corrected to be Φ _2d *, and the torque fluctuation is compensated. In any of the compensation methods, v _1q * is compensated on the basis of the variation ΔΦ of the magnetic flux, and control is performed so that the vector relationship of FIG. 2 is maintained.

As another compensation method, a method of correcting the exciting current command i _1d * by using ΔΦ and controlling the magnetic flux so as to be Φ _2d * can be considered. In this case, the vector relationship between the voltage and the current cannot be maintained in the transient state as shown in FIG. 2, but the magnetic flux can be constantly controlled to be constant.

In this way, even if the influence of the primary resistance drop becomes large and Φ and Φ _2d * do not match, v _1q *, ω ₁ * or i _1d * is passed through an appropriate compensation element.
Can be made to coincide with each other by correcting, and high-precision vector control can be maintained.

As a result, the problem of deterioration of accuracy of speed and torque in the low speed rotation range as described above is solved.

FIG. 8 shows another embodiment of the present invention. This is an example in which the present invention is applied to a vector control system without a speed sensor, which is provided with an AC current control system that causes the instantaneous output current i ₁ of the inverter to follow the sine wave current command i ₁ *.

In the figure, 81 is an inverter which outputs a current i ₁ proportional to the sine wave current command i ₁ *, 82 is an induction motor, 83 is current commands i _1d *, i _1q *, and a phase reference θ.
A coordinate converter that calculates a three-phase current command i ₁ * based on *
84 is an AC current controller that inputs the difference between i ₁ * and i ₁ and outputs a voltage command v ₁ *, and 85 is a PWM signal that converts v ₁ * into a pulse width modulation signal and PWM-controls the inverter output voltage Generator, 86 is a current detector for detecting electric motor current, 87
Is a current component detector for detecting a torque current i _1q by converting i ₁ into a rotating magnetic field coordinate with a magnetic flux obtained by integrating the detected value v ₁ of the motor voltage or its command value v ₁ * as a phase reference. Is a current controller that outputs an estimated speed value ω _r ^ according to the difference between the torque current command value i _1q * and the detected value i _1q , 89
Is a slip frequency calculator for calculating the slip frequency estimated value ω _s ^ based on i _1q *, and 90 is the frequency reference ω ₁ * obtained by adding ω _r ^ and ω _s ^, and the phase reference signal θ * output phase computing units, 91 speed command circuit for outputting a speed command omega _r *, 92 outputs i _1q * depending on the difference between ^ omega _r * and omega _r, speed regulator controlling the speed , 93 is a voltage component detector that detects the d-axis voltage v _1d by converting the detected value v ₁ of the motor voltage or its command value v ₁ * into a rotating magnetic field coordinate based on θ *, and 94 is i _1d *, i _1q * is a sine wave signal i _1d ″, i _1q ″
, And a magnetic flux calculator for estimating the magnitude Φ of the magnetic flux based on the component of v _1d generated by this.
_This is a compensation element used when modifying _1q *.

Next, the basic operation of this control system will be described. This system can also be roughly divided into three parts. The first part is an output current controller, which is composed of a coordinate converter 83, a current controller 84, a pulse width modulator 85, and a current detector 86. The coordinate converter 83 calculates a three-phase current command value i ₁ * from the d-q axis current command values i _1d *, i _1q *. Each phase command value is 120
Since only the phases differ by degrees, the u-phase current command value i
_If only _u * is shown, then the number is 16.

[0064]

[Equation 16]

In the current regulator 84, v ₁ * is calculated according to the difference between i ₁ * and i ₁ . Furthermore, the PWM modulator 85
At, v ₁ * is converted into a pulse width modulation signal, and the output voltage v ₁ of the inverter ₁ is controlled accordingly,
i ₁ is controlled in proportion to i ₁ *. As a result, i ₁ becomes i
Controlled according to _1d *, _i1q *, and θ *.

The second part is a speed estimator, which is a current component detector 87, a current controller 88, and a slip frequency calculator 8
9 and a phase calculator 90. Current component detector 8
In FIG. 7, first, the electric motor magnetic flux Φ used for current detection is calculated according to equation (17).

[0067]

[Equation 17]

Φ is divided by the amplitude value | Φ |, and a magnetic flux phase signal (sin θ, cos θ) having a constant amplitude and a sine wave is calculated. Based on the signal, according to equation 3 (replace θ * with θ), i
_1q is calculated. In the current regulator 88, ω _r ^ is calculated based on the difference between i _1q * and i _1q . That is, as described above, under the condition that i ₁ is controlled according to i _1d *, i _1q *,
The difference between i _1q and i _1q * is caused by the fact that θ *, which will be described later, does not match the magnetic flux phase θ, and ω ₁ * is controlled by the current regulator 88 to correct this. As a result, θ * = θ holds and vector control is performed correctly. In the vector control, the motor magnetic flux Φ _2d is maintained at the predetermined value Φ _2d * without being affected by the torque change.
In 9, the ω _s ^ is correctly estimated according to the _equation 5 (replace i _1q with i _1q *). In addition, ω _r ^ (= ω ₁ * -ω _s ^) is also correctly estimated. The θ * is the phase calculator 90
Is obtained by integrating ω ₁ * at, and used as the phase reference of the coordinate converter 83 and the voltage component detector 93.

The third part is a speed controller, which is composed of a speed command circuit 91 and a speed adjuster 92. I _1q * is calculated in the speed adjuster 92 based on the difference between the speed command values ω _r * and ω _r ^, and the torque τ _e is controlled according to equation ₁ according to the equation 4, so that ω _r ^ becomes ω _r *. Speed control is performed so that they match.

The above is the basic operation of the vector control without the speed sensor. However, even in this case, the control accuracy is deteriorated due to the fluctuation of the primary resistance, especially in low frequency operation. This is because when the magnetic flux Φ is calculated in the current component detector 87, the motor constant (r ₁ ,
This is because Φ has an error when L _σ ) does not match the actual value. A detection error also occurs in i _1q due to the magnetic flux estimation error, and θ * = θ does not hold, resulting in incomplete vector control. In this way, the speed and torque control accuracy deteriorates as in the previous embodiment.

Therefore, in this embodiment, the voltage component detector 9
3 and a magnetic flux calculator 94 are added to solve this problem. FIG. 9 shows the outline of the calculation contents of the magnetic flux calculator 94. In the figure, 101 is a two-phase sine wave signal (sinωt, cosωt)
The signal generator 102 outputs the signal sin ωt, and (1 / √2) sin ωt,
(1 / √2) A switch circuit that outputs signals sin ωt and sin ωt, and 103 corresponds to modes (1, 2 and 3) (1 /
√2) sinωt, − (1 / √2) sinωt, a switch circuit that outputs each signal of 0, 104 is a voltage v _1d and a signal (cosω
t) a multiplier, 105 an integrator that integrates the output of the multiplier 104, and 106 an inductance value L _σ1 , L _σ2 in each mode based on the output value of the integrator 105.
An inductance calculator for measuring L _σ3 , and 107 for each L
It is a calculator that calculates the electric motor magnetic flux Φ based on _σ .

Next, the principle and contents of the magnetic flux calculation will be described. This basic concept is the same as that described above.
Further, since the content of operation of the arithmetic unit 107 are also the same as the operation unit 39 of FIG. 3, described contents until obtaining the _{L σ1, L σ2, L σ3} .

Now, a sine wave current i (= sinωt) having a frequency different from the fundamental wave is passed through the electric motor, and the AC voltage v generated from this is observed. Under the condition that the angular frequency ω of i is sufficiently higher than the reciprocal of the quadratic time constant T ₂ , the transfer function of v / i can be approximated by a first-order lead system, so v is expressed by the following equation 18.

[0074]

V = (R _σ + jωL _σ ) i (Equation 18) On the other hand, the detected v is Fourier-transformed with i as a reference, and a component in-phase with i and a 90 ° phase difference component are obtained, each of which is a number. 18 is equal to the first term and the second term on the right side of L
_{When σ} is obtained, it is Formula 19.

[0075]

[Formula 19]

Here, | i | is the magnitude of the current, which is a preset amount.

As described above, the difference from the previous embodiment is that in the previous embodiment, the AC voltage v is applied to the winding and the current i flowing therethrough is used to measure L _σ. An alternating current i is passed through the winding, and the voltage v generated from this is measured. Subsequent operations, that is, φ =
i is sequentially passed through three windings of φ ₁ + π / 4, φ ₁ −π / 4, φ ₁ , and L _σ1 , L _σ2 , and L _σ3 are measured from each v,
The content of calculating Φ is the same as in the previous embodiment.
That is, in FIG. 9, the switch circuits 102 and 103
, I _1d ″ determined for each mode as described above,
i _1q ″ is added to i _1d *, i _1q * to obtain the motor current i ₁
The above-mentioned sine wave current i is superimposed on. As a result, in each mode, the alternating current i flows in the directions of modes 1, 2, and 3 shown in FIG. 7, and accordingly, the alternating voltage v is generated in each direction. Since v is observed as an in-phase quantity on the d-axis,
v can be detected from the d-axis voltage v _1d . v _1d is calculated and detected according to the equation (20).

[0078]

(Equation 20)

In the multiplier 104, the integrator 105 and the inductance calculator 106, v _1d and the signal (cosω
Based on t), the operation of _Equation 18 is performed, and L _σ1 , L
It is possible to obtain _σ2 and L _σ3 . Then, each L _σ
Based on the above, Φ is calculated in the calculator 107. Since the contents are the same as those of the arithmetic unit 38 of FIG. 3, the description thereof will be omitted.

Next, the operation of the entire system shown in FIG. 8 will be described. The basic operation of the system is as described above. In order to solve the accuracy deterioration due to the fluctuation of the motor constant,
A magnetic flux calculator 94 is added, but its output calculation value Φ
Is measured regardless of the operating state of the electric motor. Φ is
It is used in the calculation of ω _s ^ in the slip frequency calculator 89, and as a result, the estimation accuracy of ω _s ^ and ω _r ^ is improved.

The fluctuation of the primary resistance in this embodiment causes a calculation error in the magnetic flux calculation in the current component detector 87. As a result, i _1q includes a detection error, which needs to be compensated. As a compensation method, as shown in FIG.
Φ (= Φ _2d * −Φ) is added to i _1q * via the compensating element 95, and i _1q is modified, and the flux calculator (flux phase) of the current component detector 87 is modified according to ΔΦ. A possible method is to add it. In this case, it is necessary to compensate for the polarity of i _1q *.

Also, a compensating method for correcting the exciting current command i _1d * according to ΔΦ can be considered. In this case, Φ _2d
* And Φ can be matched, but they cannot be matched during a transition. However, when the electric motor is operated in the vicinity of no load, it is more stable to correct the exciting current command i _1d *.

From the above, Φ _2d * ≠ Φ due to primary resistance fluctuation
Even in such a case, Φ _2d * = Φ can be obtained by adding appropriate compensation using Φ, and vector control can be performed with high accuracy. That is, also in the present embodiment, similar to the previous embodiment, high-accuracy control of speed and torque can be performed in the entire range including zero speed.

Next, another method of compensating for the secondary resistance fluctuation in vector control with a speed sensor, which is another object of the present invention, is a method in which compensation is possible especially including zero speed and a temperature sensor attached to a motor is unnecessary. Describe.

FIG. 10 shows a voltage control type vector control system which is another embodiment of the present invention. In the figure,
The reference numerals 1, 2, 4 to 7 and 11 are the same as those shown in FIG. 121 is a magnetic flux calculator for calculating the magnitude Φ of magnetic flux, 122 and 123 are current regulators for outputting values corresponding to the respective current deviations i _1d * -i _1d and i _1q * -i _1q , and 124 is a motor. Is a speed detector for detecting the rotation speed ω _r , 125 is a speed controller that outputs i _1q * according to the difference between ω _r * and ω _r , and controls the speed, and 126 is a current command i _1d *, i _1q * And a non-interference controller 127 that calculates induced electromotive force components of the voltage commands v _1d * and v _1q * based on the angular frequency ω ₁ and outputs a slip frequency command ω _s * by multiplying i _1q * by a coefficient. , A slip frequency calculator, 128 is a secondary resistance setter for setting the secondary resistance value r ₂ * used as the coefficient, and 129 is a signal for adding ΔΦ to the output of 128 to correct r ₂ *. Is a compensation element, and 130 is Δ
It is a compensation element used when ω ₁ * is modified using Φ.

Next, the operation of this control system will be described. The system is roughly divided into four parts. The first part is an output voltage controller, which is composed of a non-interference controller 126, a coordinate converter 4 and a pulse width modulator 5.

In the non-interference controller 126, the induced electromotive force components e _1d *, e _1q * of the motor voltage are calculated according to the equation (21).

[0088]

[Equation 21]

The current regulators 122 and 123 are added to e _1d * and e _1q *.
_Are added, and the voltage commands v _1d *, v _1q * are calculated. With the coordinate converter 4 and the pulse width modulator 5, FIG.
The inverter output voltage v ₁ is controlled in the same manner as in the above embodiment.

The second part is a current controller, which includes the current detector 6, the current component detector 7 and the two current controllers 12.
It is composed of 2,123. The current component detector 7 detects the current components i _1d and i _{1q in the same} manner as in the embodiment of FIG. Since the current regulators 122 and 123 correct v _1d * and v _1q * according to the respective control deviations of i _1d and i _1q , i
_1d is controlled to match i _1d *, and i _1q is controlled to match i _1q *. At this time, the motor-generated torque τ _e is expressed by the above-mentioned _equation 4, and is controlled in proportion to i _1q *.

The third part is a speed controller, which is composed of the speed command circuit 11, the speed detector 124 and the speed adjuster 125. I _1q * is calculated according to the velocity deviation ω _r * −ω _r , and the torque τ _e is controlled in proportion to i _1q * as described above, so that ω _r is equal to ω _r * Control is performed.

The fourth part is a slip frequency control section,
It is composed of a slip frequency calculator 127 and a secondary resistance setter 128. The slip frequency command ω _s * is calculated by the calculator 126 according to the equation (22).

[0093]

[Equation 22]

[0094] Next omega _r and omega _s * a seek to omega ₁ addition, theta is obtained by further integrating the omega ₁ at the phase calculator 10.

The above is the basic operation of the vector control with speed sensor. By the way, since the slip frequency ω _s * is calculated by using the secondary resistance setting value r ₂ * as shown in Formula 22, when the secondary resistance changes due to the temperature change of the motor secondary winding, _{Since s} * contains an error, the magnetic flux and torque cannot be controlled according to the command values (i _1d *, i _1q *), and high-precision control cannot be performed.

Therefore, in this embodiment, a magnetic flux calculator 121 is added to solve this problem. That is, the magnetic flux Φ is calculated using the magnetic flux calculator as in the embodiment of FIG. 1, and ΔΦ (=
Φ _2d * −Φ) is used to correct the fluctuation of the secondary resistance. Specifically, the set value of r ₂ * is corrected via the compensating element 129 to correct the slip frequency, or the compensating element 130 is used.
Correct ω ₁ directly via. As a result, ω _s * is set correctly, and the magnetic flux and torque can be controlled with high accuracy according to the command value.

FIG. 11 is a block diagram of a vector control system showing another embodiment of the present invention. This is an example in which the present invention is applied to perform secondary resistance variation compensation in a vector control system including an AC current control system for feedback controlling the output instantaneous value i ₁ of the inverter. In the figure, 81
Since ~ 86, 90, 91, 93, and 94 are the same as those of the same numbers shown in FIG. Also, 12
4, 125, 127 to 130 are the same as those having the same numbers shown in FIG.

Next, the operation of this control system will be described. The current control is the same as that of FIG. 1 and the speed control is the same as that of FIG. Moreover, since the magnetic flux calculation is the same as that of FIG. 8, an outline will be described.

The output current i ₁ of the inverter 81 is controlled according to the AC current command value i ₁ * calculated in the coordinate converter 83 based on the torque current command value i _1q * and the exciting current command value i _1d *. . Further, the inverter output frequency is controlled by the added value ω ₁ * of the rotation speed ω _r and the slip frequency command value ω _s *. The slip frequency is controlled according to the above-mentioned equation 22, but if the secondary resistance value r ₂ fluctuates, the magnetic flux phase will not coincide with θ and high-precision control cannot be performed. Therefore, a magnetic flux position calculator 94 is added to obtain the magnetic flux Φ in the same manner as described above.
ΔΦ modifies r ₂ * via compensation element 129 or ΔΦ modifies ω ₁ * via compensation element 130. As a result, high-precision control can be performed as in the above-described embodiment.

As described above, even if the magnetic flux position and the coordinate axis are deviated due to the secondary resistance fluctuation, the magnetic flux position can be matched with the coordinate axis, and it is possible to perform compensation in the vicinity of zero velocity, which was difficult in the past. You can

In the above embodiment, the magnetic flux calculator is always operated during operation to compensate for the secondary resistance fluctuation. However, in the high speed rotation range, a conventional compensation method such as induced electromotive force or motor magnetic flux is used. Even if you use the method to correct the slip frequency based on the fluctuation of, the sufficient compensation accuracy can be obtained, so even if you suspend the operation of the magnetic flux calculator in the high-speed rotation range and perform the conventional compensation, However, the same effect as the above embodiment can be obtained.

Further, in the above-described embodiments, the magnetic flux calculation is performed on the dq axes which are the rotating coordinates, but the same effect can be obtained even when it is performed on the α-β axes which are the fixed coordinates. .

As described above, according to the present invention, a vector control system without a speed sensor capable of performing highly accurate speed control in the entire speed range including the vicinity of zero speed, and a motor constant (secondary resistance) in the entire speed range including the vicinity of zero speed. ) It is possible to provide a vector control system (with a speed sensor) that can perform fluctuation compensation.

Next, examples of applying the magnetic flux position calculation method of the present invention and the motor control method using the same to various systems will be described.

The magnetic flux calculation of the induction machine according to the present invention can be basically realized by the apparatus shown in FIGS. In FIG. 12, 150 is a command value generator that outputs a final command voltage applied to an induction machine such as a vector control device (without a speed sensor or with a speed sensor) or a V / F control device, and 151 is a voltage command. An exciter such as an inverter or a linear amplifier that outputs a corresponding voltage, 152 is a current sensor for detecting the primary current of an induction machine, 153 is an induction motor, and 154 generates an identification voltage v _h * and the induction voltage v _h * A magnetic flux calculator for calculating the magnetic flux Φ from the machine primary current, 1541 is a signal extractor such as a filter for extracting only the current component i _h having the same frequency as the identification voltage from the induction machine primary current, 154
Reference _numeral 2 denotes a magnetic flux calculator that calculates the magnetic flux Φ of the induction machine from v _h * and i _h .

Similarly, in FIG. 13, reference numeral 160 denotes a command value generator 161 for outputting a final command current applied to an induction machine such as a vector controller (without speed sensor or with speed sensor) and slip frequency controller. Is an exciter such as an inverter or a linear amplifier that outputs a current according to a current command, 162 is a voltage sensor for detecting the primary voltage of an induction machine, 153 is an induction motor, and 163 generates an identification current i _h *. A magnetic flux calculator that calculates the magnetic flux Φ from _h * and the induction machine primary current, 1631 is a signal extractor such as a filter that extracts only the voltage component v _h that is the same as the frequency of the identified current from the induction machine primary voltage, and 1632 is i _h It is a magnetic flux calculator that calculates the magnetic flux Φ of the induction machine from * and v _h .

The magnetic flux Φ inside the motor can be obtained by the calculation method described so far with reference to FIG. 12 or FIG. It is also possible to combine FIG. 12 and FIG. 13 and provide both a current sensor and a voltage sensor, and to calculate the magnetic flux position from those values, which is rather superior in accuracy. Reference numerals 1541 and 1542, 1631 and 1632 in FIGS. 12 and 13 are devices for realizing the magnetic flux calculation algorithm represented by the equations 7 to 14.

FIG. 14 shows an embodiment of an AC servo system incorporating a magnetic flux calculator. By using this system, the control response at the time of starting and low speed is improved. In FIG. 14, 170 is a position command generator that generates a position command p *, and 171 is an actual position p and a position command p *.
The position controller 172 calculates a speed command value more, and 172 uses the speed command output from the position controller 171 and the magnetic flux information Φ (or a value related to Φ) from the magnetic flux calculator to generate a command voltage to the inverter. A speed controller for calculation, 173 is an inverter for applying a voltage to the induction machine according to the input command voltage v ₁ *, 174 is a mechanical system system (control target) driven by the induction machine, and 175 is a position for detecting the position of the control purpose. It is a sensor. Also, 152, 153 and 154 are shown in FIG.
Follow

The servo system of the induction motor is used for driving the spindle. Since this type of motor rotates at a high speed, mounting the speed sensor of the electric motor becomes a problem from the mechanical strength viewpoint. Therefore, it is desired to eliminate the need for a speed sensor. In the spindle drive, sufficient torque is required from the low speed range, but if the servo system shown in FIG. 14 is used, it is possible to comply with this without a speed sensor, and high-responsive position control can be realized.

FIG. 15 shows an embodiment of a rolling mill drive system incorporating a magnetic flux calculator. By using this system, high rolling accuracy can be realized without a speed sensor. In FIG. 15, reference numeral 180 is a speed command generator for generating a speed command ω _r * of the electric motor, and 181 is the speed of the electric motor ω.
_r , speed command ω _r *, speed controller (vector controller, etc.) that calculates the command voltage to the inverter using the magnetic flux information Φ (or a value related to Φ) from the magnetic flux calculator, 182
Is a speed calculator that estimates the motor speed from the motor primary current, and 183 is a rolling mill system driven by an induction machine. Also, 152, 153, 154 and 173 follow FIG.

At present, a large number of vector control induction machines with speed sensors are used in steel process lines and the like. However, the location where the motor is installed is often in a bad environment where dust, vibration, heat (temperature rise), etc. accompany it, and therefore the speed sensor mounted on the motor is placed under severe conditions.
Many failures occur in this. Further, maintenance may be difficult depending on the installation location of the electric motor, and it takes a lot of time to restore the electric motor. For this reason, the application of a motor control system without a speed sensor is drawing attention. Conventionally, there is a problem in control performance that the speed control accuracy is low in the low speed range, and therefore torque nonuniformity occurs between electric motors used in the same line, and smooth operation cannot be performed.

On the other hand, according to the rolling mill drive system of the present invention, since the high-precision control can be performed in the entire speed range including the zero speed, the above-mentioned problem is solved and the maintenance-free operation without the speed sensor is realized. .

FIG. 16 shows an embodiment of a torque control system incorporating a magnetic flux calculator, and examples of the system include electric trains and electric vehicles. By using this system, high efficiency and downsizing of the electric motor can be realized.

In FIG. 16, reference numeral 190 is a torque current command generator for generating a torque current command i _1q * for the electric motor, and 19
1 is the actual motor torque current i _1q , torque current command i _1q
*, A torque current controller that calculates the command voltage to the inverter using the magnetic flux information Φ (or a value related to Φ) from the magnetic flux calculator, 192 is a torque current calculator that estimates the motor torque from the motor primary current, Reference numeral 193 is a drive system of a train or an electric vehicle driven by an induction machine.
Also, 152, 153, 154 and 173 follow FIG.

Electric trains and electric vehicles are required to have sufficient torque even at low speed operation such as start-up acceleration. However, especially when starting uphill, it is necessary to overcome gravity. Therefore, sufficient torque is required even at zero speed. . Therefore, conventionally, a system with a speed sensor has been used which detects the motor speed and controls the inverter output frequency using this.

However, because of the mounting location of the electric motor, there are many vibrations and there is a problem with the reliability of the speed sensor. Therefore, it is desired to eliminate the need for the speed sensor. According to the present invention, since a sufficient torque can be obtained without a speed sensor in the entire speed range including zero speed, a highly reliable system can be realized. Further, even in the low speed region, since the proportional relationship between the torque current and the actual torque is maintained, it is not necessary to supply an unnecessary current, so that the system efficiency can be improved and the electric motor can be downsized.

Further, even in the case of a system with a speed sensor,
The reliability of the system is improved by always providing the speed sensorless vector control according to the present invention as a backup when the speed sensor fails.

FIG. 17 shows an elevator system incorporating a magnetic flux calculator. By using this system,
The system configuration can be simplified and downsized. In the figure, 200 is an elevator position command p.
Position command generator that generates *, 201 is the actual position p,
A position controller that calculates the speed command ω _r * from the position command p *,
202 is a speed controller that calculates a torque current command i _1q * from the motor speed ω _r and speed command ω _r *, and 203 is a torque that calculates a command voltage to the inverter from the torque current component i _1q and the torque current command i _1q * A controller, 204 is a speed calculator that estimates the motor speed from the motor torque, 205 is a torque current calculator that estimates the torque current component i _1q from the motor primary current, 206 is an elevator system driven by an induction machine, and 207 is an elevator. It is a position sensor that detects the position of the car. Magnetic flux information Φ from the magnetic flux calculator 154
(Or a value related to Φ) is provided to the controllers 201 to 203 as needed. Also, 152, 15
3,154,173 follow "FIG. 14".

In the elevator system, a large starting torque that overcomes the gravity or the static frictional force is required in the stopped state of rotation. Therefore, when the conventional technique is applied, a torque shortage occurs in the low speed range, and a large current flows through the electric motor and the inverter to compensate for this, and there is a problem that these become large.

According to the present invention, since high-precision control can be performed in the entire speed range including zero speed without a speed sensor, the system configuration is simplified, and at the same time, the above-mentioned torque shortage and the problems associated therewith are solved. .

The embodiments have been described above with respect to several systems incorporating the magnetic flux calculator of FIG. Similar effects can be obtained by using the magnetic flux calculator of FIG. 13 instead of FIG. 12 and using a current control type inverter as the inverter.

Although the embodiment has been described with respect to the system without the speed sensor, the control accuracy and the response speed can be expected to be improved in the system with the speed sensor by using the magnetic flux information of the motor.

For example, although vector control with a speed sensor is used for the main machine drive and process line in the rolling mill drive, it has been described in the section of "Prior Art" due to secondary resistance fluctuation due to temperature change of the electric motor. Such a problem occurs. Motor voltage (magnetic flux) according to torque
Fluctuation increases the allowable maximum value of the inverter output voltage and causes the size of the inverter to increase. Further, the torque control delay makes high response control difficult. According to the vector control system with a speed sensor according to the present invention, the above-mentioned problems are solved, and an economically excellent system with high response and high performance is realized.

In addition, a train with a speed sensor is often used in the current trains, electric cars, and AC servo systems. If the control system according to the present invention is applied to these, the same effects as those in the above-described rolling mill drive can be obtained. Further, according to the present invention, it is possible to simultaneously estimate the resistance together with the magnetic flux of the electric motor, and it is also possible to diagnose an abnormal state such as a failure of the electric motor by monitoring the change of the magnetic flux and the change of the resistance.

[0125]

According to the present invention, since the magnetic flux in the electric motor can be estimated without being affected by the primary resistance of the induction motor, vector control based on the electric motor magnetic flux becomes possible, and near zero speed. It is possible to realize highly accurate speed and torque control including.

[Brief description of drawings]

FIG. 1 is a configuration diagram of a vector control device without a speed sensor according to an embodiment of the present invention.

FIG. 2 is a vector diagram of motor voltage and current.

FIG. 3 is a configuration diagram of a magnetic flux position calculator in FIG.

FIG. 4 is a model of an induction motor.

FIG. 5 is a measurement result of leakage inductance according to the present invention.

FIG. 6 is a diagram showing a positional relationship between a magnetic flux of a motor and windings according to the present invention.

FIG. 7 is a vector diagram of a leakage inductance measurement mode according to the present invention.

FIG. 8 is a configuration diagram of a vector control device without a speed sensor according to another embodiment of the present invention.

9 is a configuration diagram of a magnetic flux position calculator in FIG.

FIG. 10 is a configuration diagram of a vector control device according to another embodiment of the present invention.

FIG. 11 is a configuration diagram of a vector control device according to another embodiment of the present invention.

FIG. 12 is a configuration diagram of a magnetic flux position calculator according to the present invention.

FIG. 13 is another configuration diagram of the magnetic flux position calculator of the present invention.

FIG. 14 is a configuration diagram of an AC servo system of the present invention.

FIG. 15 is a configuration diagram of a rolling mill drive system of the present invention.

FIG. 16 is a configuration diagram of a train and electric vehicle system of the present invention.

FIG. 17 is a configuration diagram of an elevator system according to the present invention.

[Explanation of symbols]

1 ... Inverter, 2 ... Induction motor, 3 ... Voltage command calculator, 4 ... Coordinate converter, 5 ... PWM signal generator, 6 ... Current detector, 7 ... Current component detector, 8 ... Current regulator, 9 ... Current regulator, 10 ... Phase calculator, 11 ... Speed command circuit, 1
2 ... Slip frequency calculator, 13 ... Speed controller, 14 ... Magnetic flux calculator.

Claims (17)

[Claims] 1. In an induction motor driven by an AC power supply, an AC component having a frequency different from its output frequency is superimposed on the output of the AC power supply, and as a result, the amount of AC flowing through the motor according to the AC component. And the AC component, the physical quantity related to the saturation state of the iron core of the electric motor is detected, and the magnitude of the electric motor magnetic flux is determined based on the detected physical quantity. 2. The magnetic flux calculation for an induction motor according to claim 1, wherein the leakage inductance of each winding of the electric motor is measured as the physical quantity, and the magnitude of the magnetic flux of the electric motor is obtained from the difference in the leakage inductance between the windings. Law. 3. The power supply according to claim 2, wherein the AC power supply is a power converter capable of arbitrarily controlling an output voltage or an output current and a frequency, and an AC signal is applied to a command value for commanding an output voltage or a current of the converter. A magnetic flux calculation method for an induction motor, wherein the AC component is superimposed on the output by adding the magnetic flux. 4. The voltage command value or current command value and frequency on the α, β axes which are orthogonal on the stator coordinate axes or the d and q axes which are orthogonal on the rotating magnetic field coordinate axes, according to claim 3. A power converter for outputting AC power based on a command value, wherein the AC signal is added to each voltage command value or each current command value on the α, β axes or d, q axes, and the electric motor at that time. The leakage inductance of the electric motor is measured based on the AC or β-axis or d, q-axis converted value of the current or voltage of
A magnetic flux calculation method for an induction motor, characterized in that the magnetic flux of the motor is obtained based on this. 5. The alternating current signal according to claim 4, wherein each of the voltage command values or the current command values on the d, q axes or the α, β axes corresponds to at least two modes, and the AC signal is a predetermined value. An alternating magnetomotive force is generated in two different directions inside the electric motor for each mode, and 2 corresponding to each mode based on each axial component of the electric current or voltage of the electric motor and the AC signal at this time. A magnetic flux calculation method for an induction motor, which is characterized by measuring two inductances and finding the magnitude of the fundamental wave magnetic flux of the motor based on this. 6. The alternating current signal according to claim 4, wherein a predetermined value is added to each of the voltage command values or the current command values on the d, q axes or the α, β axes in accordance with at least three modes. Then, an alternating magnetomotive force is generated in three different directions inside the electric motor for each mode. Based on each axial component of the electric current or voltage of the electric motor and the AC signal at this time, three magnetomotive forces are generated corresponding to each mode. A method for calculating the magnetic flux of an induction motor, characterized by measuring the inductance and finding the magnitude and position (angle) of the fundamental wave magnetic flux of the motor based on this. 7. The method for calculating magnetic flux of an induction motor according to claim 5, wherein the two different directions have an electrical angle difference of 90 °. 8. The magnetic flux calculation method for an induction motor according to claim 6, wherein the three different directions each have an electrical angle difference of 45 °. 9. The method according to claim 4, wherein before the actual operation, an alternating current is applied to the electric motor from the converter to measure the leakage inductance of the electric motor in advance, and the characteristic amount (either the average value and / or the fluctuation range) is determined. It is calculated and stored, and during actual operation, the leakage inductance is measured using two of the modes, and the magnetic flux of the motor is obtained based on the inductance value and the characteristic amount. Magnetic flux calculation method. 10. An induction motor control method for driving and controlling an induction motor by a power converter that arbitrarily controls an output voltage or an output current and a frequency based on these command signals. A method for controlling an induction motor, characterized in that at least one of the amplitude, frequency and phase of the output voltage or output current is changed according to a signal of the electric motor magnetic flux obtained by a magnetic flux calculation method. 11. The command signal of the frequency according to claim 10, which is obtained from an addition value of a slip frequency command value calculated by multiplying a torque current of the electric motor by a coefficient and a rotational speed detection value, A method of controlling an induction motor, wherein the coefficient is changed according to the magnetic flux of the electric motor obtained by the magnetic flux calculation method according to claim 1. 12. The at least one of the amplitude, frequency, and phase of the output voltage or output current of the power converter according to the magnetic flux signal only when the output frequency of the converter is a predetermined value or less. A method for controlling an induction motor, characterized by changing one of the two. 13. An induction motor controller for driving and controlling an induction motor by a power converter for arbitrarily controlling an output voltage or an output current and a frequency based on these command signals. A method of diagnosing an induction motor for diagnosing an operating state of the induction motor based on a signal of the electric motor magnetic flux obtained by a magnetic flux calculation method. 14. A power converter for outputting AC power, an induction motor driven and controlled by the converter, a mechanical system connected to the induction motor, and means for detecting at least the position of the mechanical system. A means for instructing the position, a position control means for generating an output signal so that the detected value of the position becomes the instructed value, and an output signal of the position control means is used as a speed command to obtain a command for the converter. An AC servo system for controlling the position of the mechanical system, comprising speed control means, wherein position control is performed based on a signal of the electric motor magnetic flux obtained by the magnetic flux calculation method according to claim 1. The characteristic AC servo system. 15. A power converter that outputs AC power, an induction motor that is drive-controlled by the converter, a rolling mill that drives a rolling roll using the induction motor as a power source, and at least the rotation speed of the induction motor. Alternatively, means for estimating or detecting the rotational speed of the roll, means for instructing the rotational speed, and speed control means for generating an output signal so that the estimated value or the detected value of the rotational speed matches the command value. ，
A rolling mill drive system for controlling the speed of the rolling rolls, comprising a means for controlling the converter based on the output signal, wherein the electric motor obtained by the magnetic flux calculation method according to any one of claims 1 to 9. A rolling mill drive system characterized by performing speed control of rolling rolls based on a magnetic flux signal. 16. A power converter that outputs AC power, an induction motor driven and controlled by the converter, a train or an electric vehicle that uses the induction motor as a power source, and at least a driving torque of the train or the electric vehicle. Means for detecting or estimating the torque, means for instructing the torque, torque control means for generating an output signal so that the detected value or estimated value of the torque matches the command value, and the torque control means based on the output signal. In a train or electric vehicle system including means for controlling a converter, speed control of the train or electric vehicle is performed based on a signal of the electric motor magnetic flux obtained by the magnetic flux calculation method according to claim 1. A train or electric vehicle system characterized by that. 17. A power converter for outputting AC power, an induction motor driven and controlled by the converter, an elevator using the induction motor as a power source, and means for detecting the position of a car of the elevator. , Means for instructing the position of the car, position control means for generating a first output signal so that the detected value of the car position matches the command value, and estimating or detecting the rotational speed of the induction motor And a speed for generating the second output signal such that the estimated value or the detected value of the rotation speed of the induction motor matches the command value with the first output signal as the rotation speed command of the induction motor. Control means, means for estimating or detecting the torque of the induction motor, and the second output signal are used as the torque command for the induction motor, and the command value of the induction motor is set to this command value. In an elevator drive system comprising: torque control means for generating a third output signal so that the estimated value or the detected value of the torque match, and means for controlling the converter based on the third output signal, An elevator drive system characterized in that the elevator car position or speed is controlled based on the signal of the electric motor magnetic flux obtained by the magnetic flux calculation method according to any one of claims 1 to 9.