JPH0776871B2 - Tone generation system - Google Patents

Description

The present invention relates to musical instruments, and more particularly to digitally controlled electronic musical instruments and methods of producing musical tones.

A digitally controlled method of producing musical tones is performed by producing a series of digital numbers and converting them into electrical analog signals. The analog signal is amplified,
Generate a musical sound through an ordinary speaker. Musical instruments that perform digital control include keyboards and other input devices,
It consists of a digital electronic circuit that responds to the keyboard. The electronic circuit digitally processes the signal in response to the keyboard and digitally produces audible vibrations at the speaker. The vibrations generated by these digital equations are different from the vibrations generated by analog oscillators, and also the vibrations generated mechanically by ordinary orchestra and other types of musical instruments.

All musical tones, be they electronic or mechanical, can be described by the Fourier spectrum. In the Fourier spectrum, musical tones are described by expressing their component frequencies as sinusoids. Therefore, the whole musical sound is the sum of component frequencies, that is, the sum of sinusoids.

Under Fourier analysis, sound quality is classified into overtones and dissonances. Overtones are periodic and can be represented as the sum of sinusoids with frequencies that are integer multiples of the fundamental frequency. The fundamental frequency is a pitch of sound quality. Harmonious instruments for orchestra include string instruments, brass instruments, and woodwind instruments. Discord is not periodic, but is often expressed as the sum of sinusoids. However, frequencies containing discords usually have fairly complex relationships. Discord music instrument,
It usually has no pits associated with it. Percussion instruments, such as bass drums, snare drums, cymbals, etc., are examples of orchestral instruments that produce dissonance.

Electronically controlled instruments have relied on the selection of the Fourier spectrum as the basis for generating musical tones. One of the known types of digital musical instruments uses a harmonic total tone generation method. In this method, sound quality is obtained by adding (or subtracting) a large number of amplitude scale sinusoids of different frequencies. However, the harmonic synthesis method requires a complicated addition (or subtraction) process to form each sample. This process requires an expensive and inflexible digital circuit. Therefore, the digital design required to implement the harmonic synthesis method requires complicated computer calculations and is often unsatisfactory.

Another known type of instrument uses the overtone method.
In this method, a desired frequency component is selected by passing a complicated electric waveform such as a square wave or a sawtooth waveform sequence with one or more filters. Then, the passed frequency components are combined to form an electronic signal for driving the speaker. This overkill is commonly used to synthesize human words,
Often has an analog electronic organ. The pass-through method is rather inflexible because it relies on the value of a constant sample stored by each sample. Obtaining a natural sound requires a large number of multiplication steps, which is expensive.

Both the harmonic synthesis method and the over-law method depend on a linear combination of sinusoids, and therefore have characteristics as a linear method for generating a musical sound. By multiplying the amplitude of the input function (sinusoid in the harmonic synthesis method, pulse train in the past method) by a factor of 2, we obtain an output waveform with the same sound quality and an amplitude with a factor of 2 The sex is clear.

Chowning, U.S. Pat. No. 4,018,121, entitled "METHOD OF SYNTHESIZING A MUSICAL SOUND", describes a non-linear method of generating musical tones. This non-linear method uses a closed form representation (based on frequency modulation) to represent the sum of innumerable sinusoids. This non-linear frequency modulation method produces a large number of sinusoids, these sinusoids having a frequency that is the sum of the carrier frequency and an integer multiple of the modulation frequency. The multiple amplitude of the modulation frequency is the sum of the Bessel functions. Chowning's nonlinear frequency modulation method is superior to the above-mentioned linear harmonic synthesis and over-methods, and has found commercial applications in the field of music synthesis.

`` MUSICAL INSTRUMENT AND METHOD FOR GENERATING MUS
Moorer U.S. Pat. No. 4,215,61 entitled "ICAL SOUND"
No. 7 describes an improved non-linear method of producing musical tones. In this method, the amplitude of the frequency component is not restricted by the Bessel function, and a finite spectrum, that is, a spectrum composed of a total of a finite number of sinusoids can be used.

Many linear and non-linear methods similar to the ones described above have been successfully used in digital tone synthesis, but all of these methods require fast and complex computational power to obtain rich and natural sounds. To do. As a result, the cost of the instrument increases,
It gets complicated. This defeats the purpose of widespread use of digital synthesis.

Therefore, there is a need for an improved musical instrument that uses digital synthesis that is slower and requires less computational power than the prior art, and is capable of producing rich natural sounds. is there. There is also a need for an improved digital tone synthesizer which can be constructed using conventional computer processors and conventional semiconductor chip technology.

SUMMARY OF THE INVENTION In light of the above background, it is an object of the present invention to provide a tone generating system that produces a rich and natural sound by utilizing a simple ordinary digital circuit that does not require complicated calculations.

The tone generation system according to the present invention includes a readable / writable storage unit for storing a waveform sample signal, and a first address designating unit which is incremented by a clock signal and designates a first address based on the increment value. And a pitch instructing means for supplying numerical data corresponding to the pitch of a musical tone to be generated, and an address value instructed by the first address according to the numerical data supplied by the pitch instructing means. A second address designating means for designating a second address according to an address value and a waveform sample signal read from the storage means are input, processing for changing or not changing the waveform sample signal is performed, and the output signal is stored in the storage. Waveform processing means to be input to the means, and one of the first address and the second address as a read address The waveform sample signal is read from the storage means, and the output signal of the waveform processing means is written in the storage means by using the other of the first address and the second address as a write address. And a tone signal having a periodicity corresponding to this difference is generated.

Corresponding to the embodiments of the present invention described below, the readable / writable storage means is the wavetable unit 13, and the first address indication means is the latches 66, 60, 61, 67, 49 in the digitizer unit 35. A portion including the arithmetic unit 62,
The pitch indicating means is a portion including the input unit 2 and the shift register 56 in the digit unit 35, and the second address indicating means is the latch 6 in the digit unit 35.
0, 61, 67, 49, a portion including the arithmetic unit 62, the waveform processing means is a portion including the latches 45, 65, 60, 61, 67 and the arithmetic unit 62 in the modifier unit 14 or the digital unit 35, the control unit Generally correspond to the portion of the digital unit 35 including the control logic circuit 71 and the like.

A difference corresponding to the numerical data is generated between the write address and the read address, and the storage means functions as a delay circuit that delays the difference. In this way, a tone signal having a periodicity corresponding to the variable delay time is generated, and the generated tone has a tone pitch designated by the numerical data. Since the frequency of the clock signal for address advancement may be constant regardless of the pitch of the musical sound to be generated, it is possible to generate a musical tone having an independent pitch corresponding to each of a plurality of voices. When configuring the device, time division processing can be easily performed. In addition, the configuration for variable control of the musical tone pitch also offsets the Aledos value of the address signal stepped at a constant rate so that a constant offset corresponding to the desired pitch occurs between the write address and the read address. The configuration is simple because it only needs to be controlled.

An instrument according to an embodiment of the present invention described below includes a keyboard and other input devices, a wave table change generator for generating a digital signal by changing a wave table, and an output device for converting a digital signal into a musical sound.

The generator contains a wavetable which is periodically accessed to provide an output signal which determines the musical note. The output signal from the wave table is changed, then returned to the wave table and stored as change data. After a delay, the modified data is accessed from the wavetable and becomes the new output signal. This process is repeated periodically, with each new output signal being modified and returned to the wavetable for storage therein. According to the invention, a new output signal is thus generated by the wavetable modification and is used to generate a rich and natural tone.

According to the invention, at any time t, the modification signal y _{t, which} is stored back in the wave table, is a function of the original contents of the wave table and the modification component m _t . Therefore, the signal y _t is a function of x _t and m _t as shown below.

y _t = f (x _t , m _t ) In the digital sample embodiment, the nth sample of y _t is given by y _n . With the delay operator, N
For wavetables showing cycle delays, y _n is x _t
Is a function of the n-th value of x _n, and the delay value of y _n given as d ^N y _n times the change component, and is given by the following equation.

y _n = f (x _n , d ^N y _n m _n ) According to an embodiment of the type suitable for producing string-plucked notes, it is carried out to produce the following modified value y _n. The change is the first delay output y _nN and the previous delay output y _{n- (N + 1)}
Is the average value of.

In the repellent string instrument embodiment, the nth modified value stored back in the wavetable is:

y _n = x _n + [y _nN + y _{n- (N + 1)} ] / 2, where x _n is the nth from the wavetable when it is equal to 0 when n is greater than (N + 1) , Y _n is the modified output at the nth sample, N is the wavetable delay (approximately the desired pitch period of the note in the sample), and y _nN is the sample delayed by N minutes, y _{n- (N + 1)} is a sample delayed by N + 1 minutes.

In the plucked string embodiment, the modification is performed, for example, as a simple addition and binary shift (divided by two) of the data stored in the wavetable. In one digital storage embodiment, the location of the wavetable data is determined by a storage address pointer. The read pointer points to the location of the delay sample y _nN . The "read pointer + 1" is offset by 1 from the read pointer, and the delay sample
Specifies the location of y _{n- (N + 1)} . The modified value y _n is stored in the wave table at the location designated by the write pointer. The write pointer is offset from the read pointer by the pitch delay N.

In multi-voice embodiments, the pitch delay N is typically different for each voice. The read pointer and the write pointer are determined for each voice.

The wavetable modification method of the present invention achieves the object of providing an improved digital instrument that can be implemented with low computational requirements.

The above and other objects and features of the present invention will be apparent from the following detailed description of the preferred embodiments with reference to the accompanying drawings.

In FIG. 1, a digital synthesizer musical instrument is shown here. This musical instrument includes an input unit 2 for designating a musical sound to be generated, a wave table change generator 3 for generating a signal representing the musical sound to be generated, and an output unit 4 for generating a desired sound. To do.

In FIG. 2, this shows the instrument of FIG. 1 in more detail. The input unit 2 usually contains a keyboard 5 for connecting electrical signals to the interface 6. This keyboard 5 is of conventional design and produces an electrical signal when a key is depressed. The keyboard 5 is a typical device for designating the sound to be generated, but any other type may be used. Furthermore, the input unit 2 usually includes means for specifying the force (amplitude) of the sound and the duration of the sound.

The interface unit 6 encodes keyboard information (pitch, amplitude, duration) and transmits it to the control unit of the wavetable change generator 3. The generator 3 produces a signal on the output bus 8 in response to the signal from the input unit 2, which signal is sent to the output unit 4. The output unit 4 converts this signal into a desired musical tone. Usually, the output unit 4 contains a digital-to-analog converter 9. The analog signal from the converter 9 is a reduction filter 10
And to the speaker 12 through the amplifier 11. The speaker 12 emits a desired musical sound.

In FIG. 2, the wave table change generator 3 includes a wave table 13, a changer unit 14, and a control unit 15.
Include. The wave table 13 acts as a delay device having a delay time p. The modification signal from the modifier unit 14 is stored in the wavetable 13 where it is delayed by a time p before appearing on the bus 16. The output signal on the bus 16 from the wavetable is modified by the modifier unit 14,
It is stored in the wave table 13. For the digital example, the delay time p is represented by N samples of data. Wavetable 13 as one digital device
Is a digital store for storing N samples of data. For time t values smaller than p, the wavetable 13 stores the first value of x _t . In a digital system, t is quantized to N values of n, so that
In the initial state, x _n has N initial values.

Due to the background, if there is no change in the modifier unit 14, the wavetable 13 will produce an output signal y _t which is periodic with a delay time p. When the first contents in the delay line is x _t, the output signal y _t
Is represented as follows.

y _t = y _(tp) = x _t (1) When time t is quantized into individual values and p is equal to N values of n (where N is an integer and x _n is N of x _t ). (Representing each individual value), equation (1) can be written as:

y _n = y _(nN) = x _n (2) Since the change is made by the modifier unit 14, the formula (1),
(2) does not apply to the output signal used in the present invention.

According to the invention, at any time t, the modification signal y _t stored back in the wave table is a function of the minimum content x _t of the wave table and the modification component m _t . Therefore, the signal y _t is x _t , as given by the following equation (3),
It is a function of m _t .

y _t = f (x _t , m _t ) (3) In the digital sample embodiment, the nth sample of y _t is given as y _n , and the delay operator d ^N is used to indicate a wave table showing N cycles of delay. , Y _n is x _n
Is a function of the _n-th value of y and the delay value of y _n given by d ^N y _n times the change factor m _t , and can be expressed by the following equation (4).

y _n = (x _n , d ^N y _n m _n ) (4) With reference to equation (4), the maximum number of samples stored for the value N may be any number. The larger the number of samples, the lower the frequency components that can occur. In one example, 256 8-bit samples are stored. Also, many different forms of change component m _t can be selected. However, considering simplification and economy,
It is desirable that m _t be easy to calculate. One simple modification component will be described in connection with the generation of musical notes of a repelling string instrument.

Flip-Play String Musical Tones FIG. 3 shows the details of the modifier unit 14 for generating the tone of the flip-play stringed instrument. Modifier unit
14 includes a delay unit 26 which delays the signal y _nN on the bus 16 by one cycle of n. The output from this TY26 forms one input y _{n- (N + 1)} on line 28 and sends it to arithmetic unit (AU) 27. The other input to the arithmetic unit 27 is derived directly from the bus 16. The arithmetic unit 27 calculates the value on the bus 16
Form the sum of the previous values above (output from TY26). A shift input 29 to the arithmetic unit 27 shifts this sum by one binary bit and divides this sum by two.

When the modifier unit of FIG. 3 is connected to the wave table modification generator 3 of FIG. 2, a repulsive string instrument sound is generated. The wavetable change generator 3 of FIG. 2 formed using the modifier unit 14 of FIG. 3 resembles a digital filter in some respects. A digital filter in the form of a generator 3 mimics the strings of a mechanically vibrating string instrument. The short input signal to the digital filter represents the "plucking" of the "string" and excites the generator 3 to produce the desired output signal during the next delay time. The output signal from the modifier unit 14 when presented on line 8 in FIG. 2 is given as follows.

y _n = x _n + [y _nN + y _{n- (N + 1)} ] / 2 (5) where x _n is the input signal amplitude at sample n and y _n is the output amplitude at sample n And N is the desired pitch time (rough value) of the sound in the sample.

The basic fluffy string sound is obtained by exciting the generator 3, for example, with a short burst of "white noise". The burst of white noise is obtained by selecting the value of x _n in equation (5) as follows.

_xn = Au _n , n = 0, 1, 2, ... (N + 1) ... (6
₁ ) x _n = 0, n ≧ N (6 ₂ ) where A has the desired amplitude and u _n has the value +1 or −1 as a function of the output of the random number generator. n is 0, 1, ...
The value of x _n when equal to (N + 1) is the modifier unit 14
Is stored in the wave table 13 as an initial value existing prior to any change by.

Finally, the output y _n is utilized by the output device 4 of FIG. 2 to generate the beginning portion of the sound when n equal N.

Analysis of flutter performance string instrument sound The input / output relation of expression (5) can also be expressed by the "delay operator" binary method. Here, the unit sample delay operator d is defined by the following relationship. That is, d ^k x _n = x _nk (7) where x _n is an arbitrary signal and k is an integer. Multiplying one signal by d ^k delays this signal by k samples. As a result, the equation (5) becomes as follows.

y _n = x _n + [d ^N y _n + d ^{N + 1} y _n ] / 2 = x _n + d ^N [(1 + d) / 2] y _n ··· (8) When the output signal y _n is calculated, y _n = x _n / {1 - [(1 + d) / 2] d ^n} ··· (9) linear delay operator expression, it replaces each time signal at the z deformation, replacing d with z ^-1 Then, it is immediately converted into the z-variant equation. The time signal (eg, x _n or y _n ) is shown in lower letters, and the corresponding z variant values are shown in upper-case lower case letters, such as X (z) or Y (z).

The transfer function of the digital filter (linear time constant expression) is the z-transform value of the input. The transfer function of the repellent string simulator of FIG. 2 is obtained by subtracting z ⁻¹ for d in equation (9) and transposing as follows.

H (z) = Y (z) / X (z) = 1 / {1- [1 + z- ¹ ) / 2] z- ^N } = 1 / {1- [Ha (z)] [Hb (z)] } (10) where Ha (z) = (1 + z ⁻¹ ) / 2 Hb (z) = z ^−N Equations (9) and (10) are the wave table generator 3 of FIG.
An example of the above will be described. The feedback loop consists of a length of N delay line Hb (z) which is considered as a wavetable 13 in series with a two-point average Ha (z) and also as a modifier unit 14. Wave table
It is at the input terminal 20 to 13 and the output from the modifier unit 14 is at terminal 21. Terminal 21 is connected to terminal 20 in a feedback relationship forming a closed loop.

The frequency response of the generator of FIG. 3 is z = e ^jωT = cos
It is defined as the transfer function calculated by (ωT) + jsin (ωT), where T is the sampling time (seconds),
Is the reciprocal of the sampling rate f _s , ω = 2πf is the radian frequency, f is the frequency (Hz), Is. The frequency response of the plucking string instrument of FIG. 2 is given as follows.

H (e ^{j ωT} ) = 1 / [1-Ha (e ^iωT ) Hb
(E ^iωT )] (11) Here, H (e ^jωT ) = [1 + e ^−jωT ] / 2 = e ^{−jωT / 2} c
os (ωT / 2) = e ^−πfT cos (πfT) (12) Hb (e ^jωT ) = e ^−jωNT = e ^−j2πfNT (13) Separate amplitude response and phase transition of wavetable component Hb and modifier component Ha You should think about it. The magnitude response is defined as the magnitude of the frequency response and gives the gain as a function of frequency. Phase delay is defined as the negative complex angle of the frequency response divided by the radian frequency, giving the time delay (in seconds) caused by the sinusoid for each frequency.

The amplitude responses Ga and Gb of the components Ha and Hb are given as follows, respectively.

G (f) = ^1Ha (e ^jωT ) = cos (ωT / 2) = cos (πf
T) (14) G (f) = Hb (e ^jωT ) = 1 (15) ^Therefore , the delay line component Ha is lossless, and the two-point average Ha is the first quadrant of the cosine. Shows a gain that decreases with frequency according to. All frequencies are limited to the Nyquist limit. That is, f ≦ f _s / 2. In this range, cos (πfT) = cos (πfT).

It is convenient to have a phase delay defined in units of samples rather than seconds. The phase delay of Ha and Hb of the sample is given as follows.

Pa (f) = − [ / Ha (e ^jωT )] / ωT = 1/2 (16) Pb (f) = − [ / Hb (e ^jωT )] / ωT = N (17) where “ / ” Indicates the complex angle of z.

The two-point average Ha (z) has a phase delay equal to half the sample and the delay line has a phase delay equal to its length N. The entire loop consists of Ha (modifier unit 14) and Ha (wavetable 13) in series, so the loop gain is:

Loop gain = Ga (f) Gb (f) = cos (πfT) (18) Then, the effective loop length is as follows.

Loop gain = Pa (f) = Pb (f) = N + 1/2 (sample) (19) This is for each sinusoidal frequency f (Hz).

In synthesizing a single plucked chord, N samples of white noise at amplitude A are provided by the wavetable 13 and the output tone occurs immediately after the first N samples. Delay line represented by wavetable 13
Hb is essentially filled with random numbers normalized at time 0,
No other input signal need be used. Modifier unit 14
The two-point average Ha from is always changing, and the contents of the loop and the output signal y _n are not periodic. However,
The output signal is chosen to be periodic, and thus the term "periodic" in this application means approximately periodic or quasi-periodic. Each "cycle" of the synthesized string sound corresponds to the contents of the delay line (wave table 13) at a specific time,
Each cycle is equal to the intermediate reduction of the previous cycle. Speaking in detail, run a sample that was offset for one cycle 2
Point averaging gives the output waveform the next period. Since the effective loop length is N + 1/2 samples, this period is NT + T / 2
Is best defined as (seconds), where t is the sampling period. Also, this is equal to 1 / f _s ,
Where f _s is the sampling frequency. Experiments show that NT + T / 2 agrees well with the perceived pitch period.

Since the overtone collapse output signal y _n is quasi-periodic, it does not consist of individual sinusoids. In essence, the output signal has many narrow energy bands that decay to zero at different rates. When these energy bands are concentrated at frequencies that are integer multiples of the minimum frequency, we call them "overtones". If the frequency components are not necessarily evenly spaced, we will use the expression "partial sound" to emphasize the possibility of dissonance.

Here, a partial sound having a frequency f (Hz) circulating in the loop is considered. Each pass through the loop causes the partial to be attenuated equal to the loop amplitude response, resulting in Ga (f) Gb (f) equal cos (πfT). That is, one cycle of attenuation = cos (πfT) Since the lap time in the loop is equal to N + 1/2 samples, the number of laps through the loop after n samples (nT seconds) is n / (N + 1/2). ) = Tf _s / (N + 1/2). Therefore, the "damping factor" a (t) at time t = nT is given by:

For example, the first partial sound amplitude A at time 0 becomes the amplitude Aaf (t) at time t. Here, f is the frequency of the partial sound.

The "time constant" cf of exponential decay is traditionally defined as the time when the amplitude collapses to 1 / e, or about 0.37 of its initial value. The time constant at frequency f is given by the formula (2
0) equal to e ^{−t / (cf)} and is found by computing for cf (where f _s = 1 / T).

cf = -t / [1nat (t)] =-[(N + 1/2) T] / [1n cos (πfT)] (seconds)
(21) In the case of audio, it is usually more advantageous to define the decay time constant as -60 dB of decay, that is, 0.001 times the initial value. In this case, equation (20) is set equal to 0.001 and t is calculated. This value of t is often referred to as t ₆₀ . Conversion from cf to t ₆₀ (f) is performed as follows.

t ₆₀ (f) = 1n (1000) cf (22) That is, about 6.91c.

For example, if the sinusoid at frequency f (Hz) has amplitude A at time 0, amplitude Aaf (t at time t ₆₀ (f)
₆₀ (f)) = A / 1000, ie 60 below the departure point
has dB.

The above analysis describes the attenuation due to propagating components around the loop. However, it does not "fit" into the loop and does not contain components that decay rapidly due to self-interference. This self-interference is similar to a mechanically vibrating physical string. A string of excitation effects can be applied to the string, but after the excitation is stopped, the remaining energy rapidly exhibits quasi-periodicity, which is largely determined by the length of the string.

Similarly, for example, even if the loop of the instrument of FIG. 2 is started with a random number, in a very short time, the main frequency existing in the loop will have an integer of the period in N + 1/2 samples. , These frequencies are all multiples of the minimum frequency called the fundamental frequency, and their period exactly matches the loop length N + 1/2. This minimum frequency f ₁ gives the pitch frequency of the sound and is given as follows.

f ₁ = 1 / [N + 1/2) T] = f _s / (N + 1/2) (23) where f _s is the sampling frequency and T = 1 / f _s is the sampling period.

If we set f to the sequence of harmonics starting at f ₁ , we get:

f _k = ω _k / 2π = k [f _s / (N + 1/2)] ・ ・ ・ (24) Here, k = 1, 2, ... N / 2 is the k-th Give the decay factor at time t for the overtones.

Similarly, the time constant per harmonic is given in seconds as follows.

c _k = -t / [1n _ak (t)] =-{1 / [f ₁ 1n (cos (πf _k T))]} (26) Fig. 4 shows the equipment of the type shown in Figs. Figure 9 shows the spectral evolution during the first 15 periods of a sound with a period of 128 samples. A fast Fourier transform (FFT) of length 128 is performed every other period, that is, n equal to 0, 2, 4, ...
Was calculated. Note that the higher the overtones, the faster the collapse, which is a sympathy to the actual behavior of the strings. The higher the overtone, the more quickly its energy dissipates.

In one alternative embodiment (where modifier unit 14 has Ha which is considered to be significantly larger than the two-point average), k after t seconds.
The attenuation factor of the second overtone is as follows.

a _k (t) = Ga (f _k ) {tf _s / [N + Pa (f _k )]} (27) Here, Ga (f) ≦ 1 for stability. Phase collapse Pa
(F _k ) is the same for all overtone frequencies f _k .

Similarly, when Ha is considered to be a big key, the decay time constant for each harmonic is given in seconds as follows.

c _k =-[(N + Pa (f _k )) / (f _s 1nGa (2πf _k T))]
(28) 16-speech embodiment FIG. 5 shows a 16-speech embodiment of the musical instrument of FIG. 2 using a modifier unit in the form of FIG.

In FIG. 5, the wave table 13 is a random access storage device (RAM), one for each of 16 voices.
Has 6 different storage areas. Each of these storage areas has a wavetable storage location for 256 8-bit bytes. Wave table modifier generator 3 is 8
It includes a bit output register 36 and a digit unit (DIGITAR) 35. This digitizer unit 35 makes the modifications described above in connection with the modifier unit of FIG. 3 for all 16 voices of the instrument of FIG. The digital unit 35 has a 12-bit address bus 38 connected to address the wavetable 13. further,
The digital unit 35 is connected to the 8-bit data bus 37. A data bus 37 connects the wavetable 13 to the output register 36. The input unit 2 is connected to a common data bus 37 by a bus 7. Output register 36
Are connected to the output unit 4 as inputs and to the wave table 13 as data inputs by an 8-bit bus 8.

Details of the digitizer unit 35 of FIG. 5 are shown in FIG.

In FIG. 6, the data bus 37 is connected as an input to the data-in latch 45 and receives the output from the tristate gate 46. Data-in latch 45 and tristate gate 46 are connected to a common data bus 47, shown as the B bus. The data is the input going to and from the digitizer unit of FIG. 6 through the B bus 47 and the data bus 8.

In FIG. 6, A bus 48 is used to carry an address to address register 49. Address register 49 has its output connected to a 12-bit address bus 38,
This address bus gives an address to the wave table 13 in FIG. The four lower output bits on bus 50 from address register 49 are encoded and specify one of 16 voices. The bus bar 50 is connected to the voice matching comparator 52 as an input unit.

In FIG. 6, Mu latch 53 receives data from bus 47 when latch signal Mu-L is indicated. The latch 53 is shown by bit positions c0, c1, ... C7. The 8 bit output from latch 57 is sent to multiple inputs. The high bits c1, ..., C7 are sent as an input to the zero detector 40. Zero detector 40 detects when bits c1, ... C7 are all zero and provides an output on line 98. The output on line 98 is the latch signal to the 1-bit mode latch 91. Mode latch 91 is latch 53 to c0
Get a bite. When the c0 bit latched in latch 91 is a logic zero, it indicates that the parameter mode is selected. If the bit in latch 91 is a logic one, it indicates that the pitch / amplitude mode is selected. The output from latch 91 is sent as an input to NOR gate 97. Gate 97 receives the other input from zero detector 40 and the c4 bit from latch 53. Zero detector
When 40 senses a non-asserted output that all of bits c1, ... c7 are not logic 0's, and latch 91 stores one 0's, indicating a parameter mode is called, gate 97 turns on c4. Is zero and the latch signal is provided to the speech latch 90. Voice latch 90
Responds to the latch signal from gate 97 by the Mu latch 53.
Memorize bits c0, ... c3 from. In this way, a new voice is selected, allowing different parameters for that voice to be changed. The output from the audio latch 90 is provided to the audio matching comparator 52 as another output. The output from comparator 52 is addressed on line 54 if the audio shown in audio latch 90 corresponds to the audio being addressed by address latch 49.

In FIG. 6, the output line 54 from the voice matching comparator 52
Applies an enable input to AND gates 94 and 95. Data 9
4, 95 are therefore enabled whenever the voice identified by latch 90 is the same voice being addressed by address latch 49.

The other input to AND gate 94 is the output from mode latch 91 on line 85. Therefore, voice matching occurs,
Gate 94 will be satisfied whenever mode latch 91 indicates that the unit is in pitch / amplitude mode. AND gate 94 controls the operation of multiplexer 92.

Multiplexer 92 receives one data input from Mu latch 53 and another input from bus 47. When gate 94 is satisfied, multiplexer 92 selects the lower input from latch 53 to shift Pbot stage 56 ₁ of shift register 56 _1.
Load a new pitch or amplitude value into. Gate 94
Is not satisfied, the multiplexer 92
Select the data derived from the Ptop stage 56 ₁₆ of the shift register 56 on the bus.

Similarly, AND gate 95 is satisfied when mode latch 91 is in parametric mode. then,
The gate 95 controls the parameter multiplexer 93 to select the lower input, and the Pbot stage 55 of the shift register 55 is selected.
Load -1 with the new parameter value from Mu latch 53. When gate 95 is not satisfied, multiplexer 93 selects Ptop data from stage 55 ₁₆ via bus 44 and reinserts it into Pbot stage 55 ₁ .

In the above manner, in response to a command in latch 53, a new pitch or amplitude may be inserted into the corresponding voice location of shift register 56. Alternatively, the new parameters may be inserted into the corresponding voice location of shift register 55.

In FIG. 6, the arithmetic unit 62 is an ordinary 12-bit device, and is particularly adapted to add the 12-bit data received at each of its two input ports. The arithmetic unit 62 is 8
It is divided into a high bit part and a low bit part. The carrier outline 43 signal from the lower part is ORed to form the carrier in to the higher part. Carrier 1 on line 43
Is used to add +1 to the higher 8 bits under the control of carrier unit 64 when the lower four bits in latch 61 have been cleared to zero by the ZB signal. One of the input ports for the arithmetic unit 62 is 12 bit A
It is sent from the latch 60, and the other input port is sent from the 12-bit B latch 61. A latch 60 inputs its input to A bus
Pull out from 48, B-latch pulls its input from B-bus 47. The 8 bit R latch 65 provides the data from the high 8 bits of the B bus 47 to the corresponding high 8 bits of the A bus 48. The 12-bit T-latch 66 is an A bus 48 for use as a register to store a normal write pointer address.
Interact with. The 13-bit output from the arithmetic unit 62 is 13
Sent to Bit C Latch 67. This C latch 67 is C-
When the R / A signal is indicated, the lower 12
B bus 47 when bit is sent and C-R / B signal is shown
Send high 8 bits directly to. By selecting the higher 8 bits from the 13 bit latch 67, the shift of 1 bit is effectively performed. That is, the arbitrary 8-bit number of C latch 67 is divided by two.

In FIG. 6, the collapse / calculation unit 58 is the field.
Coherence bits from fields 55-2 of Ptop stage 55 ₁₆ by pitches c0, c1, c2 from 55-1.
It is a selector that selects different probability values from the random bit generator as a function of velocity specified by c3. The output from this unit 58 is PROBd, "PROB 1/2"
It is a line.

The output on the PROB 1/2 line is a logic one for half the time and a logic zero for the other half of the time except when the coherent bit c3 is shown. In this case, the output on the PROB 1/2 line is the same as the current cycle of the previous cycle.

The PROB 1/2 line is one of the inputs to the control logic circuit 71 and is also the input to the AND gate 39. Gate 39 pulls its remaining input from c7, the dither bit from the output field of stage 55 ₁₆ . The output from gate 39 is one of the control inputs to carrier unit 64.

The PROBd line output from the unit 58 has a probability of 1 being 3 bits output from the field 55-1, the command bit c0,
When controlled as a function of c1 and c2, it is in the 1 or 0 state. In this way, the PROBd line has a logic state of 1 or 0 and relative probabilities can be chosen.

The PROBd line is sent to the carrier unit 64 as one input and to the E latch 59 as another input. E
The input to latch 59 selects either the true or intended value of the quantity that is qualified for E-latching and sends it to B-bus 47 as a function of the 1 or 0 states of PROBd, respectively.

In FIG. 6, the PROBd line is connected to the carrier unit 64 as another input section. The carrier unit 64 is a selector unit that selects either the PROBd line or the output from the AND gate 39 as a signal for controlling the carrier line 43. Control logic circuit 71
The gate 39 line is selected when the "d _k " line from is shown. When the control line line is shown from control logic 71, the PROBd line from unit 58 is selected.

The effect of selecting the PROBd line during the collapse operation is to make it possible to prolong the collapse. In the non-collapse delay operation, the carry-in from unit 64 on line 43 adds +1 to the read pointer address to allow the calculation of the read pointer +1 address. However, in the case of collapse prolongation, the addition of +1 is prohibited a certain number of times, so that the read pointer address is used twice as much as when using the read pointer +1 address. Under these conditions, no change is made to the read data value from the read pointer address. The same data value is returned to the line pointer address. +1
The larger the frequency at which addition is prohibited, the longer the extension. The PROBd line is also used during repellant roadside operations and selects different initial amplitude values.

Digital Control The control of the digital unit shown in FIG. 6 is implemented by a control (CTRL) logic circuit 71. The control logic circuit 71 continuously circulates seven control cycles. At this time, each cycle has a first phase and a second phase. The control cycle is 1 ₁ ,
It is shown by 1 ₂ ; 2 ₁ , 2 ₂ ; ...; 7 ₁ , 7 ₂ . Cycle 1 ₁ ,
... 7 ₂ All are collectively called one logic array, or simply a logic cycle.

A number of control lines 73 are outputs from the control logic circuit 71 and are connected through the unit of FIG. 6 to the device of FIG. 5 in the case of some masters.

The control lines 73 form the control logic circuit 71 whose functions are shown in Table I below.

In Table I, Postscript L for each of the signal lines indicates that the latch to latch function has occurred. Postscript R indicates that the read (gate out) function from the latch circuit has occurred.

Table II below shows the binary and other states of each of the lines shown in Table I for each phase of the seven control cycles generated by the control logic circuit 71 of FIG.

P for the lines E-R and C-R / B, the signal is
Register stage 55 ₁₆ stages 55-5,
It is determined by the c5 and c6 bits from 55-4. Bit
When both c5 and c6 are 1, P is 1 and is 0. P is 1
, It indicates a flipping cycle which selects the amplitude from the E latch 59 to the B bus 47. When P is 1, the output data is selected from C latch 67 to B bus 47. Line C-L
Signal is in the state of 1 or 0 of line-L. The R signal on line-L is 1 in harp mode. That is, the local parameter bit c4 is 0 and c5 is 1.
Is. In drum mode (0 for both c5 and c6), R is the "PROB 1/2" signal from unit 58. The D signal on the "read" line is the c7's complement. Z-B
In the line, the symbol φ indicates that either 0 or 1 can be present.

Command Control The operation of the digitizer unit in FIG. 6 is under command control. Each command has 8 bits, and these 8 bits are shown as c7, c6 ... There are two methods of interpreting commands, parameter mode and pitch / amplitude mode. During the parameter mode, parameters useful for the operation of the FIG. 6 device are loaded from an external source, for example, input unit 2. During pitch / amplitude mode, the pitch or amplitude is specified from an external source.

In phase 1 ₁ of each logical cycle, data from the data-in latch 45 is stored in the Mu latch 53. During this phase 1 ₁
The cycle mode is detected and the mode latch of FIG. 6 is detected.
91, and determines the Pbot data entry location for the control shift registers 55,56.

To enter a parameter, all command bits on bus 7 are zero. To enter the pitch / amplitude mode, on the bus 7, command bits c7, c6, c5, c4, c3, c
2, c1 are all 0, and command bit c0 is 1.

Parameter mode Parameter mode (mode latch 91 set to 0)
Then, there are two types of parameters. That is, the wide range parameter determined when the command bit c4 is 0, and c4
And the local parameter when is 1. To explain the command, a high-order nybble containing four high-order bits c7, c6, c5, c4 and four low-order bits c3, c2, c1,
It is good to divide it with the lower nibble including c0. Each of these bits can have a binary state of 1 or 0 represented by hexadecimal characters. For example, when the four high-order bits are 0, the hexadecimal character _0h indicates that all bits are 0s. If the four lower bits is 0011, hexadecimal characters 1 _h is used to indicate that these bits have the respective values. The hexadecimal character _Fh is used to indicate that all four bits are ones. Hexadecimal characters are indicated by the subscript "h", which represents the base hexadecimal number. That is, each hexadecimal character can be expanded into four binary bits.

Parameter mode command codes are shown in Table III below.

The various parameters shown in Table III are for obtaining many different changes in the fundamental tone of the plucked string. In this embodiment, there are three types of repulsive musical instruments defined by high nibbles. Drum (1 _h X _h ), guitar (3
_h X _h ) and harp (5 _h X _h ). In addition to these three types of instruments, the bias, that is, "repellency" is a local parameter 7 _h.
Specified by X _h . The presence or absence of modified bits (called dither bits) is also controlled by the local parameter high nibble. Character X _h of Table III shows that Naiburu may have any value.

In Table III, the local parameter lower nibble for the last 16 entries specifies the collapse property.

The eight entries in Table IV below show representative examples of the specified parameters.

In operation, the data values of the command bits c2, c1, c0 during the local parameter mode are stored in the 3-bit field 55-1 of the shift register 55. The coherence control bit c3 is stored in the 1-bit field 55-2. The control bits c7, c6, c5 are 1 bit field 55-3, 55-4,
It is stored in 55-5.

Mode latch 91 indicates the presence of the local parameter mode, when the identification information indicates that the comparator 52 is of the voice is right at the bottom stage 55 ₁ of the shift register 55, the local parameter shifted from the B bus 47 Gate to bottom stage 55 ₁ of register 55. Table II cycles (both phases)
Shift register stage 55, during each logical array cycle (logical cycle) including all 1 through 7,
The contents of 56 are shifted one stage. That is, the data of the bottom stages 55 ₁ and 56 ₁ are shifted to the next adjacent stages 55 ₂ and 56 ₂ which are performed one after another and finally
55 ₁₅ and 56 ₁₅ are shifted to the top stages 55 ₁₆ and 56 ₁₆ . 16
5 stages 55 ₀ , 55 ₁ , ... 55 ₁₆ and 56 ₁ , 56 ₂ , ...
56 ₁₆ corresponds to the 16 different voices in FIGS. 5 and 6.

Many different control parameters can be stored in shift register 55. However, the basic operation of the instrument is PM
These control parameters are independent and unchanged, as will be explained in connection with the operation therein.

Pitch / Amplitude Mode During pitch / amplitude mode, each note is constructed by an initial flutter at some amplitude, followed by the collapse of some designated pitch. In FIG. 6, sixteen 8-bit stages of shift register 56 are used during the "repelling" period to store the maximum amplitude of this repelling and to store the pitch period during the subsequent collapse period.

Pitch / amplitude mode is initiated when the first 1 0 _h 1 _h encoded in the phase of 7 cycles logical cycle of Table II was detected in the Mu latch 53. If pitch / amplitude mode is becoming an entry for the first time associated with the sound of a given pitch, 0 _h 1 _h amplitude flap consonant after the code is followed, the bottom stage of the amplitude shift register (Pbot) Stored in 56 ₁ . This stored amplitude is used to fill the wavetable unit 13 of FIG. 5 with the appropriate initial values.
Whether the amplitude value is positive or negative, the sixth
It is stored in each location of the wave table for that voice as a function of the 1 or 0 output of the random bit generator 57 shown.

The way in which the circuit of FIG. 6 loads the amplitude into the wavetable is as follows. In the first cycle 1 _1, the next address Rokeshiyon obtained from C latch 67 is stored in the T latch 66. Amplitude values from Ptop stage 56 ₁₆ is accessed and transferred through B bus 47 to the E latch 59.

In cycle 6 _1, address T latches 66 is transferred to the address latch 49 and the A latch 60. B latch 61 is a
-1 loaded, in the result, the cycle 7 _2, C latch
The 67 address is decremented by -1. In cycle 7 _1, E
The plus or minus amplitude value stored in latch 59 is gated through data out gate 46 and stored in the storage location of address latch 49. This process is repeated until the wavetable is filled with positive or negative amplitude values. While the wavetable is being filled, output data equal to the positive or negative amplitude value is sent to the output unit 4. These amplitude values make up the flipping sound. Flipping sounds exist until the wavetable is full of positive or negative amplitude values.

At the appropriate time, designated by the commands to end the flip phase and start the collapse phase, the pitch / amplitude mode will re-enter and the pitch frequency N will be placed in the bottom stage (Pbot) 56 ₁ of the shift register. When the logic cycle is complete, the command is executed for each of the 16 voices.

In the collapsed portion of the pitch / amplitude mode, the instrument of FIGS. 5 and 6 continues to operate as follows, as shown in equation (5).
The device of FIG. 6 uses a common write pointer for all 16 voices. This write pointer indicates the address of the wave table which is going to store the data value which is changed at the same time. The write pointer is stored in the T latch 66 of FIG. The lower 4 bits of the T-latch represent the voice field of the address.
Corresponds to the voice field output from address latch 49 on 50. The eight high order bits in T-latch 66 (correspondingly on bus 51 from address latch 49) represent the write address in the wavetable location for any particular voice. Each logic cycle (7 in Table II)
Cycle), the write pointer in latch 66 is decremented by one count.

In operation, the read pointer for each voice is calculated at each logic cycle by adding the number of bits N obtained from the top location 56 ₁₆ (Ptop) of the shift register 56.
In the embodiment of FIG. 6, the address of T-latch 66 is decremented by one each logical cycle. Therefore, latch 66
N location behind the light pointer inside is a latch 66
It is obtained by adding N to the address in. When the latch 66 address is incremented, this N will be subtracted from the latch 66 address. The read pointer selects the data previously stored in N cycles. This data is modified and stored at the address specified by the write pointer.

In this example, the modification is according to equation (5). The data designated by the "read pointer + 1" is the data N + 1 address apart from the address of the T latch 66. N and N + 1
The location data is summed, divided by two and rewritten to the address specified by the write pointer (in T latch 66).

Normal collapsing operation implements the modified portion of equation (5) in the area shown in Table V when no new designation is given.

In Table V, the last cycle 7 _2, the address in the T register by 1 reduced. The the reduced value is filed in the write pointer, the new logic cycle is stored in the C latch 67 when starting in cycle 1 _1. Cycle 1
_{At 1} , the write pointer is gated from latch 67 to A bus 48, from which it is gated to T latch 66 and A latch 6.

In cycles _1, it appears at the B bus (represented by the lower bits of the T latch 66) bit length N of the current voice (indicated by Ptop in Table V). This value is latched in the B lattice 61. In cycle 1 _1, C latch in pitch length N of T values and B latch 61 of the A latch 6 are added by the adder 62 in cycle 1 ₂ 67
Give the read pointer in.

In cycle 1 and ₂ , the read pointer in the C latch is the A bus 48
Through address bus 49, where it propagates through bus 38 to address the wavetable 13 of FIG. The wavetable 13 thus addressed provides the data on the bus 8 and stores it in the data-in latch. Data in latch in cycle 3 ₁
The data-in value of 45 is gated through the B bus 47 and stored in the R latch 65. In cycle 3 _1, Kiyariin-Yunitsuto 64 are subject to a +1 A, the sum of the contents of the B latch conditionally, it'll connexion, add +1 to the value of C latch. From the previous values in C latch was found to be the read pointer (T + N), a new number in the C latch after the cycle 3 ₁ becomes read pointer +1 (T + N + 1) .

In cycle 4 _1, the read pointer +1 in the C latch 67 is A
Transferred to address latch 49 via bus. From the address latch 49, the read pointer +1 to address the wave enable 13 to latch new data values to the data-in latch 45 in cycle 5 _1.

In cycle 5 _1, data values obtained cowpea the read pointer is gated to the A latch 60 through R latch 65, other data values from the read pointer +1 B through B bus 47
Gated to latch 61. These two data values are then added by the adder 62 and the C latch 67 in cycle 5 _1.
Give the total to.

In cycle 6 _1, the write pointer from T latch 66 is gated to the address latch 49 and the A latch 60 through A bus 48.

In cycle 6 _1, Purisetsuto value of -1 on the B bus is latched in the B latch, then the adder 62 to be added to the write pointer of Yotsute A latch new write pointer (T
-1) is formed. It is latched to the C latch in cycle 7 _2. Also, in cycle 6 _1, a total of either C latch 1
It is gated out from the C latch 67 to the B bus 47 by bit shift, gated out to the bus 8 through the tristate gate 46, and stored in the wavetable unit 13 at the write pointer (T) address. 2 at this point
The two transferred data values are read pointer and read pointer + 1 address, and are fetched, added, and averaged according to the equation (5) described above. Also cycle
7 _2, decrement the write pointer (T-1) is formed,
It is stored in the C latch 67.

The new value of the write pointer defines a different voice because the lower four bits have been changed. Similarly, the shift register 56 undergoes a step operation for one stage, and as a result, the bit number N previously in the Ptop location 56 ₁₆ is returned to the Pbot location 56 ₁ through the B bus.

New values for bit length for different voices are now Ptop
Stored in Rokeshiyon 56 _16. This bit value is reused to form the read pointer by executing a complete logic cycle of the Vth table format.

This calculation for each of the 16 voices is done by 16 executions of the V-tabular logic cycle. afterwards,
T-Ratch 66 continues to decrease. At this time, the carry-out after decrementing 16 times is performed up to the high-order 8 bits in the T latch 66, so that the new location in the wave table is accessed for each voice. In this way, all of the locations in the Wavetable Unit 13
For each of the 16 voices, it is accessed in the manner that the calculation described above in relation to equation (5) is performed for each voice.

Transfer the sample to the output Yunitsuto 4 of FIG. 5 occurs once at the end of Table V cycle in cycle 7 _1. After executing Table V logic cycle 16 times, 16 samples, one for each voice, are output to the output unit 4. Then the V
With 16 or more executions of the table logic cycle, 16 or more samples are output to the output unit 4, one for each voice. It has to be noted here that the outputs of the output unit 4 are not added, but only time multiplexed one at a time for each voice. Each sample is converted into an analog signal by the D / A converter 9, and this signal is low-pass filtered by the filter 10. The signal on line 18 represents the sound from all 16 voices, such as when the outputs on bus 8 are summed before being converted in D / A converter 9. Time-multiplexing through the D / A converter 9, adding low-frequency waves first adds, and then D / A
It is the same as performing A conversion.

The output signal on bus 8 occurs at a sampling frequency of about 20 KHZ for each voice. 1 for all 16 voices in a cyclical manner
Since two outputs are provided, a new signal appears on the data bus 8 at a rate 16 times as fast as the sampling frequency of 320 KHz (which is the logic cycle frequency). The logical cycle frequency is the time taken to complete all seven cycles in Table II. Each cycle within the Table II format logic cycle occurs at seven times the frequency of the complete logic cycle, or 2.24 MHz. The values of the sampling frequency, the logic cycle frequency and each sub-cycle frequency within the logic cycle are shown merely as an example. You can choose the frequency of your post. The preferred embodiment described here is 2 MHz
Operated at a fundamental clock frequency between 3MHz. Therefore, the sampling frequency Fs for each voice is 1/112 of the clock frequency of DY35. In this example, the sampling frequency Fs is the same for all 16 voices.

Command Sequences In the musical instruments shown in FIGS. 5 and 6, NY2 gives input commands in an appropriate sequence. In this command sequence,
The amplitude that defines the amplitude of a single repellant is entered in the shift register 56 as part of the repellant performance mode. Similarly,
The pitch number that defines the pitch of each voice is entered as part of the collapse mode for that particular voice. In this connection, note that the two values for the command code prohibit the amplitude or pitch number from being entered. The two prohibited values are 0h 0h and 0h 1h. These values are prohibited because they are used in the conversion between pitch / amplitude mode and parameter mode. However, these values are not particularly useful in the pitch. This is because it is the frequency when there is no pitch of 2. Moreover, to this, these amplitudes are also obtained by using their inversions (Fh Fh, Fh Eh). When performing symmetrical flip, the maximum amplitude is Fh Fh
And the minimum is 8h 0h. Typical examples of command sequences for playing the maximum amplitude guitar sound are shown in Table VI below.

It is useful to use a simple example to show that the instrument of FIG. 5 performs the function of equation (5). As an example, M
Use a storage device with a number of storage locations (M is 10). Here, it is assumed that the pitch length N is 6. The write pointer (think only by high order bits in combination with a single voice) has the values 9, 8, 7 ... 1,0.

Assume that the ten locations 0, 1, ..., 9 of the storage device are initially filled with data values A, B ,. Each of these data values has either a positive or negative quantity as determined by a 1 or 0 in the random bit generator. The logical cycles in Table II are numbers 1, 2, 3, ... Corresponding to cycles for only one voice.
・ ・ 18, ... are attached. Actually, one for each voice
There are 16 times the number of cycles to the destination, but only the cycles corresponding to one voice are numbered. Under these simplified assumptions, Table VII shows the relationship between the write pointer, the data stored at the storage device address associated with the write pointer, and the read pointer, read pointer + 1.

In Table VII, prior to the logic cycle 1, the stored data are the data values A, B, ...
It is J. These values are, for example, +8, -8,-
It may be 8, +8, -8, +8, +8, +8, -8, +8.

In logical cycle 1, the storage location 9 indicated by the write pointer is read pointer and read pointer +
It is satisfied by the average value of the amounts accessed at the address defined by 1. The read pointer indicates the address 5 stored with the data value F. The lead point +1 indicates the address 6 of the data value G. Therefore, in the logical cycle 1, the stored data is (F + G) / 2, that is, the value given as an example above is 8.

In logic cycle 2, the stored data is (E + F) / 2, or 0. Similarly, each cycle is a logical cycle 6
The above is the same, and the recording location of the write pointer 4 is (A + B) / 2, that is, 0.

In logic cycle 7, however, the read pointer shows the location 9 stored at (F + G) / 2 in logic cycle 1. Logic cycle 7 is the sixth cycle counting from logic cycle 1. The value averaged by the values stored in the logical cycle 1 is the value stored in the cycle before the write pointer cycle 10, that is, the initial data value A. Data value A is 7 cycles displaced. Therefore, the average value between the 6-cycle displacement and the 7-cycle displacement is 6 1/2, that is, the pitch number is N + 1/2 cycles.

Summary of 16 Voice Examples Each of the 16 voice examples of FIG. 5 has a voice that is independently controlled for a pitch value determined by the pitch number N. Each voice has a sampling rate of about 20 KHz. Each sample time is 16 voice cycles and each cycle is a logic cycle consisting of 7 clock cycles. The sampling frequency is 1/112 of the clock frequency of DY.

Each voice has one of four modes: rehearsal, guitar, collapse, drum collapse, and harp collapse. Each of these disintegration algorithms allows for disintegration strech reduction during disintegration motion, with the striking factors 1, 2, 3,
4, 8, 16, 32, 64, infinite. The factor s multiplies the stretch collapse by the value of s. Stretching is intended not to increment the read pointer by +1 in the select logic cycle.

During the repulsive performance, the output to the D / A converter in the output unit 4 is randomly amplitude A or 225-A, that is,
It can be the complement. The probability of reversal to 225-A is 1 / S,
Here, S is a stretching factor determined by the "PROBd" control for the E-latch 59 in FIG. Under these conditions, the D / A converter 9 of FIG. 2 is located at 128.

For guitars and drums, the blend factors (probabilities of choosing the complement from 1 minus C latch) are 1 and 1/2, respectively. Harp is a drum with a blending factor of zero. The blend factor of 0 is
It means that the complement is always selected from C latch. Therefore, the values in the wave table are complemented for each bus, the frequency is lowered by one octave, and only odd overtones are left. This operation delays the range down by one octave and adds a somewhat unusual tone to the higher octaves.

The ditherbit c7 is given as an option to offset the effects of round-off errors.

The coherence bit c4 is provided as a means of connecting several voices. This technique can be used to increase the total amplitude of a sound over that which can be achieved with a single voice. It can also be used to give a "sound enhancement" at the beginning of a note by first exciting the two coherent voices with equal and opposite amplitudes (perfect cancellation and hence silence). The reverse amplitude turns off the coherence bit.

In the embodiment shown in FIG. 5, the digital unit 35 tests the input bus 7 from the input unit 2 only once per logic cycle (once every 7 clock cycles in the SpCy cycle). Therefore, the control byte must be held by the interface unit 6 for at least 7 cycles before being issued a new command. The interface unit 6 may be any conventional device, such as a control register gated out from the microprocessor chip or digital unit 35 by the SpCy signal. Further, the control storage device of the digital unit 35 is the sixth.
Since the shift registers 55 and 56 shown in the figure are used, the command that affects only one voice is that the voice has a bottom location.
Must be held until stepped to 55 ₁ or 56 ₁ . Therefore, a command intended to be effected by one voice should be held by the interface unit for at least 112 clock cycles.

Other Embodiments Above The 16-voice embodiment described above utilized a common write pointer and different read pointers calculated for each voice. The sampling frequency f _s was the same for each voice. Since these conditions are normal, the present invention is not limited thereto. Furthermore, the write pointer and the read pointer can be independently determined for each voice and each sampling frequency f _s , and can also be determined separately for each voice.

Considering the modifier unit 14 as an 8080 microprocessor in the embodiment of FIG. 2, this is yet another embodiment of the present invention. A suitable program for making changes in such a microprocessor example is shown in Table VIII.

In Table VIII, the entry point is START. The registers in the 8080 processor are C latch, DE register and HL.
Includes registers. DUR is a storage device location. The C register stores the lower half of the write pointer. The DE register stores the read pointer ₂ for voice 2. The H register stores the address of the current byte. Therefore, the DUR performs the step operation through the count number of 256 samples. After counting through 256 counts, the DUR register begins counting through the next set of 256 counts, wrapping around to do this one after another.

In Table VIII, the program routine is excited whenever the decay time expires by advancing from instruction 30 to instruction 31. If the collapse time has not expired, command 30 jumps to CONT command 15 and continues processing.

The method of determining the change of the equation (5) is that of the circular buffer technique. The common light pointer is shared by all voices. Each voice has a separate read pointer. Both the write pointer and the read pointer undergo a step operation once per LOOP execution. The pitch number N for each voice V is not explicitly stored, but rather the difference between the write and read pointers for that voice. The write pointer for voice 1 is formed using the B, C registers by moving the contents of register H to B. The write pointer ₂ is formed by moving the contents of register D to register B. The lower byte of register C is incremented or decremented in register pair BC without servicing the higher byte B.

The routine in Table VIII uses timer D once every 256 samples.
Handle the sampling frequency timing by decrementing the UR.

Although the two-voice embodiment is another embodiment of the wavetable modifier method of the present invention, a particular implementation using current microprocessor technology is suitable for processing two voices. Of course, the more powerful the microprocessor, the more processing it will perform, so the circular buffer technique described above can be used to process a much larger amount of speech.

In the example of FIG. 5, the higher the overtone, the faster the decay rate, and thus the sound decays to almost a pure sine wave, which is evenly decayed to a certain value (silence) in the initial spectrum. With or without

Many initial states can be specified. Specifically, this involves preloading the wavetable with the appropriate values. The initial value may form a sine wave, a triangular wave, or any other desired waveform. However, in general, it need not be so complicated. Since it is desirable to have many high overtones first, the buffer of the FIG. 5 embodiment is filled with random values. This produces a repulsive string sound, much like a guitar. One of the fastest ways to fill the buffer is to use two-level randomness. Mathematically speaking, the initial state is given as follows, where n is between N and 0.

y _n = + A, probability 1/2 (29 ₁ ) y _n = -A, probability 1/2 (29 ₂ ) The single bit randomness of this form is fed back shift due to the random bit generator 57 of FIG. It easily occurs in the register. Such an embodiment is simpler than a fully random language generator. The parameter A is for amplitude control, and the output amplitude is directly proportional to A.

After playing a note, it is always necessary to reload the buffer with a random value before playing the next note. If the sound does not decay too early, a slur is generated between the two pitches. This technique is particularly useful when using the circular buffer technique. This is because the increase in N can be captured more in the previous sample. A similar increment using a decrement counter will obscure the value passing through the buffer (wave table) bridge.

If the initial buffer load is periodic and this cycle elicits N, then the sound will be a harmonic of N, with a pitch corresponding to the positive cycle of the buffer load. This overtone trick is done by filling one half (or one third) of the buffer with randomness and adding these samples to fill the buffer rest. This gives a way to extend the yes pitching sound, since the short buffers (N small) decay faster. The disintegration stretching method described below is more general, powerful, and
This is a fast method.

One way is to replace y _nN-1 with y _{n-N + 1} and change the pitch to N-1 / 2 instead of N + 1/2. In the single-speech algorithm, in this variant, the period N can be corrected by using the extra time of wraparound in the decrementing counter technique. If this extra time is set to half the normal sample time, the average sampling rate is T
(1 + 1 / 2N). This has a sound frequency of 1 / [(T)
(N−1 / (2N)].

Reducing the disintegration time is more difficult than prolonging it. One possibility is to change the iterations to make the waveform smoother faster. For example, y _n = x _n [y _n [y _nN-1 ^{+ 2y} _nN ^{+ y} _{n-N + 1} ] / 4 (30) The algorithm of Eq. (30) enhances computational power and shortens the decay time. Offset by an increasing amount of time to compute a sample. The modified example described below can be applied to equation (30) more easily than the algorithm of equation (5).

A simple variant of the basic equation (5) produces a drum timbre. The simplest explanation of the drum example is the following stochastic circular relation.

_{y n = 1/2 (y} nN + y nN-1), the probability _{b ··· (31 1) y n} = -1 / 2 (y nN + y nN-1), the probability b-1 ··· (31 ₂ ) The normal initial state is 2-level randomness.

Parameter b is called the blend factor. When the Bledfactor is 1, the algorithm reduces to the basic plucked string algorithm, N braking the pitch. When the blend factor is half, the sound is like a drum. The middle value emits a sound in the middle between the repelling string and the drum,
Some of them are musically very interesting. 1/2
Smaller values are also interesting. Note that b = 1/2 requires only a single randomness bit for each sample. When using any value for b, a comparison with all random languages is required.

If b is close to 1/2, the buffer length does not control the pitch of the sound. Instead, it controls the decay time of the noise burst. Large N (about 200), sampling time about 50
For microseconds, the effect is a Nesua drum. For small N (about 20), the effect is tom tom. Intermediate values give an intermediate timbre, allowing a smooth transition from one drum to another. In these drum sounds, buffers are seen with a minimum constant (A), as the algorithm produces randomness itself.

Using a small language size (8 bits) causes round-off error problems. In the above algorithm,
The round-off error is not random but causes a matched rounding down of the sample. This effect sufficiently reduces the decay time of the fundamental frequency (when compared to the theoretical decay time, or when the algorithm has a large linguistic size). This effect can be mostly eliminated by randomly adding 0 or 1 to y _nN + y _nN-1 before dividing by 2. This bit-to-ziddle technique extends the basic final collapse almost to its logic collapse time. Moreover, the initial attack of the sound is not lengthened.

For longer decay times (plucked strings or drums),
Disintegration stretching may be used. In the case of a drum, this has the effect of increasing the "shining" sound, with a small N
Allow the value to be used. The circulation relation for the stretched sound is as follows.

y _n = + y _nN probability b (1-b) ... (32 ₁ ) y _n = -y _nN probability (1-b) (1-d) ... (32
_{1) y n = + 1/} 2 (y n-1 + y nN-1), the probability _{bd ··· (32 3) y n} = -1 / 2 (y nN + y nN-1), a probability (1-b) d ...
(32 ₄₎ A new parameter d is called the decay rate multiplier, 0 to 1
Is in the range. Sometimes, for convenience, Stretch actors
Try as = 1 / d. Note that the collapse rate multiplier and the blend factor are independent, the algorithm is run in two separate tests and does not require any multiplier. The decay time of a sound is approximately proportional to s. The pitch of the sound is also affected by d. This is because the time is about N + 1 / 2d. The optimal choice for d depends on the sampling rate N and the desired effect. Yotsute to selecting d to approximately proportional to N or N ^2, so that the higher pitch of the decay rate is comparable to a lower pitch of the decay rates. d
Note that when = 1, the cyclic relationship is simplified to that of the steam-stretch algorithm. When d = 0, the sound does not collapse. When b = 1, this is the wavetable synthesis algorithm of equation (5), and when b = 1/2, white noise occurs.

If no drum sound is desired, b is set to 1 which can simplify the algorithm. In the case of a random buffer load, the sound is a plucked string and the decay time is proportional to d. When a non-random buffer load is used with b = 1 and the value of s is large, a woodwind sound is produced.

The above example uses a single sampling frequency f _s for all of the one or more voices. Of course, it may be different for each f _s voice. For example, the clock frequency for the clock unit (CLK) shown in FIG. 6 may be made variable by the program command of the quantity Q from the bus 47 which controls the frequency in the control logic circuit 71. The clock frequency is Q
It is also possible that each voice is given a different value of Q, and each voice has a different sampling frequency. Also, the sampling frequency of any voice may change as a function of time by changing Q as a function of time.

Embodiments of the present invention contemplate the use of a single D / A converter for all of the voice of a multi-voice instrument. In another embodiment, each audio may have its own D / A converter and sum the analog output from multiple converters. For example, this is done with a summing amplifier in front of the low pass filter 10 of FIG.

Although the present invention has been described with reference to preferred embodiments, those skilled in the art will appreciate that various changes and modifications can be made without departing from the scope and spirit of the invention.

[Brief description of drawings]

FIG. 1 is an electric block diagram of a musical instrument incorporating the present invention. FIG. 2 is a developed electric block diagram of the musical instrument of FIG. FIG. 3 is a schematic electric block diagram of a modifier unit which forms part of the wavetable modifier generator of the instrument of FIG. FIG. 4 is a graph of SS amplitude versus frequency during the first 15 periods for a representative note of a musical instrument of the type shown in FIGS. FIG. 5 is an electrical block diagram of one embodiment of the present invention utilizing a digital semiconductor chip. FIG. 6 is a schematic electric block diagram of a digital chip used in the musical instrument of FIG.

Claims (15)

[Claims] 1. A readable / writable storage means for storing a waveform sample signal; a first address designating means which is stepped by a clock signal and which designates a first address based on the step value. Pitch instructing means for supplying numerical data corresponding to the pitch of the musical tone, and changing the address value instructed by the first address by the numerical data supplied by the pitch instructing means. Second address designating means for designating a second address, and the waveform sample signal read from the storage means are input, processing for changing or not changing the waveform sample signal is performed, and the output signal is input to the storage means. Waveform processing means for performing waveform processing from the storage means by using one of the first address and the second address as a read address. Control means for reading the sample signal and writing the output signal of the waveform processing means in the storage means by using the other of the first address and the second address as a write address. A tone generation system characterized in that a difference corresponding to numerical data occurs and a tone signal having a periodicity corresponding to this difference is generated. 2. The second address indicating means is the first address
2. The musical tone generating system according to claim 1, wherein the second address is designated by a value obtained by offsetting the address value designated by the address of FIG. 3. The musical tone generation according to claim 1, wherein an arbitrary initial waveform is written in the storage means, and the stored waveform is gradually changed in accordance with the change processing by the waveform processing means. system. 4. The musical sound generating system according to claim 3, wherein the initial waveform is a noise waveform. 5. The musical tone generating system according to claim 1, wherein the first address designating means designates a write address and the second address designating means designates a read address. 6. The first address designating means stores storage means for storing a write pointer address for designating a current write address, and increases or decreases the write pointer address in accordance with the clock signal so that the storage means stores the write pointer address. Arithmetic means for updating the stored write pointer address, the pitch instructing means includes storage means for storing the numerical value data indicating the delay amount N as an address offset, and the second address instructing means 6. The musical tone generating system according to claim 5, further comprising calculation means for calculating a read pointer address indicating the read address by adding or subtracting N to the write pointer address. 7. One of the first address designating means and the second address designating means includes means for storing a write pointer address for designating a current write address,
The other includes means for storing, as the read address, a read pointer address offset from the write pointer address by the number of addresses corresponding to the numerical value data supplied by the pitch instructing means, and the first address instructing means, 2. The musical tone generating system according to claim 1, further comprising calculation means for sequentially changing the stored write pointer address and read pointer address according to the clock signal while holding the offset amount therebetween. 8. The frequency of the clock signal is constant,
The tone generation system according to claim 1, wherein the sampling frequency of the tone signal generated is constant regardless of the tone pitch. 9. The storage means is capable of storing the waveform sample signals for a plurality of voices, respectively, and the pitch instructing means has a numerical value corresponding to the tone pitch of each of the plurality of voices. Data can be supplied respectively, and the control means independently controls reading and writing of the waveform sample signal for each of the plurality of voices, and corresponds to the numerical data for each of the plurality of voices. The musical tone generating system according to claim 1, wherein a musical tone signal having a pitch to be generated can be generated. 10. The tone generation system according to claim 9, wherein said waveform processing means independently performs a process of changing or not changing said waveform sample signal for each of said plurality of voices. 11. The tone generation system according to claim 9, wherein one of the write address and the read address has a common value for each of the plurality of voices. 12. The first address designating means uses the clock signal having a common frequency for each of the plurality of voices, and a sampling frequency of a generated tone signal is common to each of the plurality of voices. A musical sound generating system according to any one of claims 9 to 11. 13. The control means controls the reading and writing of the waveform sample signal with respect to the storage means for each of the plurality of sounds in a time-divisional manner, respectively. A tone generation system according to any one of items. 14. The first address designating means and the second address designating means.
Of the address indicating means includes a means for storing a write pointer address indicating a current write address, and the other means is offset from the write pointer address by the number of addresses corresponding to the numerical data supplied by the pitch indicating means. A read pointer address is stored as the read address, and at least one of the write pointer address and the read pointer address is stored for each of the plurality of sounds. 10. The storage device according to claim 9, further comprising calculation means for sequentially changing the stored write pointer address and read pointer address in accordance with the clock signal while respectively holding the offset amount between them for each of the plurality of voices. The tone generation system described. 15. The tone generation system according to claim 1, wherein said waveform processing means performs a waveform changing process so that the overtones of the tone signal gradually decrease.