JPH0775873A - Inverter control type welding power source - Google Patents

Abstract

PURPOSE:To enhance the safety and reduce the capacity of an inverter transformer by operating a forward converter of each set by shifting phase in order. CONSTITUTION:AC input terminals Tm1 to Tm3 of the primary input is connected to one DC power source 1 in parallel or electric power is supplied from separate DC power sources and forward converters of n sets ((n) is an integer >=2) connected with output terminals in parallel are provided. A control circuit where the forward converter of each set is operated by shifting by one nth period in order is then provided. N pieces of capacitors to divide the output voltage of the DC power source 1 are provided. Consequently, a stable and safe arc welding machine with the low no-load voltage can be provided.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter-controlled power source for arc welding, and provides a stable arc welding machine which is stable and has a low no-load voltage against arc welding accompanied by a drastic load change. Is what you are trying to do.

[0002]

2. Description of the Related Art Conventionally, a forward converter has been used as an inverter control system having a stable operation in an arc welding machine with a drastic load change. An example of a two-stone forward converter is shown in Fig. 1, and its operation waveform is shown in Fig. 2. In FIG. 1, 1 is a DC power supply, 2 is an inverter 1
Next circuit, 3 is an inverter transformer, 4 is an inverter secondary circuit, 5 is a torch, and 6 is a base material. FIG. 2 is a diagram for explaining the operation of the device of FIG. 1, (a) is the base current of the transistors Tp1 and Tp2, and (b) is the transistor Tp.
1, collector current of Tp2, (c) is transistor Tp1,
Collector-emitter voltage of Tp2, (d) current of rectifying diode Ds1, (e) flywheel diode D
The current of s2, (f) is the voltage of the flywheel diode Ds2, and (g) is the current of the DC reactor L, that is, the load current.

In FIG. 1 and FIG. 2, as shown in FIG. 2A, the base current flows, so that FIG.
The collector current as shown in FIG. 2 flows through each transistor, and the collector-emitter voltage becomes as shown in FIG. While the base current flows through the transistors Tp1 and Tp2 and is turned on, a forward voltage is applied to the rectifier diode Ds1 to conduct it, and a reverse voltage is applied to the flywheel diode Ds2, and the secondary current is the rectifier diode Ds1 and the DC reactor L. , Torch 5, arc, base metal 6, and rectifier diode Ds1 on the secondary side of the inverter transformer 3.

When the base currents of the transistors Tp1 and Tp2 are exhausted, these transistors are turned off and a voltage opposite to that at the time of turning on appears in each of the primary and secondary windings of the inverter transformer 3. On the primary side, this voltage causes the diode Dp
3, Dp4 becomes conductive, and the inverter transformer 3 returns the magnetic energy stored when it is ON to the DC power supply 1. Next ON
By the time the return of the magnetic energy is completed, the voltage appearing in the winding of the inverter transformer disappears. On the other hand, during this period, the rectifier diode Ds1 is turned off and the flywheel diode Ds2 is turned on, and the magnetic energy stored in the DC reactor L supplies power to the load. At this time, the secondary current flows through the circuit of the DC reactor L, the torch 5, the arc, the base material 6, the flywheel diode Ds2, and the DC reactor L.

[0005]

When the output voltage of the DC power source is Eo and the times when the transistors Tp1 and Tp2 are ON and OFF are Ton and Toff, respectively, the voltage of the flywheel diode Ds2 is as shown in FIG. 2 (g). And the current flows at Toff. If the number of turns of each primary and secondary winding of the inverter transformer 3 is Np and Ns, the maximum value Vmax of the load voltage is

Vmax = Eo. (NS / NP)

The average value of the load voltage Vavr is

Vavr = Eo. (NS / NP). (Ton /
(Ton + Toff))

[0008] Therefore,

(Vmax / Vavr) = (Ton + Toff) / Ton

[0010] However, in the forward converter, it is necessary to provide a time for returning the magnetic energy at least for the time required for accumulating the magnetic energy, and the ON DUTY is limited to 50% or less. That is,

Ton / (Ton + Toff) <0.5

To be

(Vmax / Vavr) = (Ton + Toff) / Ton> 2

[0013] On the other hand, other inverter systems

Ton / (Ton + Toff) <100%

Therefore,

(Vmax / Vavr) = (Ton + Toff) / Ton> 1

It is

Even if a welding worker wears safety shoes and wears insulating gloves, there is a danger that a voltage is always applied between the torch and the base metal. When the inverter is operating and no arc is coming out, a small capacitance capacitor that exists as stray capacitance between the base metal and the torch or a small capacitance capacitor connected between the output terminals to prevent noise from entering It is charged with the aforementioned Vmax and appears as a no-load voltage. If this no-load voltage is high, it is dangerous for the welding operator. However, as described above, the forward converter has an ON DUTY, that is, Ton / (Ton + Toff) of 50% or less, so in order to obtain the same average voltage as other methods with an ON DUTY of almost 100%, for example, a bridge type inverter. It requires about twice the Vmax, resulting in about twice the no-load voltage, which is a high risk.

[0019]

The present invention provides a plurality of sets of forward converters in order to solve the problems of the above-mentioned conventional apparatus, connects the secondary side in parallel, and sequentially turns on and off each forward converter. By controlling it, the overall ON DUTY is improved and the difference between the average voltage and the maximum voltage is reduced.

[0020]

The present invention will be described below with reference to the illustrated embodiments.

FIG. 3 shows the first embodiment of the present invention. The inverter primary circuits 2a and 2b are the inverter primary circuits 2 of FIG.
The inverter transformers 3a and 3b have the same configurations as the inverter transformer 3 in FIG. 1, and the inverter secondary circuits 4a and 4b have the same configurations as the inverter secondary circuit 4 in FIG. Figure 4
FIG. 4 is a diagram for explaining the operation of the embodiment of FIG. In FIG. 4, (aa) is the base current of the transistors Tp1 and Tp2 of the inverter primary circuit 2a, (ba) is the collector current of the transistors Tp1 and Tp2 of the inverter primary circuit 2a, and (ca) is of the inverter primary circuit 2a. The collector-emitter voltage of the transistors Tp1 and Tp2, (ab) is the base current of the transistors Tp1 and Tp2 of the inverter primary circuit 2b, (bb) is the collector current of the transistors Tp1, Tp1 and Tp2 of the inverter primary circuit 2b, ( cb) is the collector-emitter voltage of the transistors Tp1 and Tp2 of the inverter secondary circuit 2b, and (da) is the inverter secondary circuit 4
current of rectifier diode Ds1 of a, (ea) current of flywheel diode Ds2 of inverter secondary circuit 4a,
(Fa) is the voltage of the flywheel diode Ds2 of the inverter secondary circuit 4a, (db) is the inverter secondary circuit 4
current of rectifier diode Ds1 of b, (eb) current of flywheel diode Ds2 of inverter secondary circuit 4b,
(Fb) is the voltage of the flywheel diode Ds2 of the inverter secondary circuit 4b, and (ga) is the inverter secondary circuit 4
The current of the DC reactor L of a, (gb) is the inverter 2
The current of the DC reactor L of the next circuit 4b, (h) is the load current.

4 (aa) and (ab), (ba) and (bb), (ca) and (cb), (da) and (db),
Although (ea) and (eb), (fa) and (fb), and (ga) and (gb) have the same waveform, their phases are shifted from each other by a half cycle. As for the load voltage, (ga) passes through the DC reactor L of the inverter secondary circuit 4a,
b) appears through the DC reactor L of the inverter secondary circuit 4b. Therefore, the maximum value of the load voltage is V1max, and the average value of the load voltage is 2 · V1max · (Ton / (Ton
+ Toff)). Since the average value of the load voltage is the required value for welding and may be the same as that of the circuit of FIG. 1 described above, Vmax = 2 · V1max. From this, the turns ratio (secondary / 1
The following will be 1/2 of the inverter transformer 3 in FIG. This indicates that the no-load voltage of the circuit of FIG. 3 is half that of the circuit of FIG. 1, indicating that the circuit of FIG. 3 is safer for the operator than the circuit of FIG. It is a thing.

Each inverter transformer 3a in FIG.
The maximum value of each current flowing through the secondary winding of 3b is also one-half of that of the device of FIG. 1, and considering the turns ratio, the maximum current flowing through the primary winding of each inverter is It is a quarter of two. Therefore, the total capacity of the two inverter transformers in FIG. 3 is one half of the capacity of the one inverter transformer in FIG. Further, in the case of FIG. 3, the product of the rated current and the rated voltage required for the transistors forming each inverter may be one quarter of that in the case of FIG.

FIG. 5 shows an example of the control circuit used in the embodiment shown in FIG. In the figure, 7 is a clock pulse generator, 8 is a control pulse generator, and 9a and 9b are transistor drive circuits. FIG. 6 is a diagram for explaining the operation of the control circuit of FIG. 5, where (a) is a clock pulse Vck, (b) is a control pulse Vca, (c) is a transistor drive pulse Ip1a, (c). d) is a transistor drive pulse Ip2a, (e) is a control pulse Vcb, (f) is a transistor drive pulse Ip1b, and (g) is a transistor drive pulse Ip2b.

The clock pulse generator 7 generates a clock pulse Vck at a constant cycle and inputs it to the control pulse generator 8. The control pulse generator 8 divides the clock pulse to generate the transistor Tp of the inverter primary circuit 2a shown in FIG.
1, Tp2 and control pulse Vca corresponding to each maximum ON DUTY of the transistors Tp1 and Tp2 of the inverter primary circuit 2b
And Vcb are generated, and the transistor drive circuits 9a and 9b are generated.
To enter. The transistor driving circuits 9a and 9b output pulses Ip1a, Ip2a and Ip1b, Ip2b having a time width corresponding to another input, that is, the output control signal Vcnt, in synchronization with the rising edges of the control pulses Vca and Vcb, and output the inverter 1 shown in FIG. The transistors Tp1 and Tp2 of the next circuit 2a and the transistors Tp1 and Tp2 of the inverter primary circuit 2b are driven. The transistor drive pulses Ip1a and Ip1a
The circuits p2a, Ip1b and Ip2b are isolated from each other.

Here, a clock pulse Vck is input as a control pulse generator, and this input pulse is divided into two phases to generate two sets of pulse trains whose ON DUTY is less than 50% and whose phases are shifted by 180 degrees. Any circuit will do. For this purpose, for example, an integrated circuit for controlling a stepping motor (such as TD62803P manufactured by Toshiba Corporation) can be used. FIG. 7 shows an embodiment of the control pulse generator 8. In the figure, IC1 is an integrated circuit for controlling a stepping motor,
Rck1 and Rck2 are resistors, and SWP is a switch. In the figure, when the clock pulse Vck is input, control pulses Vca and Vcb whose phases are different from each other by 180 degrees are obtained.

FIG. 8 shows the control pulse Vca and the output control signal Vcn.
Transistor drive circuit 9a that receives t and as inputs and obtains transistor drive pulses Ip1a and Ip2a corresponding to the input signal
It is a connection diagram showing an example of. In the figure, OP1 to OP3 are operational amplifiers, E1 to E5 are control power supplies, and PC.
1 to PC3 are photocouplers, TR11 to TR14 are transistors, DR1 to DR3 are diodes, AN
D1 is an AND gate. In the figure, the operational amplifier OP1 and the resistors R1 and R2 form an inverting amplifier. Further, a resistor R3 and a control power source E1 are connected in series with the polarity shown in the figure to the negative input terminal of the operational amplifier OP1. The output of the operational amplifier OP1 is negative when the control pulse Vca is positive and positive when the control pulse Vca is zero. The output of the operational amplifier OP1 is input to the integrator composed of the operational amplifier OP2, the resistor R4 and the capacitor C1 and the reset circuit for the integrator composed of the resistor R5 and the photocoupler PC1.

FIG. 9 is a diagram for explaining the operation of the transistor drive circuit 9a shown in FIG.
(A) is the control pulse Vca, (b) is the output voltage of the operational amplifier OP1, (c) is the output voltage Vth of the operational amplifier OP2,
(D) is the output voltage of the operational amplifier OP3, (e) is AND
1 output voltage. In FIG. 8 and FIG. 9, when the control pulse Vca is zero, the output of the operational amplifier OP1 becomes positive, and a current flows through the resistor R5 and the light emitting diode of the photocoupler PC1. Therefore, the output transistor of the photocoupler PC1 is turned on to discharge the electric charge of the capacitor C1 and reset the integrating circuit. When the control pulse Vca is positive, the output of the operational amplifier OP1 becomes negative, and no current flows through the resistor R5 and the light emitting diode of the photocoupler PC1. Therefore, photo coupler PC
The output transistor of No. 1 is turned off and the integrating circuit performs the integrating operation. As a result, the triangular wave Vth shown in FIG. 8C is obtained as the output of the integrating circuit.

This triangular wave Vth is output control signal Vcnt, operational amplifier OP3, resistors R6 and R7 and diode D.
It is input to the comparator composed of R2 for comparison. This comparator outputs a plus when Vth <Vcnt and a minus when Vth> Vcnt. The output of this comparator is AND gate A
It is added to one input terminal of ND1. A control pulse Vca is applied to the other input terminal of the AND gate AND1. As a result, when the control pulse Vca is positive and the output of the comparator is positive, the AND gate AND
The output of 1 becomes positive and a current flows through the resistor R8 to the light emitting diodes of the photocouplers PC2 and PC3.
Photocouplers PC2 and P having these light emitting diodes
The output transistors of C3 are control power sources E2 and E, respectively.
3, resistors R9, R10, transistors TR11, TR12
The transistor drive pulses Ip1a and Ip2a are generated together with the control power supplies E4 and E5, the resistors R11 and R12, and the transistors TR13 and TR14.

The transistor drive circuit 9b of FIG. 5 is also the same as 9a, but since the input control pulse Vcb is out of phase with Vca by a half cycle, the transistor drive pulses Ip1a and Ip2a are 180 degrees out of phase. The shifted transistor drive pulses Ip1b and Ip2b are applied to the transistor drive circuit 9b.
Will be output.

FIG. 10 shows a second embodiment of the present invention. In the embodiment shown in FIG. 3, the flywheel diodes Ds1 and Ds2 and the DC reactor L of the inverter secondary circuits 4a and 4b are taken out to the outside, and one is provided. It is the one that is commonly used. As shown in FIG. 11, the operation is such that the waveforms (e) and (f) of the flywheel diode Ds2 and the waveform (g) of the DC reactor L are different from those of the diagram of FIG. 4 for explaining the operation of the embodiment of FIG. Since they are only different, detailed description will be omitted.

FIG. 12 shows a third embodiment of the present invention. The difference between the third embodiment and the first embodiment is the DC reactor. In the first embodiment, the direct current reactor L of the inverter secondary circuit 4a and the direct current reactor L of the inverter secondary circuit 4b, which are independent from each other on the inverter a side and the inverter b side, are used, whereas in the third embodiment, they are magnetically coupled to each other. A DC reactor Lsc is used, and the currents flowing through the windings of Lsc (1/2) on the side of the inverter a and Lsc (2/2) on the side of the inverter b are magnetically coupled so as to cancel out the magnetic flux.

FIG. 13 shows a diagram for explaining the operation of the embodiment shown in FIG. 12 and 13, a DC power supply 1, a torch 5 and a base material 6, an inverter primary circuit 2
a and 2b operate in the same manner as in FIG. 3 of the first embodiment. The operation waveforms are (aa) to (ca) and (a) in FIG.
b) to (cb). The operations (aa) to (ca) and the operations (ab) to (cb) in FIG. 13 are out of phase with each other by a half cycle. The operation waveforms of the rectifier diode Ds1 and flywheel diode Ds2 of the inverter secondary circuit 4a and the rectifier diode Ds1 and flywheel diode Ds2 of the inverter secondary circuit 4b are shown in FIG.
As shown in a), (ea) and (db), (eb).

Transistor T of inverter primary circuit 2a
When p1 and Tp2 are ON and the transistors Tp1 and Tp2 of the inverter primary circuit 2b are OFF, the inverter secondary circuit 4
The current of the rectifying diode Ds1 of a is the same as the current flowing through the coupled DC reactor Lsc (1/2). The current flowing through the coupled DC reactor Lsc (2/2), torch 5, arc, base metal 6, flywheel diode Ds2 of the inverter secondary circuit 4b, and coupled DC reactor Lsc (2/2) is the coupled DC reactor Lsc. It becomes the same as the current flowing in (1/2).
The transistors Tp1 and Tp2 of the inverter primary circuit 2a are O
In the FF, the transistor Tp1 of the inverter primary circuit 2b,
When Tp2 is ON, the current of the rectifying diode Ds1 of the inverter secondary circuit 4b is the same as the current flowing through the coupled DC reactor Lsc (2/2). At this time, the coupled DC reactor Lsc
(1/2), the torch 5, the arc, the base material 6, the flywheel diode Ds2 of the inverter secondary circuit 4a, and the current flowing through the circuit of the coupled DC reactor Lsc (1/2), the coupled DC reactor Lsc (2/2 ) Is the same as the current flowing through.

Transistor T of inverter primary circuit 2a
When p1 and Tp2 are OFF and the transistors Tp1 and Tp2 of the inverter primary circuit 2b are also OFF, the coupled DC reactor Lsc (1/2), torch 5, arc, base metal 6, inverter 2
The current flowing through the flywheel diode Ds2 of the next circuit 4a and the circuit of the coupled DC reactor Lsc (1/2), the coupled DC reactor Lsc (2/2), the torch 5, the arc, the base material 6, and the inverter secondary circuit 4b. Flywheel diode Ds
2. The current flowing through the circuit of the coupled DC reactor Lsc (1/2) becomes the same. The voltage waveforms of the flywheel diode Ds2 of the inverter secondary circuit 4a and DRS2 of the inverter secondary circuit 4b are shown in (fa) and (fb). The voltage of the flywheel diode Ds2 of the inverter secondary circuit 4a becomes V3max when the transistors Tp1 and Tp2 of the inverter primary circuit 2a are ON. The voltage of the flywheel diode Ds2 of the inverter secondary circuit 4b is the inverter primary circuit 2
When the transistors Tp1 and Tp2 of b are ON, V3max is reached.

Therefore, the maximum value of the load voltage is V3max, and the average value of the load voltage is 2 · V3max · (Ton / (Ton /
+ Toff)). Since the average value of this load voltage may be the same as that of the circuit of FIG. 1 described above, Vmax = 2 · V
It will be 3max. This shows that the no-load voltage of the circuit of FIG. 12 is also half that of the circuit of FIG. 1, like the circuit of FIG. 3, and the circuit of FIG. 12 is safer than the circuit of FIG. It means that.

[0037]

Other Embodiments The above-mentioned Embodiments 1 to 3 are
Although the description has been given for the two-stone forward converter, the same thing can be said for the one-stone forward converter. Also,
The inputs of each inverter primary circuit may be connected to different DC power supplies.

The output of the DC power supply may be divided by capacitors, and the input of each inverter primary circuit may be connected to each capacitor. FIG. 14 is a connection diagram showing an example of such a case. In the figure, C1 and C2 are capacitors connected to the output section of the DC power supply 11, and divide the output of the DC power supply 11 into two. The terminal voltage of each capacitor is supplied to each inverter primary circuit 2a, 2b. Other parts of the figure are similar to those of the embodiment shown in FIG.

Further, in addition to the embodiment shown in FIG. 14, FIG. 15 shows that the output voltage of the DC power supply is set to 2 by using two capacitors.
Divide and connect the input terminals of each inverter primary circuit in parallel and connect them directly to the output terminals of the DC power supply.
Switch S for selecting whether to connect the input terminal of the next circuit to each capacitor that divides the output voltage of the DC power supply into two
It has a WP. In the case of the figure, for example, three-phase 20
A common device can be used by selecting the change-over switch Swp for two kinds of power supply voltages whose input voltages have a one-to-two relationship with each other such as 0V or 400V.

Although two sets of inverter circuits have been described above, three or more sets of inverter circuits may be used to connect the secondary side in parallel. In this case, the control pulse generators have phases that are one-third of each other (12
It suffices to prepare three sets of transistor drive circuits that obtain three control pulses shifted by 0 degrees) and generate a pulse having a time width corresponding to the output control signal in synchronization with the control pulses.

[0041]

As described above, according to the present invention, the no-load voltage is reduced to about half as compared with the conventional inverter-controlled arc welding machine using the forward converter.
The safety can be improved and the total capacity of the inverter transformer can be reduced to about half.

According to the second aspect of the present invention, it is possible to provide an inverter-controlled arc welding machine using a forward converter that can handle a high input voltage with a transistor having a lower voltage rating.

The third aspect of the present invention is, for example, 200 V
It is possible to provide an inverter-controlled arc welder of a forward converter that can be commonly used for two types of input voltages that are in a one-to-two relationship such as 00V.

In the inventions of claims 4 and 5, the flywheel diode of the inverter secondary circuit and the DC reactor or the DC reactor can be integrated into one part, so that the number of parts can be reduced.

[Brief description of drawings]

FIG. 1 is a connection diagram showing a circuit of a conventional two-stone forward converter,

2 is a diagram showing operating waveforms of the forward converter of FIG. 1,

FIG. 3 is a connection diagram showing a first embodiment of the present invention,

FIG. 4 is a diagram for explaining the operation of the apparatus of the embodiment of FIG.

5 is a connection diagram showing an example of a control circuit used in the embodiment of FIG.

6 is a diagram for explaining the operation of the control circuit of FIG.

7 is a connection diagram showing an example of a control pulse generator 8 used in the control circuit of FIG.

8 is a connection diagram showing an example of a transistor drive circuit 9a used in the control circuit of FIG.

9 is a diagram for explaining the operation of the transistor drive circuit 9a in FIG.

FIG. 10 is a connection diagram showing a second embodiment of the present invention,

11 is a diagram for explaining the operation of the apparatus of the embodiment of FIG.

FIG. 12 is a connection diagram showing a third embodiment of the present invention,

FIG. 13 is a diagram for explaining the operation of the apparatus of the embodiment of FIG.

FIG. 14 is a connection diagram showing a fourth embodiment of the present invention,

FIG. 15 is a connection diagram showing a fifth embodiment of the present invention.

[Explanation of symbols]

 1,11 DC power supply D1 to D6 Rectifier C, C1, C2 smoothing capacitor Tm1 to Tm3 AC input terminal 2, 2a, 2b inverter primary circuit Tp1, Tp2 transistor TR11 or TR14 transistor Dp1, Dp2, Dp3, Dp4 diode 3 inverter Transformer 3a, 3b Inverter transformer 4 Inverter secondary circuit 4a, 4b Inverter secondary circuit 41a, 41b Inverter secondary circuit 42a, 42b Inverter secondary circuit Ds1 Rectifying diode Ds2 Flywheel diode L DC reactor Lsc Coupling DC reactor 5 Torch 6 Mother Material 7 Clock pulse generator 8 Control pulse generator IC1 Stepping motor control IC Rck1, Rck2 Resistor Swp switch 9a Transistor drive circuit 9b Transistor drive circuit OP1 to OP3 Operational amplification It E5 control power PC1 to no E control power E1 to PC3 photocoupler AND1 AND gates DR1 DR3 diodes R1 to R12 resistor

Claims (5)

[Claims] 1. n sets of primary input terminals connected in parallel and connected to one DC power supply, or power supplied from different DC power supplies and output terminals connected in parallel (where n is an integer of 2 or more). Inverter-controlled welding power source, comprising: the forward converter and the control circuit for operating the above-described forward converters in each set by sequentially shifting the phase by 1 / n cycle. 2. N capacitors (where n is an integer of 2 or more) for dividing the output voltage of the DC power supply, and n sets of forward converters in which the terminal voltage of each capacitor is input and the output terminals are connected in parallel. An inverter-controlled welding power source, comprising: a control circuit for operating the forward converter by sequentially shifting the phase by 1 / n cycle. 3. Two capacitors that divide the output voltage of the DC power supply into two equal parts, two sets of forward converters whose output terminals are connected in parallel, and two sets of the forward converters that correspond to the output voltage of the DC power supply. And a switching circuit for selecting whether to directly connect the input terminals of the capacitors to the output terminals of the DC power supply or to connect the input terminals of the forward converters of each set to the respective terminals of the capacitors, and the forward converters of each set. Inverter-controlled welding power source equipped with a control circuit that operates by shifting the phases of each other by a half cycle. 4. The inverter-controlled welding power source according to claim 1, wherein the output terminals of the forward converters of each set are connected in parallel, and then one DC reactor is connected. 5. The inverter-controlled welding power source according to claim 1 or 2, wherein there are two sets of said forward converters, a DC reactor is provided at an output terminal of each set, and a winding of each DC reactor is a common iron core. An inverter-controlled welding power source that is composed of windings that are wound and have polarities in which the magnetic fields created by the currents flowing in each winding cancel each other out.