JPH0746708A - Controller for induction motor for electric car - Google Patents

Abstract

PURPOSE:To reduce the transient loss by providing a steady loss minimum flux calculating part, a target flux calculating part, and a target torque calculating part, and changing the transfer functions of the target flux calculating part and the target torque calculating part according to the residual capacity of a battery. CONSTITUTION:A torque command value Te' is inputted to calculating parts 11 and 15, and a voltage Vb and a current ib are inputted to an operation mode detecting part 14, and a steady loss minimum flux phir', a target flux response time constant tauphi, a target torque response time constant tauT, and a target torque Tm are found. A target flux phir and a single differentiated value dphir/dt are found and an exciting current command value iphi' is calculated 13, by calculation 12 of the minimum flux phir' and the time constant tauphi by the use of a transfer function determined by the residual capacity of a battery. A torque current command value itau' is calculated 16 by the magnetic flux phir and the target torque Tm, and outputted. A slip frequency (wse) is calculated 17 from the magnetic flux phir and a torque current command value itau', and a power frequency (w) is formed by adding the number of rotations (wre) of a motor to it, and is integrated, and the phase angle theta of the current is outputted. Consequently, it becomes possible to enhance responsibility and lower the transient loss.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an induction motor used in an electric vehicle, and more particularly to a high efficiency drive control technique for the same.

[0002]

2. Description of the Related Art As a conventional induction motor control method,
For example, there is one described in JP-A-2-23085. This control method, only newly provided a core-loss resistance of R _M to d-axis component of the induction motor model according to the conventional d-q axis coordinate, describes the loss of the induction motor (copper loss, iron loss), these The induction motor is driven so that the loss of is minimized in the steady state. Specifically, vector control generally used as a torque control method for an induction motor is used, and control is performed at a slip frequency derived from the condition of minimum steady loss. Since the iron loss resistance R _M changes according to the motor rotation speed, the minimum loss slip frequency can be obtained by reading the table map of the iron loss resistance R _M from the motor rotation speed and solving the minimum loss conditional expression.

[0003]

However, in such a conventional induction motor control method, the minimum loss condition is derived from the steady-state characteristics of the motor and the transient characteristics are not taken into consideration. Loss is not minimal. Then, if the slip frequency is made a function of only the motor rotation speed as in the conventional method, it is necessary to change the rotor magnetic flux stepwise when the torque command value changes stepwise. Needs to flow a large current transiently. Therefore, when the minimum loss condition is set without considering the transient characteristics as in the prior art, there is a problem that the transient loss of the motor and the current capacity of the motor drive device increase. On the contrary, if the current capacity is set not to be increased, the torque response must be delayed. In order to solve the above problem, the present inventors have added a steady loss minimum magnetic flux calculation unit, a target magnetic flux calculation unit, and a target torque calculation unit to a general vector control calculation unit to obtain torque response and magnetic flux response. By having a control system configuration that can independently change and
Loss of slip frequency in steady state Minimum slip frequency ω
We have developed (unpublished) an induction motor controller configured to reduce transient loss by using _se-opt and transiently controlling the magnetic flux response to the optimum value according to the torque response. However, in the above-mentioned prior inventions of the present inventors, the target torque response calculated by the target torque calculation unit is a value that is not affected by the remaining capacity of the battery (the remaining amount of the charged electric power), and therefore even when fully charged. The torque response and transient loss are the same even when the remaining capacity is low. Therefore, the torque responsiveness to the accelerator operation performed by the driver (the torque command value is set by this) does not change at the time of full charge or at the time of low remaining capacity, so that the driver may experience a decrease in the remaining capacity. There is a fear that it will not be possible to operate without consideration for the remaining capacity, and the current will flow so as to maintain the same torque response as in the normal state even when the remaining capacity is low, so the remaining capacity will drop rapidly and the battery will rise. There is a problem that there is a fear that causes.

The present invention has been made in order to solve the problem of the prior art that the transient loss cannot be reduced, and further to solve the problems of the prior inventions of the present inventors as described above. An induction motor control device for an electric vehicle capable of reducing not only steady loss but also transient loss, reducing transient loss at a low remaining capacity more than usual, and allowing a driver to experience a reduction in remaining capacity. The purpose is to provide.

[0005]

In order to achieve the above object, the present invention is constructed as described in the claims. That is, in the invention described in claim 1, in addition to the same vector control operation unit and motor drive unit as in the conventional case, a remaining capacity detection unit that detects the remaining capacity of the battery that supplies the drive power of the induction motor, and The minimum loss steady-state magnetic flux calculation unit that calculates the rotor magnetic flux that minimizes the steady-state loss of the induction motor at the specified torque command value, and the above-mentioned minimum steady-state loss magnetic flux are input, and transmission that has a low-pass characteristic that is determined according to the remaining capacity. A target magnetic flux calculation unit that calculates a target magnetic flux and a first-order differential value of the target magnetic flux based on a function,
And a target torque calculation unit that calculates a target torque of the induction motor based on a transfer function determined from the torque command value according to the remaining capacity. The target magnetic flux calculation unit is
For example, as described in claim 2, the time constant of the transfer function is a function of the remaining capacity of the battery. In addition, the target torque calculation unit, for example, as described in claim 3,
The time constant of the transfer function is a function of the remaining capacity of the battery. The steady loss minimum magnetic flux calculation unit, the target magnetic flux calculation unit, and the target torque calculation unit are, for example, the steady loss minimum magnetic flux calculation unit 11, the target magnetic flux calculation unit 12, and the target torque calculation unit 15 in the embodiment of FIG. 1 described later. Respectively correspond to. In addition, the above-mentioned remaining capacity detection unit is, for example, as shown in FIG.
It corresponds to a part of the operation mode detection unit 14 or the remaining capacity detection unit 101 of FIG. The current command value corresponds to, for example, the exciting current command value i _φ ′, the torque current command value i _T ′, and the phase angle θ of the current in the embodiment shown in FIG. 1 or FIG. 2 described later.

Further, the remaining capacity of the battery can be obtained, for example, by subtracting the integrated value of the emitted power from the value at the time of full charge.

[0007]

As described above, in the present invention, the steady loss minimum magnetic flux calculating section, the target magnetic flux calculating section and the target torque calculating section are added to a general vector control calculating section to obtain the torque response and the magnetic flux response. By making the control system configuration that can be independently varied, the slip frequency is constantly set to the minimum loss slip frequency ω _se-opt, and transiently the magnetic flux response is controlled to the optimum value according to the torque response. Therefore, a transient capacity is configured to be reduced, and a remaining capacity detecting section for detecting the remaining capacity of the battery is further provided. According to the detected remaining capacity, the transfer function in the target magnetic flux calculating section and the target torque calculating section is determined. Is changed to reduce the torque response when the remaining capacity is reduced. By configuring as described above, transient loss can be reduced without impairing responsiveness,
In addition, in steady state, it can be driven with the same minimum loss as the conventional one. Further, when the remaining capacity is low, the transient loss can be reduced as compared with the normal state, and the driver can feel the remaining capacity decrease.

[0008]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention will be described below with reference to the drawings.
1 and 2 are diagrams showing an embodiment of the present invention. FIG. 1 is a block diagram showing the details of the high-efficiency drive control calculation unit 1 in FIG. 2, and FIG. 2 is a block diagram showing the configuration of the entire system. . First, in FIG. 2, 1 is a high-efficiency drive control arithmetic unit (described later in detail), for example, the motor rotation speed N detected by the rotational speed sensor 5 and the torque command value T _e 'corresponding to the operation amount of an accelerator pedal (Rpm) is input to calculate and output the exciting current command value i _φ ′, the torque current command value i _T ′, and the phase angle θ of the current. Reference numeral 2 is a coordinate conversion unit that calculates the excitation current command value i _φ ′, the torque current command value i _T ′, and the phase angle θ of the current calculated in the coordinate system that rotates at the motor power supply frequency into a three-phase alternating current. Current command value _iu ', _iv ',
Convert to i _w '. 3 is current control PWM (pulse width modulation)
It is an inverter and causes the three-phase alternating currents i _u , _iv , and i _w flowing in the induction motor 4 to follow the respective command values. Reference numeral 5 denotes a rotation speed sensor that detects the rotation speed of the induction motor 4, and 6
Is a DC power source (power source for driving an induction motor) that supplies power to the current control PWM inverter 3, that is, a battery.

In FIG. 1, 11 is a steady loss minimum magnetic flux calculating section, 12 is a target magnetic flux calculating section, 13 is an exciting current calculating section, 14 is an operation mode detecting section (details are shown in FIG. 3), and 15 is a target torque. An arithmetic unit, 16 is a torque current arithmetic unit, 17 is a slip frequency arithmetic unit, 18 is a motor rotational speed arithmetic unit, and 19 is an integral arithmetic unit. The exciting current calculation unit 13, the torque current calculation unit 16, the slip frequency calculation unit 17, the motor rotation speed calculation unit 18, and the integration calculation unit 19 are units that perform general vector control calculations.

Next, the operation will be described. First, an outline of the operation of each arithmetic unit will be described, and subsequently, a characteristic portion of this embodiment will be described in detail. In Figure 1, the steady loss minimum magnetic flux calculating unit 11 'enter the, the torque command value T _e' torque command value T _e calculating a magnetic flux phi _r 'to loss in the steady state (copper loss) at the minimum And output. In addition, the operation mode detection unit 14 detects the remaining capacity of the battery 6 (remaining amount of electric power being charged), and calculates a target magnetic flux response time constant τ _φ and a target torque response time constant τ _{T corresponding} thereto. . Further, the target magnetic flux calculation unit 12 determines that the above steady-state loss minimum magnetic flux φ
_r'is input and corresponds to the above minimum steady loss magnetic flux φ _r 'in the steady state, and the target magnetic flux φ _r that makes the magnetic flux response an optimum value according to the torque response in the transient state and its first derivative The value d / dt φ _r is calculated and output. In the calculation of the target magnetic flux φ _r, the calculation is performed based on the transfer function having the low-pass characteristic which is determined according to the remaining capacity of the battery. That is, calculation is performed using the target magnetic flux response time constant τ _φ as the time constant of the transfer function. Further, the target torque calculation unit 15 inputs the torque command value T _e ′ and sets the target torque T _m based on a predetermined transfer function (set according to the required responsiveness, the allowable current capacity, etc.). Calculate The predetermined transfer function has a characteristic that changes according to the remaining capacity of the battery. That is, calculation is performed using the target torque response time constant τ _T as the time constant of the transfer function. Further, the excitation current calculation unit 13 and the torque current calculation unit 1
6. The slip frequency calculator 17 is a part for performing general vector control calculation. First, the exciting current calculation unit 13 calculates and outputs the exciting current command value i _φ ′ based on the target magnetic flux φ _r and the first-order differential value d / dt φ _r given from the target magnetic flux calculation unit 12. . Further, the torque current calculation unit 16 inputs the target torque T _m of the target torque calculation unit 15 and the target magnetic flux φ _r of the target magnetic flux calculation unit 12 and calculates and outputs the torque current command value i _T ′. Further, the slip frequency calculation unit 17 inputs the target magnetic flux φ _r of the target magnetic flux calculation unit 12 and the torque current command value i _T ′ of the torque current calculation unit 16,
Calculates and outputs the slip frequency ω _se . Further, the motor rotation speed calculation unit 18 adds the number P of pole pairs peculiar to the induction motor to the motor rotation speed N given from the rotation speed sensor 5 of FIG.
Is multiplied by to calculate the motor rotation speed (electrical angle) ω _re . That is, ω _re = (π / 30) N × P. The sum of the motor speed ω _re and the slip frequency ω _se is the power supply frequency ω. That is, ω = ω _se + ω _re . Then, the integration calculator 19 outputs a value obtained by integrating the power supply frequency ω as the phase angle θ of the current. The exciting current command value i _φ ′, the torque current command value i _T ′, and the current phase angle θ are sent to the coordinate conversion unit 2 in FIG. 2 as current command values.

Next, the details of each arithmetic unit will be described.
First, since the excitation current calculation unit 13, the torque current calculation unit 16, and the slip frequency calculation unit 17 are portions for performing general vector control calculation, detailed description thereof will be omitted. By giving ω _se by the following formula (1), the output torque T _e of the induction motor is introduced into the form of the following formula (2).

[0012]

[Equation 1]

Where ω: power frequency, ω _re : motor speed (electrical angle), M: mutual inductance, L _r : rotor self-inductance, R _r : rotor resistance, i _T : torque current, φ _r : Rotor flux

[0014]

[Equation 2]

However, P: number of pole pairs Further, at this time, the relationship between the rotor magnetic flux φ _r and the exciting current i _φ is expressed by the following equation (3).

[0016]

[Equation 3]

However, S: Laplace operator Therefore, the calculation formula of the excitation current command value i _φ 'performed by the excitation current calculation unit 13 is as shown in the following (Formula 4) from the above (Formula 3). .

[0018]

[Equation 4]

Further, the calculation formula of the torque current command value i _T 'performed by the torque current calculation unit 16 is as shown in the following (Formula 5) from the above (Formula 2).

[0020]

[Equation 5]

However, T _e ′: Torque command value Note that the slip frequency ω in the slip frequency calculator 17 is
The arithmetic expression of _se is as shown in the above equation (1).
The calculations performed by the motor rotation speed calculation unit 18 and the integration calculation unit 19 are as described above.

Next, the operation mode detector 14, the minimum steady loss magnetic flux calculator 11, the target magnetic flux calculator 12, and the target torque calculator 15, which characterize this embodiment, will be described. First, the operation mode detection unit 14 has a configuration as shown in FIG. 3, for example. In FIG. 3, 101 is a remaining capacity detection unit, and 102 is a time constant calculation unit. The remaining capacity calculation unit 101 determines the terminal voltage V of the battery 6 for driving the induction motor.
_b and the current i _b flowing to the induction motor are detected, and the remaining capacity C _b ′ is calculated by the following equation, for example. C _b ′ = C _b0 −∫ (V _b · i _b ) dt where C _b0 is the remaining capacity when fully charged. That is,
In this example, the remaining capacity C _b 'is obtained as a value obtained by subtracting the total amount of electric power discharged from the battery from the fully charged value C _b0 . When the vehicle is being braked, the driving induction motor is driven in reverse from the wheels to operate as a generator, and the generated electric power is charged in the battery. During so-called regenerative braking, the direction of the current is reversed. In the equation (1), i _b becomes negative, and therefore the remaining capacity C _b 'changes so as to increase. Further, when a lead storage battery is used as the battery, the remaining capacity can be detected from the terminal voltage of the battery, the internal resistance of the battery, the specific gravity of the electrolytic solution, and the like. That is, the electrolytic solution resistance of the lead storage battery increases as H ₂ SO ₄ in the solution decreases with discharge. The internal resistance of the electrode also rises because the discharge substance PbSO ₄ having a large resistance increases with the discharge. Therefore, the terminal voltage of the lead acid battery decreases with discharge, and the internal resistance increases. Further, the specific gravity of the electrolytic solution decreases almost linearly with the discharge. Therefore, the remaining capacity can be detected by measuring the terminal voltage of the lead storage battery, the internal resistance, or the specific gravity of the electrolytic solution. Further, the time constant calculation unit 102 calculates the target magnetic flux response time constant τ _φ and the target torque response time constant τ _T according to the remaining capacity C _b ′. FIG. 4 shows a target magnetic flux response time constant τ _φ and a target torque response time constant τ with respect to the remaining capacity C _b ′ of the battery.
It is a characteristic view of _T. As shown in the figure, the target magnetic flux response time constant τ _φ and the target torque response time constant τ _T are characteristics in which the remaining capacity increases as the remaining capacity decreases in the range where the remaining capacity is smaller than a predetermined amount. In the range larger than the quantitative value, it becomes constant regardless of the remaining capacity. Therefore, below the predetermined value, the target magnetic flux response and the target torque response become slower as the remaining capacity becomes smaller.

Next, the portions of the steady loss minimum magnetic flux calculation unit 11, the target magnetic flux calculation unit 12, and the target torque calculation unit 15 will be described. In normal vector control, the rotor magnetic flux φ _r is kept constant (excitation current i _φ is kept constant), and only the torque current i _T is changed, whereby the torque current i _{T of the} output torque is changed.
To obtain linearity and quick response to. However, such a constant vector control of the rotor magnetic flux φ _r supplies a constant exciting current i _φ regardless of the load, and therefore the efficiency generally deteriorates at a light load. Therefore, the copper loss is considered as the loss of the induction motor, and the condition for minimizing this is obtained.
First, a γ-δ coordinate model as shown in FIG. 5, that is, a γ-δ coordinate model rotating at a power supply frequency well known as a model of an induction motor will be used. In FIG. 5, when the vector control is established, it is known that the following equation (6) is established between the current component of each axis and the rotor magnetic flux φ _r . However, each current component i _γs ,
In i _δs , i _γr , and i _δr , subscripts γ and δ represent respective axial components, r represents a rotor, and s represents a stator.

[0024]

[Equation 6]

On the other hand, the copper loss L _c of the induction motor is as shown in the following formula (7) from FIG. 5 and the above formula (6).

[0026]

[Equation 7]

[0027] However, R _s: stator resistance, K _1, K _2: In constant above (7) determined by the motor type, because the differential term d / dt φ _r of phi _r given the steady state becomes 0 The slip frequency ω _se-opt that minimizes the copper loss L _c can be calculated by the following equation (8) from the condition of dL _c / dω _se = 0.

[0028]

[Equation 8]

Therefore, the slip frequency ω _se is given by ( _Equation 8)
If the value of the formula is maintained, it is possible to drive the copper loss to the minimum.
Specifically, in order to keep the slip frequency at the minimum loss slip frequency ω _se-opt from the equation (1), the torque current i
It will be understood that the relationship between _T and the magnetic flux φ _r may be set as shown in the following equation (9).

[0030]

[Equation 9]

Next, by substituting the equation (9) into the equation (2) and erasing i _T , the torque command value T _e 'and the magnetic flux φ _r.
The relationship is expressed by the following equation (10).

[0032]

[Equation 10]

Therefore, when the torque command value T _e 'is input, the steady loss minimum magnetic flux φ _r ' is derived by the formula (10), and the torque current command value i _T 'is calculated by the formula (9), the exciting current. The command value i _φ 'is calculated by the equation (4) and the slip frequency ω _se is calculated by the equation (1), so that the copper loss can be constantly minimized. At this time, the slip frequency ω _se coincides with the minimum loss slip frequency ω _se-opt , and the output torque T _e follows the torque command value T _e ′. However, when the torque command value T _e 'changes stepwise, the steady loss minimum magnetic flux φ _r ' also becomes stepwise from the equation (10). Then, as shown in equation (4), since the calculation of the excitation current command value i _phi 'contains d.phi _r / dt, when the step-like torque command value changes occur, the exciting current command value i _phi 'Is a transiently large value, which increases the transient loss. The above phenomenon is expressed by the equation (7),
It can also be understood from the fact that L _c is a function of dφ _r / dt.
Further, in the present embodiment, only copper loss is considered as the loss, but the same problem occurs when copper loss and iron loss are taken into consideration as described in the above-mentioned conventional example (JP-A-2-23085). Occurs. Therefore, when the transient loss is also taken into consideration, the slip frequency is calculated as the minimum slip frequency ω
_{Keeping se-opt} is not a good idea in terms of efficiency.
Further, when the upper limit value of the current is determined from the current capacity of the semiconductor switching element used for the current control PWM inverter 3 of FIG. 2, the torque response must be delayed during the transient when the current becomes large. Therefore, in the present embodiment, the target torque calculation unit 15, which determines the torque response, the minimum steady loss magnetic flux calculation unit 11, and the target magnetic flux calculation unit 12 are provided as a general vector control calculation unit 13,
By adding to 15 and 16, the control system configuration that can independently change the torque response and the magnetic flux response makes the slip frequency steady and the minimum loss slip frequency ω _se-opt, and transiently the magnetic flux response. Is controlled to be an optimum value according to the torque response to reduce the transient loss. First, the steady loss minimum magnetic flux calculation unit 11
The calculation content of is the equation (10), and the rotor magnetic flux that minimizes the loss steadily, that is, the minimum steady loss magnetic flux φ _r ′ is calculated.

The target magnetic flux calculating unit 12 for calculating the target magnetic flux φ _r is a filter having a steady gain of 1. In this embodiment, the target magnetic flux calculating unit 12 is a first-order low-pass filter as shown in the following (Equation 11).

[0035]

[Equation 11]

However, τ _φ : time constant of target magnetic flux, S: Laplace operator Note that the time constant τ _φ of the above target magnetic flux is a variable value according to the remaining capacity as shown in FIG. Details later). Further, the target torque calculation unit 15 that calculates the target torque T _m has a transfer function as shown in the following (Equation 12) in this embodiment. This transfer function is set according to the required responsiveness and current capacity.

[0037]

[Equation 12]

Τ _T : Time constant of target torque Note that the time constant τ _T of the target torque is a variable value according to the remaining capacity as shown in FIG. 4 (details will be described later).

FIG. 6 shows the peak value of the loss of the induction motor with respect to the time constant τ _φ of the target magnetic flux when a stepwise input is applied as the torque command value T _e 'in the control system shown in FIG. It is a characteristic view which shows the calculated value of the energy loss within a certain time. It can be seen from FIG. 6 that each loss has the characteristic of having the minimum value, and each loss has the minimum value when the time constant τ _φ is a certain value. Therefore, in the control system shown in FIG. 1, when the torque responsiveness is given by the equation (12), the magnetic flux responsiveness is determined using the time constant τ _φ of the target magnetic flux that minimizes the transient loss obtained from FIG. By doing so, it is possible to drive the motor with less loss both in the transient state and in the steady state. That is, the value of τ _φ that minimizes the loss in FIG. 6 may be used in the equation (11).

FIG. 7 and FIG. 8 are characteristic diagrams showing the results of speed control simulation. FIG. 7 shows the results when the method of keeping the slip frequency at the loss minimum slip frequency ω _se-opt in both transient and steady state is used. 8 shows the characteristics when the method according to the present embodiment is used. The torque responsiveness was the same for both. 7 and 8, the characteristics (response) of the torque T and the rotation speed N are the same, but the copper loss L
The characteristic of _c is clearly reduced in the present embodiment in the transient state, and in the steady state, it is the slip frequency drive that minimizes the copper loss as in the conventional case. Therefore, in this embodiment, it is possible to reduce the loss during the transition while maintaining the same response characteristic.

Further, in the present embodiment, as described above, the value of the target torque time constant τ _T changes in accordance with the remaining capacity in the calculation in the target torque calculating section 15, as shown in FIG. Further, the characteristic of FIG. 6 is a characteristic corresponding to a predetermined time constant τ _T , and if the time constant τ _T changes, the value of the time constant τ _φ of the target magnetic flux that minimizes the loss also changes. Therefore, the time constant τ _φ of the target magnetic flux in the calculation of the target magnetic flux calculation unit 12 is also changed as shown in FIG. The time constant τ _φ of the target magnetic flux in this case is a value that minimizes the loss in the characteristic of FIG. 6 corresponding to the time constant τ _T of the target torque at that time. As described above, by changing the time constant τ _T of the target torque according to the remaining capacity of the battery, the target torque response is delayed when the remaining capacity decreases, that is, the torque responsiveness to the driver's accelerator operation. It is possible to make the driver experience a reduction in the remaining capacity. Also, transient loss can be reduced by increasing the time constant τ _T of the target torque when the remaining capacity is low, but by changing the time constant τ _φ of the target magnetic flux with it, the transient loss can be reduced. It can be effectively reduced to reduce power consumption and extend the mileage. FIG. 9 is a characteristic diagram showing changes in the torque target value, the magnetic flux target value, and the loss when the time constant τ _T of the target torque and the time constant τ _φ of the target magnetic flux are changed. In FIG. 9, curves and indicate the characteristics at the following values, respectively. τ _T = 0.087s τ _φ = 0.14s Peak Loss = 100% τ _T = 0.100s τ _φ = 0.15s Peak Loss = 86% τ _T = 0.118s τ _φ = 0.16s Peak Loss = 74 % the characteristic of FIG. 9, at time t _0, showing a characteristic when the torque command value rises as shown.
As can be seen from FIG. 9, the peak loss decreases as the time constant τ _T of the target torque and the time constant τ _{φ of the} target magnetic flux increase. As described above, when the remaining capacity of the battery decreases, the time constants τ _T and τ _φ are increased, the torque response is moderated, and the peak current during the transition is decreased to reduce the remaining capacity of the battery. By reducing the peak current during the transient, it is possible to prevent the battery voltage from instantaneously dropping during the transient and prevent the inverter from being adversely affected.
Therefore, depending on the remaining capacity of the battery, the time constant τ _T ,
With the arrangements to vary the tau _phi, slows the target torque response when the remaining capacity is decreased, i.e. experience by reducing the Torukure responsiveness to the accelerator operation of the driver, the remaining capacity decreases to the driver It can be done. When the remaining capacity is low, transient loss can be effectively reduced, power consumption can be reduced, and the mileage can be extended. By reducing the peak current during the transient, the decrease in the remaining capacity of the battery is reduced. By suppressing the peak current during the transition, it is possible to prevent the battery voltage from instantaneously decreasing during the transition and prevent the operation of the inverter from being adversely affected.

In this embodiment, the magnetic flux response is expressed by (Equation 1)
Although the equation (1) and the torque response are given by the equation (12), they are not necessarily limited to this transfer function. For example, a secondary vibration system [G (S) = ω _n ² /
S ² + 2ζω _n S + ω _n ² ], and similarly, the magnetic flux response may be given by a secondary vibration system. However, in order to obtain the first-order derivative value of the target magnetic flux without performing the differential operation,
It is necessary that the relative order between the numerator and the denominator be 1 or more. When the magnetic flux response is given by the secondary vibration system, the relationship between the damping ratio ζ and the natural frequency ω _n shown in Fig. 6 is calculated, and the magnetic flux is calculated using ζ and ω _n that minimize the transient loss. Good.

[0043]

As described above, according to the present invention, the means for calculating the steady loss minimum magnetic flux that minimizes the steady loss from the torque command value and the target torque of the induction motor from the torque command value are calculated. By providing a means and a means for causing the torque of the induction motor to follow the target torque from the above steady-state loss minimum magnetic flux and calculating the magnetic flux response that minimizes the transient loss, the slip frequency is always maintained as in the conventional case. Transient loss can be reduced if the same torque response as the method of maintaining the minimum loss slip frequency ω _se-opt is used. On the contrary, if the current capacity is the same, the torque responsiveness can be increased. Further, in the steady state, the motor drive control that minimizes the loss becomes possible as in the conventional method. Furthermore, by providing a remaining capacity detection unit that detects the remaining capacity of the battery and changing the transfer functions in the target magnetic flux calculation unit and the target torque calculation unit according to the detected remaining capacity, transient loss is reduced at low remaining capacity. It can be reduced compared to the normal state, and the driver can experience a decrease in the remaining capacity. Therefore, the travelable distance can be extended, and the driver can be automatically informed of the decrease in the remaining capacity and urged to travel to the nearest rechargeable place.
Further, by suppressing the peak current during the transition, it is possible to prevent the battery voltage from instantaneously dropping during the transition and prevent the operation of the inverter from being adversely affected.

[Brief description of drawings]

FIG. 1 is a block diagram showing details of a high-efficiency drive control calculation unit 1 in an embodiment of the present invention.

FIG. 2 is a block diagram showing the overall configuration of a system according to an embodiment of the present invention.

FIG. 3 is a block diagram of an embodiment of a driving mode detector 14.

FIG. 4 is a target magnetic flux response time constant τ with respect to the remaining capacity C _b '
A characteristic diagram of _φ and a target torque response time constant τ _T.

FIG. 5 is a diagram showing a γ-δ coordinate model of an induction motor.

FIG. 6 shows the peak value of the loss of the induction motor with respect to the time constant τ _φ of the target magnetic flux when a stepwise input is added as the torque command value T _e ′, and the calculated value of the energy loss within a certain time. FIG.

FIG. 7 is a characteristic diagram showing a speed control simulation result in a conventional example.

8 is a characteristic diagram showing a speed control simulation result in the embodiment of FIG.

FIG. 9 is a characteristic diagram showing changes in the target torque value, the desired flux value and the loss when the time constant τ _{T of the} target torque and the time constant τ _φ of the target magnetic flux are changed.

[Explanation of symbols]

1: High-efficiency drive control calculation unit 4: Induction motor 2: Coordinate conversion unit 5: Rotation speed sensor 3: Current control PWM inverter 6: DC power supply (battery) 11: Steady loss minimum magnetic flux calculation unit 16: Torque current calculation unit 12 : Target magnetic flux calculation unit 17: Slip frequency calculation unit 13: Excitation current calculation unit 18: Motor rotation speed calculation unit 14: Operation mode detection unit 19: Integral calculation unit 15: Target torque calculation unit 101: Remaining capacity detection unit 102: Hour Constant calculation part V _b : DC power supply voltage i _b : DC power supply current T _e ': Torque command value T _m : Target torque τ _φ : Target magnetic flux response time constant τ _T : Target torque response time constant φ _r ': Steady state Minimum loss magnetic flux φ _r : Target magnetic flux N: Motor rotation speed (rpm) ω: Power supply frequency ω _se : Slip frequency ω _re : Motor rotation speed (electrical angle) i _φ ': Excitation current command value i _T ': Torque current command Value θ: Phase angle of current _iu ', _iv ', _iw ': Three-phase AC current command value _iu , _iv , _iw : Three-phase AC current

 ─────────────────────────────────────────────────── ─── Continued Front Page (72) Inventor Yasuhiko Kitajima 2 Takaracho, Kanagawa-ku, Yokohama, Kanagawa Nissan Motor Co., Ltd.

Claims (3)

[Claims] 1. An induction motor controller for calculating a current command value according to a torque command value and a rotation speed of an induction motor, and driving the induction motor with a polyphase alternating current corresponding to the current command value. A remaining capacity detection unit that detects the remaining capacity of the battery that supplies the driving power, a steady loss minimum magnetic flux calculation unit that calculates the rotor magnetic flux that minimizes the steady loss of the induction motor at a given torque command value, and A steady-state minimum magnetic flux is input, and a target magnetic flux computing unit that computes a target magnetic flux and a first-order derivative value of the target magnetic flux based on a transfer function having a low-pass characteristic that is determined according to the remaining capacity, and the above-mentioned residual from the torque command value. A target torque calculation unit that calculates the target torque of the induction motor based on the transfer function that is determined according to the capacity, and the target magnet based on the circuit constant of the induction motor. And a vector control calculation unit that calculates the current command value according to the first-order differential value of the target magnetic flux, the target torque, and the rotation speed of the induction motor, and the current flowing through the induction motor follows the current command value. An electric motor induction motor control device for controlling an output torque of the induction motor to a value corresponding to the target torque. 2. The induction motor control device for an electric vehicle according to claim 1, wherein the target magnetic flux calculation unit is such that the time constant of the transfer function is a function of the remaining capacity. 3. The induction motor control device for an electric vehicle according to claim 1, wherein the target torque calculation unit is such that the time constant of the transfer function is a function of the remaining capacity.