JP2833422B2 - Induction motor control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an induction motor used in an electric vehicle or the like, and more particularly to a high-efficiency drive control technology thereof.

[0002]

2. Description of the Related Art Conventional induction motor control methods include:
For example, there is one described in JP-A-2-23085. This control method, only newly provided a core-loss resistance of R _M to d-axis component of the induction motor model according to the conventional d-q axis coordinate, describes the loss of the induction motor (copper loss, iron loss), these The induction motor is driven so that the loss of the induction motor is minimized in a steady state. Specifically, vector control, which is generally used as a torque control method for an induction motor, is used, and control is performed at a slip frequency derived from the condition of minimum steady-state loss. Since iron loss resistance R _M varies in accordance with the motor rotational speed, read a table map of the iron loss resistance R _M from the motor rotation speed, solving the loss minimum condition, the resulting loss minimum slip frequency.

[0003]

However, in such a conventional induction motor control method, the minimum loss condition is derived from the steady state characteristics of the motor and the transient characteristics are not taken into account. Loss is not minimal. If the slip frequency is made a function of only the motor rotation speed as in the conventional method, when the torque command value changes in a stepwise manner, it is necessary to change the rotor magnetic flux in a stepwise manner. Requires a large current to flow transiently. Therefore, when the minimum loss condition is set without considering the transient characteristics as in the related art, there is a problem that the transient loss of the motor increases and the current capacity of the motor driving device increases. Conversely, if the current capacity is set so as not to increase, a problem arises in that the torque response must be slowed down. In order to solve the above problem, the present inventors have added a steady-state loss minimum magnetic flux calculation unit, a target magnetic flux calculation unit, and a target torque calculation unit to a general vector control calculation unit, and have a torque response and a magnetic flux response. Can be independently varied, so that the slip frequency is constantly reduced to the minimum loss slip frequency ω _se-opt
An induction motor control device configured to reduce transient loss by transiently controlling the magnetic flux response to an optimal value corresponding to the torque response was developed (not disclosed). In this induction motor control device, the responsiveness of the target magnetic flux calculated by the target magnetic flux calculator is constant irrespective of the magnitude of the magnetic flux. As a loss of the induction motor, when considering copper loss only, loss L _c is as shown in the following equation (1) below.

[0004]

(Equation 1)

However, _Te : torque, ω _se : slip frequency, φ: magnetic flux K _{1 to} K ₄ : constant determined by the motor [Note that the following equation (8) describes the above equation (1) in more detail. Equation] As can be seen from Equation (1), the transient loss of the induction motor is affected by the magnetic flux φ in addition to dφ / dt. Therefore, since the transient loss changes depending on the magnitude of the magnetic flux φ when the magnetic flux φ changes, the transient loss can be further reduced by changing the magnitude of dφ / dt according to the magnitude of the magnetic flux φ. It becomes possible. However, in the prior invention by the present inventor, since the responsiveness of the target magnetic flux calculated by the target magnetic flux calculator is constant irrespective of the magnitude of the magnetic flux, the transient loss depends on the magnitude of the magnetic flux. Could not be minimized.

SUMMARY OF THE INVENTION The present invention has been made to solve the problems of the prior art as described above, and to further improve the inventor's prior invention. The present invention has not only a steady loss but also a transient loss over a wide operating range. An object is to provide an induction motor control device that can be reduced.

[0007]

Means for Solving the Problems In order to achieve the above object, the present invention is configured as described in the claims. That is, in addition to the vector control calculation unit and the motor drive unit similar to the conventional one, a steady-state loss minimum magnetic flux calculation unit that calculates a rotor magnetic flux that minimizes the steady-state loss of the induction motor at a given torque command value; A target magnetic flux calculation unit that inputs a minimum loss magnetic flux and calculates a first derivative of the target magnetic flux and the target magnetic flux based on a transfer function having a low-pass characteristic, and minimizes a transient loss according to the magnitude of the target magnetic flux. Thus, a variable magnetic flux response section for changing the transfer function in the target magnetic flux calculation section and a target torque calculation section for calculating a target torque of the induction motor based on a predetermined transfer characteristic from the torque command value are provided. In addition,
The above-described steady-state minimum loss magnetic flux calculation unit, target magnetic flux calculation unit, magnetic flux response variable unit, and target torque calculation unit are, for example, the steady-state minimum loss magnetic flux calculation unit 11, the target magnetic flux calculation unit 12, the magnetic flux response in the embodiment of FIG. It corresponds to the variable section 13 and the target torque calculation section 14, respectively. The current command value is
For example, they correspond to the excitation current command value i _φ ′, the torque current command value i _T ′, and the current phase angle θ in the embodiment of FIG. 1 or FIG.

[0008]

As described above, in the present invention, the steady-state loss minimum magnetic flux calculating section, the target magnetic flux calculating section, the variable magnetic flux response section, and the target torque calculating section are added to a general vector control calculating section to improve torque responsiveness. With a control system configuration that can vary the magnetic flux response independently, and by changing the transfer function of the target magnetic flux calculation unit so as to minimize the transient loss according to the magnitude of the magnetic flux, The slip frequency is set as the loss minimum slip frequency ω _se-opt, and the transient loss is transiently set by setting the value of the magnetic flux response so as to minimize the transient loss. With the above-described configuration, the transient loss can be reduced over a wide operating region without deteriorating the responsiveness, and the driving can be performed in a steady state with the same minimum loss as before.

[0009]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention will be described below with reference to the drawings.
1 and 2 are diagrams of an embodiment of the present invention. FIG. 1 is a block diagram showing details of the high-efficiency drive control operation unit 1 in FIG. 2, and FIG. 2 is a block diagram showing a configuration of the entire system. . First, the entire configuration shown in FIG. 2 will be described. FIG.
In, 1 is a high-efficiency drive control arithmetic unit (described later in detail), for example, a motor rotational speed detected by the rotational speed sensor 5 and the torque command value T _e 'corresponding to the operation amount of an accelerator pedal N (rpm) And calculates and outputs an exciting current command value i _φ ′, a torque current command value i _T ′, and a phase angle θ of the current. Reference numeral 2 denotes a coordinate conversion unit which converts the excitation current command value i _φ ′, torque current command value i _T ′, and current phase angle θ calculated in a coordinate system rotating at the power frequency of the motor into a three-phase AC. It is converted into current command values i _u ′, _iv ′, i _w ′. Reference numeral 3 denotes a current control PWM (pulse width modulation) inverter which causes the three-phase AC currents i _u , i _v , i _w flowing through the induction motor 4 to follow respective command values. Reference numeral 5 denotes a rotation speed sensor for detecting the rotation speed of the induction motor 4, and reference numeral 6 denotes a DC power supply (power supply for driving the induction motor) supplied to the current control PWM inverter 3.
It is.

Next, in FIG. 1, reference numeral 11 denotes a steady-state loss minimum magnetic flux calculator, 12 denotes a target magnetic flux calculator, 13 denotes a magnetic flux response variable unit, 14 denotes a target torque calculator, 15 denotes an exciting current calculator, and 16 denotes a torque. A current calculation unit, 17 is a slip frequency calculation unit, 18 is a motor rotation speed calculation unit, and 19 is an integration calculation unit. Note that the excitation current calculation unit 15 and the torque current calculation unit 1
6. Slip frequency calculator 17, motor rotation speed calculator 18
The integral operation unit 19 is a part that performs a general vector control operation.

Next, the operation will be described. First, a schematic operation of each arithmetic unit will be described, and then, a characteristic portion of the present embodiment will be described in detail. In Figure 1, the steady loss minimum magnetic flux calculating unit 11 'enter the, the torque command value T _e' torque command value T _e calculating a magnetic flux phi _r 'to loss in the steady state (copper loss) at the minimum And output. Further, the target magnetic flux calculation unit 12 inputs the above-mentioned steady loss minimum magnetic flux φ _r ′,
In a steady state corresponds to a steady loss minimum flux phi _r 'described above, the target magnetic flux phi _r to an optimum value according to torque response flux response in transient, first-order differential value d / dt φ _r that
Is calculated and output. Further, the magnetic flux response variable section 13
Transfer function at the target magnetic flux calculation unit 12 so as to minimize the transient loss in accordance with the size of the target flux phi _r, i.e. to vary the response of the magnetic flux in the transient. Also,
The target torque calculator 14 receives the torque command value T _e ′,
The target torque _Tm is calculated based on a predetermined transfer characteristic (set in accordance with required responsiveness, allowable current capacity, and the like). The excitation current calculation unit 15, the torque current calculation unit 16, and the slip frequency calculation unit 17 perform general vector control calculation. First, the excitation current calculation unit 15 calculates and outputs an excitation current command value i _φ ′ based on the target magnetic flux φ _r provided from the target magnetic flux calculation unit 12 and the first derivative d / dt φ _r. . The torque current calculation unit 16 inputs the target magnetic flux phi _r of the target torque T _m and the target magnetic flux calculating unit 12 of the target torque calculating section 14 calculates and outputs a torque current command value i _T '. Further, the slip frequency calculation unit 17 receives the target magnetic flux φ _r of the target magnetic flux calculation unit 12 and the torque current command value i _T ′ of the torque current calculation unit 16 and calculates and outputs the slip frequency ω _se . The motor rotation speed calculation unit 17 calculates the motor rotation speed (electric angle) ω _re by multiplying the motor rotation speed N given from the rotation speed sensor 5 in FIG. 2 by the pole pair number P unique to the induction motor. That is, ω _re = (π / 30) N × P. The sum of the motor rotation speed ω _re and the above-mentioned slip frequency ω _se is the power supply frequency ω. That is, ω = ω _se + ω _re
It is. Then, the integration operation unit 18 outputs a value obtained by integrating the power supply frequency ω as the phase angle θ of the current. The above-described excitation current command value i _φ ′, torque current command value i _T ′, and current phase angle θ are used as current command values in the coordinate conversion unit 2 of FIG.
Sent to

Next, the details of each operation unit will be described.
First, the excitation current calculation unit 15, the torque current calculation unit 16, and the slip frequency calculation unit 17 perform a general vector control calculation. Therefore, detailed description is omitted. by providing omega _se below equation 2, and guides the output torque T _e of the induction motor in the form of the following equation (3).

[0013]

(Equation 2)

Where ω: power supply frequency, ω _re : motor speed (electrical angle), M: mutual inductance, L _r : rotor self-inductance, R _r : rotor resistance, i _T : torque current, φ _r : Rotor flux

[0015]

(Equation 3)

Here, P is the number of pole pairs. At this time, the relationship between the rotor magnetic flux φ _r and the exciting current i _φ is expressed by the following equation (4).

[0017]

(Equation 4)

However, S: Laplace operator Therefore, the arithmetic expression of the exciting current command value i _φ ′ performed by the exciting current arithmetic unit 15 is as shown in the following (Equation 5) from the above (Equation 4). .

[0019]

(Equation 5)

The calculation formula of the torque current command value i _T ′ performed by the torque current calculation section 16 is as shown in the following formula (6) from the above formula (3).

[0021]

(Equation 6)

Here, T _e ': torque command value Note that the slip frequency ω in the slip frequency calculation unit 17
The arithmetic expression of _se is as shown in the above (Equation 2).
The calculations in the motor rotation speed calculation unit 18 and the integration calculation unit 19 are as described above.

Next, the portions of the minimum steady-state loss magnetic flux calculation unit 11, the target magnetic flux calculation unit 12, the magnetic flux response variable unit 13, and the target torque calculation unit 14, which are features of the present embodiment, will be described.
In the normal vector control, the rotor magnetic flux φ _r is constant (the exciting current i _φ is constant), and only the torque current i _T is changed to obtain linearity and responsiveness of the output torque to the torque current i _T. It is. However, such a constant vector control of the rotor magnetic flux φ _r supplies a constant exciting current i _φ regardless of the load, so that the efficiency generally deteriorates at a light load. Therefore, copper loss is considered as the loss of the induction motor, and a condition for minimizing the loss is determined. First, FIG.
Γ-δ coordinate model, that is, γ-δ rotating at a power supply frequency well-known as an induction motor model
We will use a coordinate model. 3, if the vector control is established, the current components of the respective axes, is between the rotor flux phi _r, it is known that the following equation (7) is established. Here, each current component i _γs , i _δs , i
_{In γr} and i _δr , the subscripts γ and δ denote each axis component, r denotes a rotor, and s denotes a stator.

[0024]

(Equation 7)

On the other hand, the copper loss L _c of the induction motor is as shown in the following equation (8) from FIG. 3 and the (number 7).

[0026]

(Equation 8)

Where R _{s is the} stator resistance, and K ₁ and K ₂ are constants determined by the motor. In the above equation (8), the differential term dφ _r / dt of φ _r is 0 in a steady state. The slip frequency ω _se-opt that minimizes the copper loss L _c can be obtained by the following equation (Equation 9) from the condition of dL _c / dω _se = 0.

[0028]

(Equation 9)

Therefore, the slip frequency ω _se is given by ( _Equation 9)
By keeping the value of the equation, it is possible to perform driving that minimizes copper loss.
Specifically, from the above equation (2), in order to keep the slip frequency at the minimum loss slip frequency ω _se-opt , the torque current i
_It can be seen that the relationship between _T and the magnetic flux φ _r may be set as shown in the following (Equation 10).

[0030]

(Equation 10)

Next, when the above equation (10) is substituted into the above equation (3) to eliminate i _T , the relationship between the torque command value _Te ′ and the magnetic flux φ _r is _{expressed by} the following equation (11). Become like

[0032]

[Equation 11]

Therefore, when the torque command value T _e ′ is input, the minimum steady-state loss magnetic flux φ _r ′ is derived by Expression (11), and the torque current command value i _T ′ is obtained by Expression (10). The command value i _φ ′ is given by (Equation 5), and the slip frequency ω _se is (Equation 2)
By performing the calculations using the equations, it is possible to constantly drive to minimize the copper loss. At this time, the slip frequency ω _se
Corresponds to the minimum loss slip frequency ω _se-opt , and the output torque _Te follows the torque command value _Te ′. However, the torque command value T _e 'when changes stepwise, the (number 11) from the steady loss minimum flux phi _r' becomes likewise stepwise. Then, as shown in equation (5),
Since the calculation of the excitation current command value i _φ ′ includes dφ _r / dt, if a step-like change in the torque command value occurs,
The excitation current command value i _φ ′ becomes a transiently large value, so that the transient loss increases. The above phenomenon can also be seen from equation (8), where L _c is a function of dφ _r / dt. In this embodiment, only the copper loss is considered as the loss. However, as described in the above-described conventional example (Japanese Patent Laid-Open No. 23085/1990), the same problem occurs when the copper loss and the iron loss are considered. Occurs. Therefore, when the transient loss is also taken into consideration, it can not be said that keeping the slip frequency at the loss-minimum slip frequency ω _se-opt is also advantageous from the viewpoint of efficiency. Further, when the upper limit of the current is determined from the current capacity of the semiconductor switching element used in the current control PWM inverter 3 of FIG. 2, the response of the torque must be slowed during the transient when the current becomes large. For this reason, in the present embodiment, the target torque calculating unit 14 that determines the torque response and the steady-state loss minimum magnetic flux calculating unit 11
And the target magnetic flux calculator 12 are added to the general vector control calculators 15, 16, and 17 to form a control system configuration in which the torque responsiveness and the magnetic flux responsiveness can be independently varied. As the loss minimum slip frequency ω _se-opt, and transiently controlling the magnetic flux response to an optimal value corresponding to the torque response, thereby reducing the transient loss. provided by variable according to the transfer characteristic of the magnetic flux response in the target magnetic flux calculation unit 12 to the size of the target magnetic flux phi _r, which was configured to maintain low transition losses in a wider operation range of the induction motor is there.

First, the operation content of the minimum steady-state loss magnetic flux calculating unit 11 is expressed by the following equation (11), and calculates the rotor magnetic flux that minimizes the loss constantly, that is, the minimum steady-state loss magnetic flux φ _r ′. Further, the target magnetic flux calculation unit 12 for calculating a target magnetic flux phi _r
Is a filter having a steady-state gain of 1, and in this embodiment, is a first-order low-pass filter as shown in the following equation (12).

[0035]

(Equation 12)

Here, τ _{φ is} the time constant of the target magnetic flux, and S is the Laplace operator. The transfer characteristics of the target magnetic flux calculator 12 will be described later. Further, in the present embodiment, the target torque calculation unit 14 for calculating the target torque _Tm is as follows (Equation 13).
The transfer characteristics are as shown in the equation. This transfer characteristic is appropriately set according to the required responsiveness and current capacity.

[0037]

(Equation 13)

Where τ _T is the time constant of the target torque. FIG. 4 shows a torque command value T _e ′ in the control system shown in FIG.
As a characteristic diagram showing the calculated values of energy loss at the peak value and within a time of loss of the induction motor with respect to the time constant tau _phi target magnetic flux in the case of adding the input changes stepwise. From FIG. 4, it can be seen that each loss has a characteristic having a minimum value, and that each loss has a minimum value when the time constant τ _φ is a certain value. Therefore, in the control system shown in FIG. 2, when the torque response is given by Expression 12, the time constant τ _{φ of} the target magnetic flux that minimizes the transient loss obtained from FIG.
If the magnetic flux responsiveness is determined by using the equation (1), it is possible to drive the motor with little loss both in the transient state and in the steady state. That is,
Basically, the value of the time constant τ _φ at which the loss is minimized in FIG. 4 may be used in equation (11). However, since the time constant τ _φ at which the loss is minimized also changes depending on the magnitude of the target magnetic flux φ _r , attention must be paid to that point. Hereinafter, this point will be described. The As can be seen from equation (8), since the expression for the loss L _c which exists _{φ r · (dφ r / dt} ) becomes section loss L _c is the target magnetic flux when the target magnetic flux phi _r is changed affected by the size of φ _r. Therefore, in the case where the target magnetic flux phi _r rises from zero, the target magnetic flux phi _r is a stand up from one value would transition losses are different. Therefore, if the transmission characteristic of the target magnetic flux calculating unit 12 variably depending on the size of the target magnetic flux phi _r, it is possible to maintain low transition losses in a wider operation range of the motor.

FIG. 5 shows the transfer characteristic in the target magnetic flux calculator 12 given by the above equation (12),
FIG. 13 is a characteristic diagram showing a relationship between a time constant τ _φ of a target magnetic flux and a transient loss peak with respect to a step torque response (response when the torque changes stepwise) when the transfer characteristic at the time is given by Expression (13). is there. In FIG. 5, L _c0 is a target magnetic flux φ _r before the torque command value changes stepwise.
Characteristics of loss L _c but in the case of 0, L _c1 indicates the characteristics of the losses L _c when the target magnetic flux phi _r before the torque command value changes stepwise non-zero. As can be seen from FIG. 5, the time constant τ of the target magnetic flux that minimizes the transient loss depends on the magnitude of the target magnetic flux φ _r before the torque command value changes stepwise (in this state, dφ _r / dt = 0). The value of _φ is changing. Therefore, a table maps the relationship constant tau _phi time to minimize transient loss with respect to the size of the target magnetic flux phi _r, if configured to vary the time constant tau _phi in accordance with the target magnetic flux phi _r, derived Motor transient losses can be reduced over a wider operating range.

FIGS. 6 and 7 are characteristic diagrams showing speed control simulation results. FIG. 6 shows the case where the conventional method for maintaining the slip frequency at the loss-minimum slip frequency ω _se-opt in both transient and steady states is used. 7 shows the characteristics when the value (fixed value) of the time constant τ _φ that minimizes the loss in FIG. 4 is used. FIG. 8 shows the case where the value (fixed value) of the time constant τ _φ that minimizes the loss shown in FIG. 7 is used, and the case of the present embodiment (the time constant according to φ _r so that the loss is minimized). it is a characteristic diagram showing the speed control simulation results with those changing the constant tau _phi). 8, the solid line indicates the characteristic of FIG.
The broken line shows the characteristics of this embodiment. Note that the torque response was the same for both. 6 and 7, although the characteristics of the torque T and rotational speed N (responsiveness) has become both the same, characteristics of copper loss L _c is decreased apparent direction of FIG. 7 in transient In addition, at the time of steady state, the slip frequency drive is performed to minimize the copper loss as in the conventional case. Therefore, the characteristic of FIG. 7 can reduce the loss at the time of transition while maintaining the same response characteristic. FIG.
In the case where the target magnetic flux φ _r rises from 0 at t = 0 (the torque command value _Te ′ rises from 0), the losses are the same in both cases, but the target magnetic flux φ _r is at a certain value ( In the torque transition state after φ _r ≠ 0), the loss of the present embodiment is suppressed lower.

In this embodiment, the magnetic flux response is expressed by (Equation 1).
Equation (2) and the torque response are given by Equation (13), but are not necessarily limited to this transfer characteristic. For example, a secondary vibration system [G (S) = ω _n ² /
S ² + 2ζω _n S + ω _n ² ], and similarly, the magnetic flux responsiveness may be provided by a secondary vibration system. However, in order to obtain the first derivative of the target magnetic flux without performing the differential operation,
It is necessary that the relative degree between the numerator and the denominator be 1 or more. When the magnetic flux response is given by a secondary vibration system, the relationship shown in FIG. 4 is obtained for the damping rate ζ and the natural frequency ω _n , and the magnetic flux is calculated using ζ and ω _n that minimize the transient loss. I just need.

[0042]

As described above, according to the present invention, the means for calculating the minimum steady-state loss magnetic flux for minimizing the steady-state loss from the torque command value and the target torque of the induction motor from the torque command value are calculated. Means and means for causing the torque of the induction motor to follow the target torque from the steady-state minimum magnetic flux and for changing the transfer characteristic so that the transient loss is minimized. Transient loss of induction motor can be reduced,
If the current capacity is the same, the torque response can be made faster. In addition, at the time of steady state, there is obtained an effect that the motor drive control that minimizes the loss becomes possible as in the conventional method.

[Brief description of the drawings]

FIG. 1 is a block diagram showing details of a high-efficiency drive control operation unit 1 in FIG. 2;

FIG. 2 is a block diagram showing the configuration of the entire system according to an embodiment of the present invention.

FIG. 3 is a diagram showing a γ-δ coordinate model of an induction motor.

FIG. 4 shows the peak value of the loss of the induction motor and the calculated value of the energy loss within a certain time with respect to the time constant τ _φ of the target magnetic flux when an input that changes stepwise as a torque command value _Te ′ is applied. FIG.

FIG. 5 is a characteristic diagram showing a relationship between a time constant τ _φ of a target magnetic flux and a transient loss peak with respect to a response when the torque changes stepwise.

FIG. 6 is a characteristic diagram showing speed control simulation results in a conventional example.

FIG. 7 is a characteristic diagram showing speed control simulation results when a value of a time constant τ _φ that minimizes loss is used.

FIG. 8 shows a case where a value (fixed value) of a time constant τ _φ at which a loss is minimized is used and a case where the time constant τ _φ is changed according to φ _r so as to minimize a loss. FIG. 4 is a characteristic diagram showing a speed control simulation result.

[Explanation of symbols]

1: High-efficiency drive control operation unit 4: Induction motor 2: Coordinate conversion unit 5: Rotation speed sensor 3: Current control PWM inverter 6: DC power supply 11: Steady loss minimum magnetic flux operation unit 16: Torque current operation unit 12: Target magnetic flux Calculation unit 17: Slip frequency calculation unit 13: Variable magnetic flux response unit 18: Motor rotation speed calculation unit 14: Target torque calculation unit 19: Integral calculation unit 15: Excitation current calculation unit _Te ': Torque command value _Tm : Target torque φ _r ': minimum magnetic flux of steady loss φ _r : target magnetic flux N: motor rotation speed (rpm) ω: power supply frequency ω _se : slip frequency ω _re : motor rotation speed (electric angle) i _φ ': excitation current command value i _T ': Torque current command value θ: Current phase angle τ _φ : Time constant of target magnetic flux _iu ', _iv ', _iw ': Three-phase AC current command value _iu , _iv , _iw : Three-phase AC Current

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁶ , DB name) H02P 21/00 H02P 5/408-5/412 H02P 7/628-7/632

Claims (1)

(57) [Claims] 1. An induction motor control device which calculates a current command value according to a torque command value and a rotation speed of an induction motor, and drives the induction motor with a polyphase AC current corresponding to the current command value. A steady-state loss minimum magnetic flux calculation unit that calculates a rotor magnetic flux that minimizes the steady-state loss of the induction motor at the specified torque command value; A target magnetic flux calculation unit that calculates a first derivative of the magnetic flux, a magnetic flux response variable unit that changes a transfer function in the target magnetic flux calculation unit so as to minimize a transient loss according to the size of the target magnetic flux, A target torque calculator that calculates a target torque of the induction motor based on a predetermined transfer characteristic from the torque command value; and a target magnetic flux based on a circuit constant of the induction motor. A vector control calculation unit that calculates the current command value according to the first derivative of the target magnetic flux, the target torque, and the rotation speed of the induction motor; and causing the current flowing through the induction motor to follow the current command value. An induction motor control device, comprising: a motor driving unit; and controlling the output torque of the induction motor to a value corresponding to the target torque.