JPH07336113A - High frequency electrode and high frequency transmission line - Google Patents

Abstract

PURPOSE:To reduce the conductor loss and surface resistance of a high frequency electrode and to make application equipment using the electrode into miniaturization and light weight by forming the film thickness of a conductor within a prescribed range. CONSTITUTION:A part of an incident electromagnetic wave advances in the inside of the high frequency electrode, and the other is reflected on the surface of the high frequency electrode. The electromagnetic wave advancing in the inside is reflected on the back plane of the high frequency electrode, and a part of it transmits the back plane. The film thickness of the high frequency electrode is formed in the one in a range of 1.14-2.75 times, desirably 1.32 to 1.92 times, and most desirably pi/2 times the facing depth of a working frequency, therefore, the electromagnetic waves on the surface and back plane of the high frequency electrode strengthen the opponent mutually, which increases a reflection coefficient. At this time, the facing effect of current density in the inside of the high frequency electrode can be relaxed by a current excited by the electromagnetic wave reflected on the back plane of the high frequency electrode compared with the one in the case that the film thickness of the electrode is sufficiently large.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a high frequency electrode used in a high frequency band of microwave, quasi-millimeter wave or millimeter wave,
The present invention relates to a high frequency transmission line using the high frequency electrode, a high frequency resonator using the high frequency transmission line, a high frequency filter including the high frequency resonator, and a high frequency device including the high frequency electrode.

[0002]

2. Description of the Related Art In recent years, with miniaturization of electronic parts, miniaturization of devices has been achieved by using a high dielectric material even in a high frequency band such as microwave, quasi-millimeter wave or millimeter wave. When miniaturizing the device, increasing the dielectric constant, while reducing the shape as a similar shape,
In principle, there was a problem that the energy loss increased in inverse proportion to the cubic root of the volume.

The energy loss of a high frequency device can be roughly classified into a conductor loss due to a skin effect and a dielectric loss due to a dielectric material. Recent dielectric materials are
A material having a high dielectric constant and a low loss property has been developed and put into practical use. Therefore, the conductor loss is more dominant than the dielectric loss in the unloaded Q of the circuit. In the high frequency band, since the high frequency current is concentrated on the surface of the conductor due to the skin effect, the surface resistance (also referred to as the skin resistance) increases toward the conductor surface, and the conductor loss (Joule loss) increases. Here, the skin effect is a phenomenon peculiar to the transmission of a high frequency signal in which the high frequency current exponentially attenuates inside the conductor as the distance from the surface of the conductor increases. The thin area of the conductor through which this current flows is called the skin depth. For example, in the case of copper, it is about 2.2 μm at 1 GHz. However, conventionally, the film thickness of the conductor used for the electrode of the high frequency application component is configured to be sufficiently thicker than the skin depth in order to avoid radiation loss that is lost by passing through the electrode. There are also problems such as the surface roughness of the substrate and electrode film when electrodes are created by metal plating or metal baking technology, and making the electrode thickness sufficiently thicker than the skin depth will reduce loss. Was tied to doing. However, recently, a technique for forming an electrode with a high film thickness accuracy on a substrate close to a mirror surface has begun, and it has become possible to configure the electrode with an optimum film thickness.

In view of this situation, an improved symmetric stripline resonator (hereinafter referred to as a conventional resonator) capable of effectively reducing the conductor loss and obtaining a high unloaded Q is particularly desirable. It is proposed in Kaihei 4-43703. The resonator of this conventional example is a symmetrical type in which a resonant circuit is formed by a symmetrical strip line in which a strip conductor is arranged between a pair of ground conductors which are opposed to each other with a predetermined distance therebetween with a dielectric material interposed therebetween. In the stripline resonator, the strip conductor, between the pair of ground conductors, a plurality of parallel to the ground conductor,
It is characterized in that the dielectrics are arranged in a laminated manner with a predetermined gap therebetween.

The following is disclosed in the official gazette disclosing the conventional resonator. (A) It is desirable that the thickness of each strip conductor be formed to be three times as large as the skin depth or larger than that in order to effectively suppress conductor loss. That is, in the strip conductor, the skin portion where the high frequency current in the microwave band flows is increased, and the effective cross-sectional area of the strip conductor is increased. (B) One end side of the pair of strip conductors is electrically connected to each other through the through hole, and the other end side is also electrically connected to each other through the through hole. (C) The electric field distribution in the resonator is as described in
As shown in the figure, the electric field is formed from each strip conductor toward the ground conductor.

[0006]

However, since it has the structure of (a) above, it is difficult to reduce the size and weight, and further has the structure of (b) above. Therefore, there is a problem that the structure is complicated and cannot be manufactured at low cost, the reduction rate of the conductor loss is relatively small, and the no-load Q is also relatively small. The object of the present invention is to solve the above problems, to provide a simple structure as compared with the conventional example, which can significantly reduce the conductor loss, and which can reduce the size and weight of the product embodying the invention. An object is to provide an electrode, a high frequency transmission line, a high frequency resonator, a high frequency filter, and a high frequency device.

[0007]

The high-frequency electrode according to claim 1 of the present invention comprises a film-shaped conductor, and the film thickness of the conductor is 1.14 to 2.75 times the skin depth of the operating frequency. It is characterized by a double range. A high frequency electrode according to a second aspect of the present invention is characterized in that, in the high frequency electrode according to the first aspect, the film thickness of the conductor is in a range of 1.32 to 1.92 times a skin depth of a used frequency. A high frequency electrode according to a third aspect of the present invention is the high frequency electrode according to the second aspect, characterized in that the film thickness of the conductor is π / 2 times the skin depth of the operating frequency. The high-frequency electrode according to claim 4 is the same as in claim 1 or 2.
Alternatively, in the high frequency electrode described in 3, the conductor is made of a superconducting material. The high-frequency transmission line according to claim 5 is a high-frequency transmission line including at least one conductor, and the conductor is the high-frequency electrode according to claim 1, 2, 3, or 4. A high frequency transmission line according to a sixth aspect is the high frequency transmission line according to the fifth aspect, wherein the high frequency transmission line is a waveguide.
A high frequency transmission line according to a seventh aspect is the high frequency transmission line according to the fifth aspect, wherein the high frequency transmission line is a microstrip line. The high frequency transmission line according to claim 8 is the high frequency transmission line according to claim 5, wherein the high frequency transmission line is a strip line. The high frequency transmission line according to claim 9 is the same as in claim 5.
In the described high frequency transmission line, the high frequency transmission line is a coaxial line. A high frequency resonator according to a tenth aspect includes the high frequency transmission line according to any one of the fifth to ninth aspects, which has a predetermined size. The high-frequency resonator according to claim 11 is the high-frequency resonator according to claim 10, wherein the high-frequency transmission line has a length in a transmission direction equal to ¼ of a guide wavelength of a signal transmitted through the high-frequency transmission line. Is characterized by. The high-frequency resonator according to claim 12 is the high-frequency resonator according to claim 10, wherein the high-frequency transmission line has a length in the transmission direction that is equal to 1/2 of a guide wavelength of a signal transmitted through the high-frequency transmission line. Is characterized by. The high frequency filter according to claim 13 has a predetermined length.
The high frequency resonator according to one of the second aspect, an input terminal for inputting a high frequency signal to the high frequency resonator, and an output terminal for outputting a high frequency signal from the high frequency resonator. The high frequency band elimination filter according to claim 14 is coupled to a transmission line that inputs a high frequency signal at one end and outputs the high frequency signal at the other end, and is coupled to the transmission line. And a high frequency resonator. 16. The dielectric resonator according to claim 15, comprising a resonator case including a conductor, and a dielectric body having a predetermined shape and mounted in the resonator case. It is configured by the high frequency electrode according to claim 1, 2, 3 or 4. A high frequency filter according to claim 16 is a dielectric resonator according to claim 15, an input terminal electromagnetically coupled to the dielectric resonator for inputting a high frequency signal to the dielectric resonator, and the dielectric. An output terminal that is electromagnetically coupled to the resonator and that outputs a high-frequency signal from the dielectric resonator is provided. A high frequency device according to a seventeenth aspect is a high frequency device that includes an electrode and performs a predetermined high frequency operation, and the electrode has the high frequency electrode according to the first, second, third or fourth aspect.

[0008]

When an electromagnetic wave of a high frequency signal is incident on the high frequency electrode according to the present invention, a part of the electromagnetic wave proceeds inside the high frequency electrode and the other part is reflected on the surface of the high frequency electrode. The electromagnetic waves that have traveled to the inside of the high-frequency electrode are partially transmitted through the back surface of the high-frequency electrode and reflected by the other surface. Therefore, of the incident electromagnetic waves, the electromagnetic waves reflected by the high-frequency electrode are a combination of the electromagnetic waves reflected by the front surface of the high-frequency electrode and the electromagnetic waves reflected by the back surface of the high-frequency electrode. Here, since the film thickness of the high-frequency electrode according to the present invention is formed to be in the range of 1.14 to 2.75 times the skin depth of the used frequency, it is reflected on the front and back surfaces of the high-frequency electrode. The electromagnetic waves generated by the high-frequency electrodes strengthen each other, and the reflection coefficient of the high-frequency electrode becomes larger than that when the film thickness of the electrode is sufficiently large. At this time, the current density distribution inside the high-frequency electrode has a gentle gradient due to the current excited by the electromagnetic wave reflected on the back surface of the high-frequency electrode, without abrupt attenuation as when only the skin effect is considered. This mitigates the skin effect and reduces the surface resistance.

Further, in the high frequency transmission line, since the conductor is formed by using the high frequency electrode, it has a smaller surface resistance R _s like the above electrode,
The high-frequency transmission line has a smaller transmission loss than the conventional example. Furthermore, since the high-frequency resonator is provided with the high-frequency transmission line having a predetermined size, its transmission loss is smaller than that of the conventional example. Therefore, the high-frequency resonator is compared with the conventional example. And has a large unloaded Q. In the dielectric resonator, since the conductor of the resonator case is formed by the high frequency electrode, the dielectric resonator has a large unloaded Q as compared with the conventional example. Further, since the high-frequency filter has the high-frequency resonator having a predetermined length and a large unloaded Q, the high-frequency filter has low loss and excellent selectivity. Furthermore, in the high frequency band elimination filter, since the high frequency resonator having a predetermined length and a large unloaded Q operates as a trap circuit, the high frequency band elimination filter has excellent band elimination characteristics. Furthermore, in the high-frequency device, since the electrode has the high-frequency electrode, the high-frequency device has a smaller conductor loss than the conventional example.

[0010]

Embodiments of the present invention will be described below with reference to the drawings. In the attached drawings, the same components are designated by the same reference numerals. <First Embodiment> FIG. 1 is a perspective view of a bandpass filter according to a first embodiment of the present invention. As shown in FIG. 1, a 1/2 wavelength microstrip line type resonance is provided by a strip conductor 21 formed to have a thickness of π / 2 times the skin depth δ ₀ and a dielectric substrate 10 having a ground conductor 11. Vessels are configured. The band pass filter of the first embodiment is characterized by including the above-mentioned 1/2 wavelength microstrip line type resonator. Where the skin depth δ
₀ is the angular frequency ω _{0 at} the resonance frequency f ₀ of the ½ wavelength microstrip line resonator and the magnetic permeability μ _{0 in} vacuum.
And the electric conductivity σ of the strip conductor 21 are used, and are expressed by Formula 1.

[0011]

## EQU1 ## δ ₀ = √ (2 / ω ₀ μ ₀ σ)

As shown in FIG. 1, the ground conductor 1 is formed on the entire back surface.
A strip-shaped strip conductor 21 having a length in the longitudinal direction of λg / 2 is formed on the dielectric substrate 10 on which 1 is formed. Here, λg is a guide wavelength. As a result, the strip conductor 21, the ground conductor 11, and both conductors 21, 1
A TEM mode 1/2 wavelength microstrip line type resonator is constituted by the dielectric substrate 10 sandwiched between the two. The resonance frequency f ₀ of the half-wavelength microstrip line resonator configured as described above is, as is well known, the length of the strip conductor 22 in the longitudinal direction and the dielectric substrate 1.
It is determined by the dielectric constant of 0 and the thickness.

Further, on the dielectric substrate 10, the input terminal conductor 12 is formed in close proximity to one end of the strip conductor 21 in the longitudinal direction by a predetermined gap g1 and electromagnetically coupled to each other. , Output terminal conductor 13
Are formed close to the other end of the strip conductor 21 in the longitudinal direction by a predetermined gap g2 and are electromagnetically coupled to each other. In addition, in the first embodiment, the coupling between the input terminal conductor 12 and one end of the strip conductor 21 and the coupling between the output terminal conductor 13 and the other end of the strip conductor 21 are capacitive coupling. Here, the dielectric substrate 10
Is made of sapphire, which is a single crystal of alumina, and the ground conductor 11 and the strip conductor 21 are made of, for example, C
It is made of a conductor having electrical conductivity such as u, Ag or Au.

The resonance frequency f is applied to the input terminal conductor 12 of the bandpass filter according to the first embodiment having the above-described structure.
_When a high frequency signal having ₀ is input, the input terminal electrode 1
2 and the strip conductor 21 are electromagnetically coupled to each other,
The two-wavelength microstrip line resonator is excited to be in a resonance state. Further, the strip conductor 21 and the output terminal electrode 13 are electromagnetically coupled, and the high frequency signal is output from the output terminal electrode 13. When a signal having a frequency different from the resonance frequency f ₀ is input to the input terminal conductor 12, electromagnetic waves corresponding to the signal reflected at the longitudinal ends of the ½ wavelength microstrip line resonator are generated. The two half-wavelength microstrip line resonators do not enter into a resonance state because they cancel each other out. Therefore, the signal is not output from the output terminal electrode 13. As described above, the bandpass filter according to the first embodiment has the bandpass characteristic.

Next, the optimum film thickness of the strip conductor 21 having the highest Q in the 1/2 wavelength microstrip line resonator will be determined. FIG. 2A is a distributed constant type equivalent circuit in the thickness direction of the strip conductor 21 including an air layer, and as shown in FIG. 2A, it is a distributed constant circuit including loss resistance. The distributed constant type equivalent circuit is a strip conductor 2
Two terminals T virtually provided on the first surface
1, T2 and two terminals T3, T4 virtually provided on the second surface of the strip conductor 21. Here, each unit circuit of the distributed constant type equivalent circuit has a unit inductance μ ₀ dx provided in a direction parallel to the thickness direction and a unit capacitance ε ₀ dx provided in a direction perpendicular to the thickness direction. A parallel circuit with a unit conductance σdx is provided, and the parallel circuit and the unit inductance μ ₀ dx are connected in an inverted L shape. The distributed constant type equivalent circuit is configured by connecting a plurality of the unit circuits in cascade in the thickness direction,
The impedance Z _{L of the} air layer is connected to the two terminals T3 and T4 on the air layer side of the equivalent circuit. Here, σ is the conductivity of the strip conductor 21, ε ₀ is the dielectric constant in vacuum,
μ ₀ is the magnetic permeability in vacuum, dx is the minute length of the strip conductor 21 in the thickness direction, Δξ is the film thickness of the strip conductor 21,
Z _L is the impedance of the air layer.

Further, the equivalent circuit of FIG.
It can be converted into the lumped constant type equivalent circuit of (b).
The lumped constant type equivalent circuit is configured by connecting two complex impedances Z provided in a direction parallel to the thickness direction and a complex admittance Y provided in a direction perpendicular to the thickness direction in a T-shape. It Where complex impedance Z,
The complex admittance Y and the impedance Z _{L of the} air layer are
These are expressed by equations 2, 3, and 4, respectively. Further, a value obtained by dividing the film thickness Δξ of the strip conductor 21 by the skin depth δ ₀ is defined as the normalized conductor film thickness ξ of the strip conductor 21, and is expressed as in Equation 5.

[0017]

[Number 2] Z = [(1 + j) / σδ 0] · tanh [(1 + j) ξ / 2]

[Formula 3] Y = [σδ ₀ / (1 + j)] ・ sinh [(1 + j) ξ]

[Formula 4] Z _L = √ (μ ₀ / ε ₀ ).

## EQU5 ## ξ≡Δξ / δ ₀

Further, the surface impedance Z _S when the equivalent circuit of FIG. 2B is viewed from the left end is expressed by the equation 6. Here, since the impedance Z _{L of the} air layer is sufficiently larger than the complex impedance Z and the complex admittance Y as shown in Expression 7, it is assumed that the right end of the equivalent circuit of FIG. 2B is the open end. You can use an approximation,
The surface impedance Z _s is expressed by Equation 8.

[0019]

## EQU6 ## Z _S = Z + [Y + (Z + Z _L ) ^-1 ] ^-1

(7) Z _L σδ ₀ = σδ ₀ √ (μ ₀ / ε ₀ ) ≈∞

(8) Z _S = Z + 1 / Y

Further, by substituting the complex impedance Z and the complex admittance Y represented by the equation 2 into the surface impedance Z _S represented by the equation 8, the surface impedance is represented by the equation 9.

[0021]

## EQU9 ## Z _S = [(1 + j) / σδ ₀ ] / [tanh (1 + j) ξ]

Further, the surface impedance Z _S can be expressed by the equation 10 using the surface resistance R _S and the surface reactance X _S.

[0023]

[Formula 10] Z _S = R _S + jX _S

Here, the 1/2 microstrip line type resonator has the highest Q when the surface resistance R _S is the smallest. Next, in order to obtain the surface resistance R _S , when the surface impedance Z _S expressed by the equation 9 is divided into a real part and an imaginary part, the surface resistance R _S and the surface reactance X _S are respectively expressed by the equation 11 and the number. It can be expressed as 12.

[0025]

## EQU11 ## R _S = (sinh2ξ + sin2ξ) / [σδ ₀ (cosh2
ξ-cos2ξ)]

## EQU12 ## X _S = (sinh2ξ-sin2ξ) / [σδ ₀ (cosh2
ξ-cos2 ξ)]

FIG. 3 is a graph showing the relationship between the standardized surface resistance σδ ₀ R _S obtained by multiplying the surface resistance R _S by σδ ₀ and the standardized conductor film thickness ξ, which is obtained by using the equation 11. As is apparent from FIG. 3, when the normalized conductor film thickness ξ is a specific value between 1 and 2, the normalized surface resistance σδ ₀ R _S has a minimum value which is an extreme value. With the normalized film thickness ξ where the surface resistance R _S is the minimum,
The differential coefficient ∂R _S / ∂ξ of the surface resistance R _S shown in 3 with respect to the normalized conductor film thickness ξ becomes zero. Therefore, in order to obtain the normalized film thickness ξ that minimizes the surface resistance R _{S, the} normalized conductor film thickness ξ that satisfies the equation 13 may be obtained.

[0027]

[Equation 13] ∂R _S / ∂ξ ＝ − (2sinh2ξ ・ sin2ξ) / (c
osh2ξ-cos2ξ) ² = 0

The normalized film thickness ξ when the differential coefficient ∂R _S / ∂ξ expressed by the equation 13 becomes 0, where n is a positive integer,
It is expressed by Equation 14. In particular, the normalized film thickness ξ when n = 1
Is expressed by Expression 15, and at this time, the surface resistance R _S becomes the minimum value R _S min expressed by Expression 16.

[0029]

## EQU4 ## ξ = nπ / 2, n = 1, 2, 3 ,. ． ． ． ．

Equation 15 ξ = π / 2

## EQU16 ## R _S min = (1 / σδ ₀ ) tanh (π / 2) ≈0.9
17 (1 / σδ ₀ )

Here, since the normalized film thickness ξ of the strip conductor 21 is a value standardized by the skin depth δ ₀ as defined by the equation 5, the strip conductor having a dimension of physical length. The film thickness Δξ of 21 is given by Expression 17.

[0031]

Δξ = (π / 2) δ ₀ = (π / 2) √ [2 / (ω ₀ μ ₀ σ)]

As is clear from the above results and FIG. 3, the normalized surface resistance σδ ₀ R _S decreases as the normalized conductor film thickness ξ increases, and the normalized conductor film thickness ξ becomes 1.14. At this time, the normalized surface resistance σδ ₀ R _S 1.0 is obtained. Furthermore, as the normalized conductor film thickness ξ increases, the normalized surface resistance σ
δ ₀ R _S further decreases, and when the normalized conductor film thickness ξ is π / 2, the normalized surface resistance σ δ ₀ R _S has a minimum value of 0.917. Beyond this minimum value, the standardized surface resistance σ δ _{0 RS} increases with the increase in the standardized conductor film thickness ξ, and when the standardized conductor film thickness ξ is 2.75, the standardized surface resistance σ δ ₀ R _S becomes 1.0 again. Although not shown in FIG. 3, the normalized surface resistance σδ ₀ R _S oscillates near 1.0 and converges to 1.0 as the normalized film thickness ξ becomes larger than 2.75.

As described above, the normalized conductor film thickness ξ is π / 2.
When the film thickness Δξ of the strip conductor 21 is π / 2 times the skin depth δ ₀ , the surface resistance R _S is such that the film thickness Δξ of the strip conductor 21 is sufficiently thicker than the skin depth δ _0. 0.91 smaller than 1 / σδ ₀ which is the surface resistance R _S at
It is the minimum value of 7 / σδ ₀ . This is a strip conductor 21
The surface resistance R _S is 8.3 as compared with the case where the film thickness ξ is sufficiently thick.
It has been reduced by%. Here, by utilizing the fact that the Q value of the resonator is proportional to the reciprocal of the surface resistance R _S , it is converted into a Q increase rate when the Q value when the film thickness Δξ of the strip conductor 21 is sufficiently thick is used as a reference. Then, Q increase rate is 9.03
%become. This is also the maximum Q rise rate. Therefore,
In the first embodiment, the strip conductor 21 is most preferably configured so that its film thickness Δξ is π / 2 times the skin depth δ ₀ at the operating frequency. This minimizes the surface resistance R _S of the strip conductor 21,
The Q value of the resonator reaches a maximum value by 9.03% higher than the Q value when the film thickness Δξ of the strip conductor 21 is sufficiently thicker than the skin depth δ ₀ .

The range of the normalized conductor film thickness ξ where the Q increase rate is 90% or more of the maximum value 9.03% is 1.42.
1 ≦ normalized conductor film thickness ξ ≦ 1.751. Therefore, in the first embodiment, the strip conductor 21 is more preferably such that its film thickness Δξ is the skin depth δ at the operating frequency.
_It may be configured to be in the range of 1.421 times to _0. 1.751 times. As a result, the surface resistance R _S of the strip conductor 21 is reduced, and the rate of increase in Q is its maximum value of 9.
90% or more of 03% can be secured.

Further, the maximum Q increase rate is 9.03%.
The range of the normalized conductor film thickness ξ, which is 80% or more of the
64 ≦ normalized conductor film thickness ξ ≦ 1.841. Therefore,
In the first embodiment, the strip conductor 21 may more preferably be configured such that its film thickness Δξ is in the range of 1.364 to 1.841 times the skin depth δ _{0 at} the operating frequency. . As a result, the strip conductor 21
The surface resistance R _{S of} is reduced, and the Q increase rate can be secured at 80% or more of the maximum value of 9.03%.

Furthermore, the Q increase rate has its maximum value of 9.0.
The range of the normalized conductor film thickness ξ, which is 70% or more of 3%, is
1.32 ≦ normalized conductor film thickness ξ ≦ 1.92. Therefore, in the first embodiment, the strip conductor 21 is more preferably configured such that the film thickness Δξ is in the range of 1.32 to 1.92 times the skin depth δ ₀ at the operating frequency. Good. As a result, the strip conductor 21
The surface resistance R _{S of} is reduced, and the Q increase rate can be secured at 70% or more of the maximum value of 9.03%.

Furthermore, when the Q increase rate is larger than 0, that is, the Q value of the resonator is the strip conductor 21.
When the film thickness Δξ of is larger than the Q value of the resonator when it is sufficiently thicker than the skin depth δ ₀ , the range of the normalized conductor film thickness ξ is 1.14 ≦ normalized conductor film thickness ξ. ≦ 2.75. Therefore, in the first embodiment, the strip conductor 2
1 may preferably be configured such that the film thickness Δξ is in the range of 1.14 to 2.75 times the skin depth δ ₀ at the used frequency. As a result, the strip conductor 21
The surface resistance R _s of the resonator is reduced and the Q increase rate becomes larger than 0. The Q value of the resonator is such that the film thickness Δξ of the strip conductor 21 is sufficiently thicker than the skin depth δ ₀ . It becomes larger than the Q value of.

Next, the reason why the surface resistance R _S of the strip conductor 21 configured as described above is reduced will be qualitatively described with reference to FIG. FIG. 4 is a cross-sectional view of the strip conductor 21 including a part of the dielectric substrate 10 and a part of the air layer, showing relaxation of the skin effect and reduction of the surface resistance R _S. Figure 4
Then, hatching is omitted. A part of the incident electromagnetic wave 102 that is incident on the strip conductor 21 through the dielectric substrate 10 passes through the boundary between the dielectric substrate 10 and the strip conductor 21 to become a transmitted electromagnetic wave 106, and the other part is the dielectric substrate 10 and the strip conductor 21. Electromagnetic waves reflected at the boundary of 10
4. Part of the transmitted electromagnetic wave 106 is the strip conductor 2
1 and the boundary between the air layer and transmitted electromagnetic wave 103,
The other is reflected at the boundary between the strip conductor 21 and the air layer,
It becomes the reflected electromagnetic wave 105. Therefore, of the incident electromagnetic waves, the electromagnetic waves reflected from the strip conductor 21 into the dielectric substrate 10 are a combination of the reflected electromagnetic waves 104 and the reflected electromagnetic waves 105. At this time, if the strip conductor 21 is thinner than the skin depth δ ₀ , the shielding effect is reduced and the amplitude of the transmitted electromagnetic wave 103 passing through the strip conductor 21 is increased.
Radiation loss increases. Further, when the strip conductor 21 is sufficiently larger than the skin depth δ ₀ , the shielding effect becomes large, but most of the energy of the transmitted electromagnetic wave 106 is lost inside the strip conductor 21 and the conductor loss becomes large.
When the strip conductor 21 has an appropriate thickness, the reflected electromagnetic wave 104 and the reflected electromagnetic wave 105 strengthen each other inside the dielectric substrate 10, and as a result, the reflection coefficient increases. At this time, the current density distribution in the direction from the dielectric substrate 10 inside the strip conductor 21 toward the air layer is the reflected electromagnetic wave 105.
Unlike the current distribution when the strip conductor 21 is sufficiently thick, the current excited by does not exponentially attenuate. As a result, the skin effect is alleviated and the surface resistance R _S is reduced.

From the above results, the bandpass filter of the first embodiment is a 1/2 wavelength microstrip line formed by using the strip conductor 21 having a film thickness of π / 2 times the skin depth δ _0. Type resonator has a high Q, so it is small
Light weight, low loss, and excellent selectivity.

In the band pass filter of the first embodiment described above, the film thickness of only the strip conductor 21 is set to π of the skin depth δ ₀ .
Although the ground conductor 11 is formed to have a thickness of 1/2, only the ground conductor 11 may be formed to have a depth of π / 2 times the skin depth δ ₀ . Alternatively, both the strip conductor 21 and the ground conductor 11 may be formed to be π / 2 times the skin depth δ ₀ . Furthermore, a protective dielectric may be formed on the strip conductor 21 of the first embodiment, or the entire bandpass filter may be surrounded by the protective dielectric.

In the first embodiment described above, the filter using the 1/2 wavelength microstrip line type resonator using the strip conductor 21 has been described, but the present invention is not limited to this, and 1 / A filter using a 4-wavelength microstrip line type resonator may be configured.

<Second Embodiment> FIG. 5 is a perspective view of a quarter wavelength microstrip line type band elimination filter according to a second embodiment of the present invention. In the second embodiment, FIG.
As shown in, the microstrip line is formed by forming the strip conductor 41 on the dielectric substrate 10 having the ground conductor 11 formed on the entire back surface. And
The strip conductor 21 having a length of ¼λg and a film thickness Δξ of δ ₀ π / 2 is connected to the strip conductor 41 of the microstrip line so that the strip conductor 21 is electromagnetically coupled to the gap g3. Only close away,
Further, the strip conductor 21 is formed such that the longitudinal direction thereof is parallel to the longitudinal direction of the strip conductor 41. As a result, the strip conductor 21 and the ground conductor 11 constitute a quarter-wavelength microstrip line type resonator.

In the circuit configured as described above, the 1/4 wavelength microstrip line resonator is electromagnetically coupled to the microstrip line to operate as a trap circuit.

The quarter-wavelength microstrip line type band elimination filter circuit configured as described above has a δ ₀ π /
1 with a strip conductor 21 having a film thickness Δξ of 2
The four-wavelength microstrip line type resonator has a high unloaded Q and thus has excellent band stop characteristics.

In the above second embodiment, the microstrip line is used, but the present invention is not limited to this, and
It may be configured by a transmission line such as a coplanar line, a slot line, or a triplate type strip line.

<Modification> The high frequency electrode according to the present invention has a TM mode single mode dielectric resonance in which a core dielectric and a cavity are integrally molded, as disclosed in, for example, Japanese Patent Laid-Open No. 3-292006. It can be applied to the electrode film portion provided on the outer surface of the cavity in the container. Further, as the TM mode dielectric resonator, the TM
The mode is not limited to the single mode type, but is disclosed in, for example, JP-A-63-
It can be applied to a dual mode type dielectric resonator as disclosed in Japanese Patent No. 313901 (for example, see FIG. 23), and further, it can be applied to JP-A-61-157101.
The present invention can be applied to a triple mode type dielectric resonator as disclosed in Japanese Patent Publication. That is, the high-frequency electrode according to the present invention can be applied to the electrode film portion of the TM mode dielectric resonator regardless of the number of modes used.

FIG. 6 shows an example of a modified dual mode type dielectric resonator 75. A double-mode dielectric resonator 75 is provided by providing a cross-shaped dielectric 76 integrally formed with the case 77 in the center of a square cylindrical resonator case 77 in which the outer surface of the dielectric is metallized. It is configured. Here, as the electrode of the resonator case 77, the high frequency electrode according to the present invention is used. As a result, the surface resistance of the electrode can be significantly reduced, so that the loss of the dielectric resonator can be reduced and the no-load Q can be increased.

FIG. 7 shows an example of a modified TM _{01 δ} mode type two-stage dielectric bandpass filter 80. The band pass filter 80 is configured as follows. Peripheral electrode 8
SMA connectors 83 and 84 for input and output are attached to both ends of a cylindrical dielectric tube 81 having 2, respectively, while the ground conductors of the SMA connectors 83 and 84 are connected to the outer peripheral electrode 82. SMA connector 83, 8
Monopole antennas 85 and 86 facing each other in the dielectric tube 81 are connected to the four central conductors, respectively. In the dielectric tube 81 between the monopole antennas 85 and 86, a cylindrical shape is provided with a ring-shaped dielectric support bases 89 and 90 that are separated by a predetermined distance and are inscribed in the inner peripheral surface of the dielectric tube 81. Two dielectric resonators 87 and 88 are provided. Also in the band pass filter 80, the outer peripheral electrode 8
2 uses the high frequency electrode according to the present invention. As a result, the surface resistance of the outer peripheral electrode 82 can be significantly reduced, so that the loss of the dielectric filter can be reduced and the no-load Q can be increased.

Further, in the following modified examples, by using the high-frequency electrode according to the present invention, the surface resistance of the electrode is significantly reduced as compared with the conventional one, and by this,
Transmission loss can be significantly reduced.

FIG. 8 is a perspective view of a microstrip line using the high frequency electrode according to the present invention. As shown in FIG. 8, the strip conductor 22 is formed on the dielectric substrate 10 having the ground conductor 11 on the back surface to form a microstrip line. The microstrip line is characterized by using the high frequency electrode according to the present invention for the slip conductor 22 and the ground conductor 11. The high frequency electrode may be used only for the strip conductor 22 or the high frequency electrode may be used only for the ground conductor 11.

FIG. 9 is a perspective view of a triplate type strip line using the high frequency electrode according to the present invention.
As shown in FIG. 9, a strip conductor 23 is formed in the central portion in the thickness direction of the dielectric substrate 10 having ground conductors 31a and 31b on both sides so as to be parallel to the two ground conductors 31a and 31b. A stripline is constructed. The strip line includes a strip conductor 23 and a ground conductor 31.
The high frequency electrode according to the present invention is used for a and 31b. The microstrip conductor 23
You may use a high frequency electrode only for the ground conductor 31a,
The high frequency electrode may be used for at least one of the electrodes 31b.

FIG. 10 is a perspective view of a symmetrical type coplanar line using the high frequency electrode according to the present invention. Figure 10
As shown in FIG. 3, a strip conductor 24 is formed on one surface of the dielectric substrate 10, and two ground conductors 32a and 32b are formed close to each other so as to sandwich the strip conductor 24 to form a coplanar line. The coplanar line is characterized by using the high frequency electrode according to the present invention for the strip conductor 24 and the ground conductors 32a and 32b. A high frequency electrode may be used only for the strip conductor 24, or at least one of the ground conductors 32a and 32b may be used.
High frequency electrodes may be used for only one of them.

FIG. 11 is a perspective view of an asymmetrical coplanar line using the high frequency electrode according to the present invention. Figure 1
As shown in FIG. 1, a strip conductor 2 is formed on the dielectric substrate 10.
5 is formed and the ground conductor 32 is formed adjacent to one side of the strip conductor 25 to form a coplanar line. The coplanar line is characterized by using the high-frequency electrode according to the present invention for the strip conductor 25 and the ground conductor 32. A high frequency electrode may be used only for the strip conductor 25, or a high frequency electrode may be used only for the ground conductor 32.

FIG. 12 is a perspective view of a slot line using the high frequency electrode according to the present invention. As shown in FIG. 12, the slot lines are connected to the ground conductors 51a and 51a on one surface of the dielectric substrate 10 so as to face each other closely.
b is formed to form a slot line. The slot line is characterized by using the high frequency electrode according to the present invention for the ground conductors 51a and 51b. The high frequency electrode may be used for at least one of the ground conductors 51a and 51b.

FIG. 13 is a perspective view of a suspended line using the high frequency electrode according to the present invention. As shown in FIG. 13, inside a rectangular conductor tube 101 composed of a conductor tube having a rectangular cross section, a dielectric substrate 10 provided with a strip conductor 26 has its side surfaces contacting both side surfaces of the rectangular conductor tube 101. In addition, the suspended line is configured so as not to contact the upper and lower surfaces and fixed in parallel. The suspended line is characterized in that the strip conductor 26 uses the high-frequency electrode according to the present invention.

FIG. 14 is a perspective view of a dielectric rectangular waveguide using the high frequency electrode according to the present invention. As shown in FIG. 14, the ground conductor 33 is formed on the surface of the dielectric prism 61 having a rectangular cross section to form a dielectric rectangular waveguide. The dielectric rectangular waveguide is characterized in that the ground conductor 33 uses the high-frequency electrode according to the present invention.

FIG. 15 is a perspective view of a dielectric ridge waveguide using the high frequency electrode according to the present invention. As shown in FIG. 15, the ground conductor 34 is formed on the surface of the dielectric prism 62 having an H-shaped cross section to form a dielectric ridge waveguide. The dielectric ridge waveguide is characterized by using the high frequency electrode according to the present invention for the ground conductor 34.

FIG. 16 is a perspective view of an image line using the high frequency electrode according to the present invention. As shown in FIG. 16, the ground conductor 35 having a width sufficiently larger than the diameter of the semicircle is formed on the bottom surface of the dielectric semicylindrical 63 having a semicircular cross section to form an image line. The image track is
The high frequency electrode according to the present invention is used for the ground conductor 35.

FIG. 17 is a perspective view of an H line using the high frequency electrode according to the present invention. The H line is shown in FIG.
As shown in FIG. 3, the ground conductor 3 having a rectangular cross section is provided on the upper and lower surfaces of the dielectric prism 64, which is sufficiently wider than the upper surface or the lower surface thereof.
An H line is formed by forming 6a and 36b. The H
The line is characterized by using the high frequency electrode according to the present invention for the ground conductors 36a and 36b. Ground conductor 36
The high frequency electrode may be used for at least one of a and 36b.

FIG. 18 is a perspective view of a dielectric circular waveguide using the high frequency electrode according to the present invention. As shown in FIG. 18, the ground conductor 37 is formed on the circumferential surface of the dielectric cylinder 65 having a circular cross section to form a dielectric circular waveguide. The dielectric circular waveguide is characterized by using the high frequency electrode according to the present invention for the ground conductor 37.

FIG. 19 is a perspective view of a coaxial line using the high frequency electrode according to the present invention. As shown in FIG. 19, the coaxial conductor is configured by forming the center conductor 27 on the inner peripheral surface and the ground conductor 38 on the outer peripheral surface of the dielectric cylinder 66 having an annular cross section. The coaxial line is characterized by using the high frequency electrode according to the present invention for the center conductor 27 and the ground conductor 38. The high frequency electrode according to the present invention may be used only for the center conductor 27, or the high frequency electrode according to the present invention may be used for only the ground conductor 38.

FIG. 20 is a perspective view of a dielectric coaxial resonator using the high frequency electrode according to the present invention. As shown in FIG. 20, the center conductor 28 is formed on the inner peripheral surface and the ground conductor 39 is formed on the outer peripheral surface of the dielectric cylinder 67 having an annular cross section and a length of λg / 4. A short-circuit conductor 29 is formed on one end face of 67 to form a dielectric coaxial resonator. The dielectric coaxial resonator is characterized by using the high frequency electrode according to the present invention for the center conductor 28, the ground conductor 39, and the short-circuit conductor 29. The high frequency electrode according to the present invention may be used for only one of the center conductor 28, the ground conductor 39, and the short-circuit conductor 29, or for any two of them.

In the above embodiments, the ground conductor 11, the strip conductor 21 and the like are made of, for example, Cu, Ag or Au.
However, the present invention is not limited to this, and a superconductor (superconductor) shown below may be used as at least one material such as the ground conductor 11 and the strip conductor 21. Good. (A) Pure metal superconducting materials such as Nb and Pb. (B) Alloy-based superconducting materials such as Nb-Ti alloy system and Nb-Zr alloy system. (C) An intermetallic compound-based superconducting material such as Nb ₃ Sn or V ₃ Si. (D) Ceramic-based oxide superconducting material of which one example is shown below (d-1) For example, La _1.85 Sr _0.15 CuO ₄ and other La
_2-x Ba _x CuO _4-δ system or La _2-x Sr _x CuO _4-δ system. (D-2) For example, YBa ₂ Cu ₃ O ₇ or the like YBa ₂ Cu ₃ O
_7-δ (oxygen deficiency δ = 0 to 1). (D-3) Bi-Sr-Ca-Cu-O system, where the Bi-Sr-Ca-Cu-O system superconducting material is Bi ₂
After calcination of the mixed powder of O ₃ , SrCO ₃ , CaCO ₃ and CuO at a temperature of 800 to 870 ° C., 850
It is obtained by sintering in air at a temperature of ˜880 ° C. (D-4) Tl-Ba-Ca-Cu-O system, where the Tl-Ba-Ca-Cu-O system superconducting material is Tl ₂
Powders of O ₃ , CaO, BaO and CuO were mixed and molded, and then sealed in a quartz tube containing oxygen at 1 atm, and 880
The main component Tl _{2 is} obtained by heating at a temperature of ° C for 3 hours.
A superconducting material of CaBa ₂ Cu ₂ O _x can be obtained. (D-5) EBCO system, (d-6) BPSCCO system. (E) Organic superconducting material of which one example is shown below (e-1) For example, tetramethyltetraselenafulvalene (tetramethyltetrasele) such as (TMTSF) ₂ ClO ₄
nafulvalene (TMTSF) based superconducting material. (E-2) Bis (ethylenedithiolo) tetrathiafulvalene (bis (et (et)) such as β (BEDT-TTF) ₂ I ₃
hylenedithiolo) tetrathiaful-valene: BEDT-TT
F) -based superconducting material. (E-3) A dmit superconducting material.

[0064]

The film thickness of the high-frequency electrode according to the present invention is in the range of 1.14 to 2.75 times the skin depth of the operating frequency, preferably from 1.32 times the skin depth of the operating frequency. Since it is formed in a range of 1.92 times, most preferably π / 2 of the skin depth of the used frequency, the current density distribution inside the high frequency electrode is an electromagnetic wave reflected on the back surface of the high frequency electrode. Due to the current excited by, there is no steep decay as when only the skin effect is taken into consideration, resulting in a gentle slope. As a result, the skin effect is alleviated, and the surface resistance of the high-frequency electrode according to the present invention can be significantly reduced as compared with the conventional one. Moreover, since the high-frequency electrode is composed of one layer, it can be manufactured at a lower cost than the conventional example. Therefore, by using the high frequency electrode of the present invention, a high frequency transmission line having a smaller transmission loss, an extremely large unloaded Q
The high-frequency resonator, the high-frequency filter with excellent selectivity, or the low-loss high-frequency device can be realized in a smaller size and lighter weight and at a lower cost.

[Brief description of drawings]

FIG. 1 is a perspective view of a bandpass filter according to a first embodiment of the present invention.

FIG. 2 is an equivalent circuit in the thickness direction of the strip conductor 21 including the air layer of FIG. 1, in which (a) is a distributed constant type equivalent circuit and (b) is a lumped constant type equivalent circuit. Is.

FIG. 3 Normalized surface resistance σδ ₀ R _S and standardized conductor film thickness ξ
It is a graph showing the relationship with.

4 is a cross-sectional view of strip conductor 21 including a portion of dielectric substrate 10 and a portion of an air layer of FIG. 1 showing relief of skin effect and reduction of surface resistance.

FIG. 5 is a perspective view of a band elimination filter that is a second embodiment according to the present invention.

FIG. 6 is a perspective view showing an example of a modified TM ₁₁₀ dual-mode dielectric resonator.

FIG. 7 is a vertical cross-sectional view showing an example of a modified TM _{01 δ} mode type two-stage dielectric bandpass filter.

FIG. 8 is a perspective view of a microstrip line using a high frequency electrode according to the present invention.

FIG. 9 is a perspective view of a triplate-type strip line using a high-frequency electrode according to the present invention.

FIG. 10 is a perspective view of a symmetrical coplanar line using a high frequency electrode according to the present invention.

FIG. 11 is a perspective view of an asymmetrical coplanar line using a high frequency electrode according to the present invention.

FIG. 12 is a perspective view of a slot line using the high-frequency electrode according to the present invention.

FIG. 13 is a perspective view of a suspended line using a high frequency electrode according to the present invention.

FIG. 14 is a perspective view of a dielectric rectangular waveguide using a high frequency electrode according to the present invention.

FIG. 15 is a perspective view of a dielectric ridge waveguide using a high frequency electrode according to the present invention.

FIG. 16 is a perspective view of an image line using a high frequency electrode according to the present invention.

FIG. 17 is a perspective view of an H line using a high frequency electrode according to the present invention.

FIG. 18 is a perspective view of a dielectric circular waveguide using a high frequency electrode according to the present invention.

FIG. 19 is a perspective view of a coaxial line using a high frequency electrode according to the present invention.

FIG. 20 is a perspective view of a dielectric coaxial resonator using a high frequency electrode according to the present invention.

[Explanation of reference numerals] 10 ... Dielectric substrate, 21, 22, 23, 24, 25, 26, 41 ... Strip conductor, 27, 28 ... Center conductor, 11, 31a, 31b, 32a, 32b, 32, 33,
34, 35, 36a, 36b, 37, 38, 39, 51
a, 51b ... ground conductor, 12 ... input terminal conductor, 13 ... output terminal conductor, g1, g2, g3 ... gap, 75 ... TM ₁₁₀ dual mode dielectric resonator, 76 ... dielectric 77 ... resonant _Container case, 80 ... TM _{01 δ} mode type two-stage dielectric bandpass filter, 82 ... _Peripheral electrode.

─────────────────────────────────────────────────── ─── Continuation of the front page (51) Int.Cl. ⁶ Identification number Internal reference number FI Technical indication H01P 11/00 ZAA E H01P 7/04 ZAA F20 11/00 ZAA J F6-7,20

Claims (17)

[Claims] 1. A high-frequency electrode comprising a film-shaped conductor, wherein the film thickness of the conductor is in the range of 1.14 to 2.75 times the skin depth of the operating frequency. 2. The high frequency electrode according to claim 1, wherein the film thickness of the conductor is in the range of 1.32 to 1.92 times the skin depth of the operating frequency. 3. The high frequency electrode according to claim 2, wherein the film thickness of the conductor is π / 2 times the skin depth of the operating frequency. 4. The high frequency electrode according to claim 1, wherein the conductor is made of a superconducting material. 5. A high-frequency transmission line comprising at least one conductor, wherein the conductor is the high-frequency electrode according to claim 1, 2, 3 or 4. 6. The high frequency transmission line according to claim 5, wherein the high frequency transmission line is a waveguide. 7. The high frequency transmission line according to claim 5, wherein the high frequency transmission line is a microstrip line. 8. The high frequency transmission line according to claim 5, wherein the high frequency transmission line is a strip line. 9. The high frequency transmission line according to claim 5, wherein the high frequency transmission line is a coaxial line. 10. The method according to claim 5, which has predetermined dimensions.
A high-frequency resonator comprising the high-frequency transmission line according to one of the above. 11. The high frequency resonator according to claim 10, wherein the high frequency transmission line has a length in the transmission direction that is equal to ¼ of a guide wavelength of a signal transmitted through the high frequency transmission line. 12. The high frequency resonator according to claim 10, wherein the high frequency transmission line has a length in the transmission direction that is equal to ½ of a guide wavelength of a signal transmitted through the high frequency transmission line. 13. The method according to any one of claims 10 to 1, which has a predetermined length.
2. A high frequency wave comprising: the high frequency resonator according to any one of 2), an input terminal for inputting a high frequency signal to the high frequency resonator, and an output terminal for outputting a high frequency signal from the high frequency resonator. filter. 14. A transmission line for inputting a high-frequency signal at one end and outputting the high-frequency signal at the other end, and the transmission line, which is coupled to the transmission line.
And a high-frequency resonator according to claim 5. 15. A dielectric resonator comprising: a resonator case including a conductor; and a dielectric body having a predetermined shape and mounted in the resonator case. A dielectric resonator comprising the high-frequency electrode described in 3 or 4. 16. The dielectric resonator according to claim 15, an input terminal electromagnetically coupled to the dielectric resonator for inputting a high-frequency signal to the dielectric resonator, and an electromagnetic wave connected to the dielectric resonator. High-frequency filter, which is coupled to the dielectric resonator and has an output terminal for outputting a high-frequency signal from the dielectric resonator. 17. A high frequency device comprising an electrode for performing a predetermined high frequency operation, wherein the electrode has the high frequency electrode according to claim 1, 2, 3 or 4.