JPH07255169A - Resonance-type dc-dc converter - Google Patents

Abstract

PURPOSE:To simplify the circuit constitution of a half bridge type of DC-DC converter being constituted so as to reduce the power loss, using resonance. CONSTITUTION:The series circuit composed of first and second capacitors C1 and C2 and the series circuit composed of first and second switches Q1 and Q2 are connected, respectively, between one end and the other end of a DC power source 1. The primary winding N1 of a transformer T is connected through a first inductance L4 between the mutual junction 3 of the first and second capacitors C1 and C2 and the mutual junction 2 of the first and second switches Q1 and Q2. A second inductance Lp larger in value than the first inductance Lr is connected in parallel with the primary winding N1. Series resonance is caused by the equivalent capacitor consisting of the parallel circuit of the first and second capacitors C1 and C2 and the first inductance Lr. The ON period of the first and second switches Q1 and Q2 is set longer than the half wave of the series resonance. The output voltage is adjusted by changing the ON period of the first and second switches Q1 and Q2.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance type DC-DC converter (D) comprising a switch, an output transformer, an output rectifying / smoothing circuit, a resonance inductance and a capacitor.
C-DC converter).

[0002]

2. Description of the Related Art In order to achieve zero current switching, a resonant DC-DC converter can be constructed as shown in FIG. In this circuit, the first and second FETs are formed between one end and the other end of a DC power supply 1 including a power supply circuit including a rectifier connected to an AC power supply and a smoothing capacitor (may be omitted) or a battery. Switch Q1
, Q2 are connected in series, and a series circuit of first and second capacitors C1, C2 having the same capacitance is connected. The primary winding N1 of the output transformer T is connected to the first and second switches Q via the resonance inductance Lr.
1, Q2 interconnection point 2 and first and second capacitors C
It is connected to the interconnection point 3 of C1 and C2. The secondary winding N2 of the output transformer T is formed in a center tap type, and both ends thereof are connected to one end of a smoothing capacitor Co via output rectifying diodes Da and Db, and the center tap is connected to the other end of the capacitor Co. There is. Output terminals 4 and 5 for supplying a DC voltage to a load (not shown) are connected across the capacitor Co. The control circuit 6 generates a signal for alternately turning on and off the switches Q1 and Q2. The first and second switches Q1 and Q2 are insulated gate type (MOS type) field effect transistors having a structure in which the sources are connected to the substrate, and equivalently, the first and second switches are provided between the drain and the source. Control switch S1
, S2 and first and second diodes D1 and D2 connected in anti-parallel thereto. The first and second switches Q1 and Q2 may be composed of a bipolar transistor and a diode connected in antiparallel thereto.

When the DC-DC converter of FIG. 1 is driven, the gates (control terminals) of the first and second switches Q1 and Q2 are controlled at a constant cycle as shown in FIGS. The signals (gate signals) Vg1 and Vg2 are supplied, and the first and second switches Q1 and Q2 are connected to each other by a slight dead line.
A time (pause period) is provided to alternately turn on and off at regular intervals. The ON periods of the first and second switches Q1 and Q2 correspond to the half-wave period of the resonance current. The series resonance circuit includes an inductance Lr and first and second capacitors C.
It is formed by an equivalent capacitor Cr corresponding to a parallel circuit of 1 and C2. The voltage Vds2 and current I of the first switch Q1
The d1 changes as shown in FIGS. Current I
The d1 rises as a sine wave after the first switch Q1 is turned on by the resonance action, and only half wave flows. If the first switch Q1 is turned off when this half wave ends and the current becomes zero, zero current switching is achieved,
Switching loss is reduced. The current and voltage of the second switch Q2 changes as well.

[0004]

By the way, in the DC-DC converter shown in FIG. 1, the first and second switches Q1 and Q2 are turned on in response to a half cycle of the resonance current determined by the circuit constant of Lr Cr. The time width is fixed. Therefore, the output voltage cannot be adjusted only with this circuit.

In order to solve the above problems, first and second
It is conceivable to control the time width of the dead time (pause period) between the ON periods of the switches Q1 and Q2. However, not only the circuit becomes complicated, but also the on / off frequency of the switch changes significantly. In addition, for voltage control, a method of turning off the switch before the half wave of the resonance current has finished is conceivable.However, not only does the zero current switch at the time of turn-off become impossible, but in the case of no load, the switch The on-time width becomes narrow and the operation becomes unstable. Further, a power supply circuit may require a plurality of output voltages having different levels. In this case, if each power supply circuit is provided, it becomes large and costly.

Therefore, an object of the present invention is to provide a DC / DC converter which can achieve power loss in a switch with a relatively simple circuit configuration. Another object of the present invention is to provide a DC-DC converter capable of stabilizing an output voltage without causing a large variation in the on / off repetition frequency of a switch. Also,
Still another object of the present invention is to provide a DC-DC converter capable of obtaining two outputs having different voltage levels with a relatively simple circuit configuration.

[0007]

The present invention for achieving the above object provides a series resonance circuit of a first inductance and a capacitor between a DC power supply and a primary winding of an output transformer, and the DC power supply. A resonance type DC-DC converter including a switch for turning on / off a voltage, wherein a rectifying / smoothing circuit is connected to a secondary winding of the output transformer,
Is connected in parallel to the primary winding with or without the first inductance or in parallel to a winding electromagnetically coupled to the primary winding, and A second inductance having an inductance value larger than that of the first inductance and a circuit for generating a control signal for turning on / off the switch, wherein an on-time width of the switch is the first inductance and the capacitor. Control that is set to be longer than the half-wave period of the series resonance current flowing based on the series resonance circuit of and that adjusts the output voltage of the rectifying and smoothing circuit by changing the ON / OFF cycle of the switch. And a DC-DC converter comprising a circuit. As described in claim 2, the series circuit of the first and second capacitors, the series circuit of the first and second switches, and the third and fourth series connected in parallel to the first and second switches. It is desirable to have a capacitor of Further, as described in claim 3, it is possible to obtain different output voltages from the two rectifying and smoothing circuits connected to the secondary windings by making the ON time widths of the first and second switches different from each other.

[0008]

In the invention according to claims 1 and 2, the magnitude of the stored and emitted energy of the second inductance affects the magnitude of the resonance current of the series resonance circuit. Therefore, when the ON time width of the switch is changed to adjust the stored and emitted energy of the second inductance, the output voltage changes and voltage adjustment is achieved. Further, the output voltage can be adjusted without significantly changing the on-time width of the switch, that is, the on / off frequency. According to the invention of claim 3, it is possible to easily obtain two output voltages having different voltage levels.

[0009]

[First Embodiment] Next, a resonance type DC-DC converter according to a first embodiment of the present invention will be described with reference to FIGS. However, in FIG. 3, the same parts as those in FIG. 1 are designated by the same reference numerals and the description thereof will be omitted. In the DC-DC converter of the first embodiment shown in FIG. 3, a second inductance Lp is added to the DC-DC converter of FIG. 1, and a control circuit 6a capable of voltage control is provided instead of the control circuit 6. Other than that, the configuration is the same as that of FIG. Second according to the invention
Of the transformer L is connected in parallel to the primary winding N1 of the transformer T. This second inductance Lp has a larger inductance value than the first inductance Lr connected in series with the primary winding N1. When the constants of the circuit elements are shown to be constant, C1 = C2 = 0.3 μF,
C3 = C4 = 200 pF, Lr = 5 μH, Lp = 10 μH
Is.

The control circuit 6a turns on / off the first and second switches Q1 and Q2 in accordance with the fluctuation of the output voltage or the input voltage.
It is configured to control the output voltage at a constant value by changing the off frequency. Therefore, the control circuit 6a has the output terminal 4,
Voltage detection voltage dividing resistors 7 and 8 connected between 5 and 5, a reference voltage source 9, an error amplifier (differential amplifier) 10, a light emitting diode 11, a phototransistor 12, a resistor 13,
It comprises a VCO (voltage controlled oscillator) 14 and a waveform shaping and driving circuit 15.

One input terminal of the error amplifier 10 is connected to the voltage dividing points of the voltage dividing resistors 7 and 8, and the other input terminal is connected to the reference voltage source 9. Therefore, the output voltage corresponding to the difference between the detected voltage and the reference voltage is obtained from the error amplifier 10. Since the light emitting diode 11 is connected between the output terminal of the error amplifier 10 and the ground, it emits light corresponding to the error output. The phototransistor 12 optically coupled to the light emitting diode 11 is connected via a resistor 13 between the power supply terminal indicated by + V and the ground. Therefore, when the output voltage increases and the output of the light emitting diode 11 increases, the voltage of the resistor 13 increases. Phototransistor 12
VCO14 connected to the voltage dividing point of
A frequency signal proportional to the voltage of 3 is output. VCO14
The waveform shaping and driving circuit 15 connected to the VCO 14 shapes the output of the VCO 14 into a square wave and supplies it to the control terminal (gate) of the first switch Q1 through the line 16a.
6a square wave phase inversion and a slight debt
A control signal (pulse) provided with a time (pause period) is supplied to the control terminal (gate) of the second switch Q2 via the line 16b.

What is important in this control circuit 6a is that
The first and second switches Q1, which are shorter than the time width of the half-wave of the resonance current based on the series resonance circuit of the capacitor Cr and the first inductance Lr, which are equivalent to the parallel circuit of the first and second capacitors C1 and C2, That is, the frequency of the VCO 14 is set so that the ON time width of Q2 becomes long.
The reason for this will be described later in detail.

[0013]

[Operation] Next, the operation of the circuit of FIG. 3 will be described with reference to FIGS. 4 and 6 showing waveforms of respective portions and FIG. 5 showing a part of FIG. 3 extracted. The left half of FIG. 4 schematically shows the state of each part of FIG. 3 at full load, and the right half schematically shows the state of each part of FIG. 3 when the load is lighter than this. It should be noted that although there is a slight dead time between the ON control pulses of the first and second switches Q1 and Q2, they are omitted. Now, at time t0 in FIG. 4, a control signal for turning on the first switch Q1 is generated as shown in FIG.
Immediately after this, when the drain-source voltage of the first switch Q1 becomes zero at t1, the first capacitor C1 and the first switch C1
Of the switch Q1, the first inductance Lr, and the primary winding N1 form a closed circuit. As described in FIG. 1, the second capacitor C2 is connected to the first capacitor C1 by the power source 1
Since it is connected in parallel through the capacitor, the equivalent capacitor of the parallel circuit of the first and second capacitors C1 and C2 is Cr.
Then, as in the case of FIG. 1, a series resonant circuit of Lr and Cr is formed, and the resonant current Ir thereby flows in a sinusoidal waveform as shown by the dotted line in FIG. 6 (B). The resonance current Ir is t
After reaching zero at 3, output rectifier diodes Da and Db
As a result, the capacitor C0 is separated from the secondary winding N2 of the transformer T, so that the AC primary winding N1
Becomes an infinite impedance, and the negative half-wave resonance current does not flow in the period t0 to t4. On the other hand, the current Ip based on the energy storage and release of the second inductance Lp
Flows as shown by the solid line in FIG. The stored energy of the second inductance Lp is released at time t0, and the current Ip flows upward during the period t0 to t2 in FIG. That is, in the period from t0 to t3, the current Ip is equal to the second inductance Lp and the first inductance Lr.
And the first switch Q1, the power supply 1 and the second capacitor C2
Current flows in a closed circuit consisting of and. As a result, the second capacitor C2 is reversely charged, and this voltage Vc2 is shown in FIG.
As shown in (C), it becomes lower than E / 2 in the period from t0 to t2. When the release of the stored energy of the second inductance L2 ends at time t2, the power source 1 and the first switch Q
A current flows in a closed circuit composed of 1 and the first and second inductances Lr and Lp and the second capacitor C2, energy of the second inductance Lp is stored, and the second capacitor C2 is charged. Then, the voltage Vc2 of the second capacitor C2 rises. Since the voltage of the second capacitor C2 is lower than E / 2 in the period from t0 to t4 of FIG. 6, the voltage Vc1 of the first capacitor C1 is obtained by subtracting the voltage Vc2 of the second capacitor C2 from the power supply voltage E. The value becomes E-Vc2. Therefore, in the period from t0 to t4, the first
The voltage Vc1 of the capacitor C1 is increased.
Therefore, the first capacitor C1, the first switch Q1, and the first inductance Lr within the period t0 to t4.
The operation of the resonance circuit including the primary winding N1 and the primary winding N1 is the same as that when the power supply voltage E is increased, and a resonance current Ir having a large amplitude can be passed. Although the second capacitor C2 contributes to the resonance during the period t0 to t4 to some extent,
Since its contribution is lower than that of the first capacitor C1, the resonance capacitance Cr can be regarded as only the first capacitor C1.

The current Id1 flowing through the first switch Q1 in the period from t0 to t4 is the sum of Ip of the second inductance Lp and Lr Cr resonance current Ir shown in FIG. 6 (B), and FIG. 4 (D). Flow as shown in. The third capacitor C3 for voltage partial resonance is charged to the power supply voltage E before t0. If the first switch Q1 is turned on, the third switch
The energy stored in the capacitor C3 of the first switch Q
Flowing to 1 causes power loss. In the circuit of FIG. 3, t0
During the period from t1 to t1, the current flowing in the direction in which the third capacitor C3 is reversely charged flows in the circuit composed of the second inductance Lp, the first inductance Lr, the third capacitor C3 and the first capacitor C1. The energy stored in the capacitor C3 is fed back to the first capacitor C1 and no power loss occurs. Further, the current Ids1 of the first switch Q1 starts to flow after the voltage Vds1 of the first switch Q1 becomes zero, so that zero voltage switching is achieved and the switching loss at turn-on is small. When the first switch Q1 is turned off, this voltage Vds1 becomes the third voltage Vds1.
Since it rises gently due to the action of the capacitor C3,
The crossing area between the current Id1 and the voltage Vds1 is reduced, and the switching loss is reduced. In addition, high frequency noise is suppressed.

In the section t4 to t6 of FIGS. 4 and 6, the same operation as in the section t0 to t4 occurs in the second switch Q2. That is, immediately after t4, the second switch Q
When the voltage Vds2 of 2 becomes zero, capacitors Cr and 1 equivalent to the parallel circuit of the first and second capacitors C1 and C2 are formed.
In the closed circuit of the secondary winding N1, the first inductance Lr, and the second switch Q2, a sinusoidal resonance current Ir as shown by the dotted line in FIG. 6B flows. Further, during the period from t4 to t5, the current Ip flows through the closed circuit of the second inductance Lp, the second capacitor C2 and the second switch Q2 based on the release of the stored energy of the second inductance Lp, During the period from t5 to t6, the second capacitor C2 is used as a power source, the second capacitor C2, the second inductance Lp, the first inductance Lr, and the second switch Q2.
A current Ip flows in a closed circuit composed of and. As a result, the voltages of the first and second capacitors C1 and C2 are shown in FIG.
It changes as shown in the section from t4 to t6. The current Id2 of the second switch Q2 is the sum of the current Ip of the second inductance Lp and the resonance current Ir. Second switch Q2
At the time of turn-on and turn-off, the same operation as that of the first switch Q1 occurs, so that the power loss is reduced.

By the way, in the circuit of FIG. 3, when the magnitude of the load connected between the output terminals 4 and 5 changes, the amplitude of the resonance current Ir is shown by the dotted line in FIGS. 4 (D) and 6 (B). To change. That is, for example, the impedance of the load increases, and the AC impedance from the primary winding N1 in FIG. 3 to the load increases for a light load, and the maximum amplitude of the resonance current Ir decreases. If the maximum amplitude of the resonance current Ir decreases, the energy supplied to the output smoothing capacitor Co and the load also decreases, and the output voltage fluctuation due to the load fluctuation can be automatically compensated. However, the compensation of the output voltage fluctuation based on the voltage fluctuation of the power source 1 cannot be achieved by the amplitude change of the resonance current Ir.

Therefore, in this embodiment, the output voltage is controlled by the fluctuation of the power supply voltage E as follows. The error amplifier 10 generates an output corresponding to the difference between the detected output voltage and the reference voltage, and the light emitting diode 11 emits light corresponding to the error output. Therefore, when the output voltage becomes higher than the desired value, the light emission intensity of the light emitting diode 11 becomes higher than the reference value. As a result, the resistance of the photocoupler 12 optically coupled to the light emitting diode 11 decreases, and the resistance 13
Is higher than ever, and the output frequency of the VCO 14 is higher than ever. As a result, t7 in the right half of FIG.
The first and second switches Q1,
The ON / OFF cycle T2 of Q2 becomes shorter than the left half T1. Therefore, the first and second capacitors C based on the energy storage and release of the second inductance Lp.
The fluctuation range of the voltages Vc1 and Vc2 of 1 and C2 becomes small as shown by the dotted line in FIG. 6 (C). Considering the on period t0 to t4 of the first switch Q1, the fact that the voltage Vc1 of the first capacitor C1 changes from the solid line to the dotted line corresponds to the reduction of the power supply voltage of the resonance circuit, and the resonance. The maximum amplitude of the current Ir drops. Therefore, the amount of power supplied to the output smoothing capacitor Co is reduced, and the output voltage is reduced toward a desired value. When the output voltage becomes lower than the desired value, the operation opposite to the above occurs. In this embodiment, only the adjustment corresponding to the voltage fluctuation of the power supply 1 is performed, so that the ON / OFF frequencies of the first and second switches Q1 and Q2 do not significantly change. Therefore, it is possible to stably operate from no load to full load. Also, the first and second capacitors C1,
Since C2 is used not only for dividing the power supply voltage but also as a series resonance capacitor, the circuit configuration is simplified.

[0018]

[Second Embodiment] Next, a DC-DC converter of a second embodiment will be described with reference to FIG. However, in FIG. 7, the same parts as those in FIG. 3 are designated by the same reference numerals and the description thereof will be omitted. The DC-DC converter circuit of FIG. 7 uses the first inductance Lr for resonance as the second inductance Lr.
It corresponds to the one transferred between p and the primary winding N1 and is otherwise formed in the same manner as in FIG. However, in this embodiment, the primary and secondary windings N1 and N2 and the windings for the first and second inductances Lr and Lp are provided in the common core in order to achieve downsizing and cost reduction. It is wound.
Since the other points are the same as those in the first embodiment, the same operational effects can be obtained.

[0019]

[Third Embodiment] A DC-DC converter according to a third embodiment of the present invention will be described below with reference to FIGS. However, in FIGS. 8 and 9, the same parts as those in FIGS. 3 to 7 are designated by the same reference numerals, and the description thereof will be omitted. Figure 8
The circuit of FIG. 7 shows the first inductance Lr of the circuit of FIG.
And the first smoothing capacitor Ca is connected between the upper terminal of the secondary winding N2 and the center tap via the first rectifying diode Da, and the lower terminal of the secondary winding N2 is connected. The second smoothing capacitor Cb is connected between the center tap and the center tap through the second rectifying diode Db, and the first and second output terminals 4a are vertically arranged around the terminal 5 connected to the center tap. 4b, and the control circuit 6b to FIG.
As shown in (A) and (B), different on-time widths Ta
, Tb pulses to the first and second switches Q1, Q2
It is configured to be applied to the gate (control terminal) of. The circuit of FIG. 8 has the same configuration as that of FIG. 3 except for the above. The upper half N2a and the lower half N2b of the secondary winding N2 are set to have the same number of turns.

In the circuit of FIG. 8, when the ON time width Ta of the first switch Q1 is made shorter than the ON time width Tb of the second switch Q2, the second switch is turned on during the ON period of the first switch Q1 from t0 to t1. The amount of reverse charging of the second capacitor C2 due to the upward release of energy of the inductance Lp is reduced, and the increment of the voltage Vc1 of the first capacitor C1 is reduced. On the contrary, the amount of reverse charge of the first capacitor C1 based on the stored energy of the second inductance Lp during the ON period of the second switch Q2 from t1 to t2 increases, and the voltage Vc2 of the second capacitor C2 increases. Is greater than the rate of increase of the voltage Vc1 of the first capacitor C1 from t0 to t1. First and second capacitors C
Since the voltages Vc1 and Vc2 of C1 and C2 act in proportion to the maximum amplitude of the resonance current Ir, the period t0 to t1 and the period t1 to t1.
Different power is supplied to the secondary side of the transformer T during the period t2. The power during the ON period of the first switch Q1 is supplied to the first smoothing capacitor Co1 and the first output terminal 4a via the first rectifying diode Da, and the power during the ON period of the second switch Q2 is 2 rectifier diode D
Since it is supplied to the second smoothing capacitor Co2 and the second output terminal 4b via b, the first output voltage Vo1 between the intermediate terminal 5 and the first output terminal 4a and the intermediate terminal 5 Two
The second output voltage Vo2 between the output terminal 4b and the second output voltage 4b has a different value, and two output voltages Vo1 and Vo2 having different voltage values can be easily obtained by a simple circuit configuration. The effect of reducing the switching loss of the first and second switches Q1 and Q2 and other effects can be obtained as in the first and second embodiments.

[0021]

[Fourth Embodiment] Next, the D of the fourth embodiment shown in FIG.
The C-DC converter will be described. However, in FIG. 10, the same parts as those in FIG. 3 are designated by the same reference numerals and the description thereof will be omitted. The circuit of FIG. 10 omits the center tap of the secondary winding N2 of the circuit of FIG. 3, connects the lower side terminal of the secondary winding N2 to the intermediate output terminal 5, and connects the upper side terminal and the lower side of the secondary winding N2. First and second smoothing capacitors Co1 and Co2 are connected to the terminals through diodes Da and Db having different directions, and the first and second output terminals 4 are connected to the capacitors Co1 and Co2.
a and 4b are connected. As a result, the first smoothing capacitor Co1 is charged with the current passing through the first rectifying diode Da, and the second smoothing capacitor Co2 is charged with the current passing through the second rectifying diode Db. Therefore,
It is possible to easily obtain the positive first output voltage Vo1 and the negative second output voltage Vo2. The circuit shown in FIG. 10 also has the same effects as the first embodiment. The control circuit 6a in FIG. 10 is replaced with the control circuit 6b shown in FIG.
It is also possible to make the ON time widths of the first and second switches Q1 and Q2 different. In this case, output voltages having different positive and negative values can be obtained.

[0022]

[Fifth Embodiment] Next, a DC-DC converter of a fifth embodiment will be described with reference to FIG. However, in FIG. 11, the same parts as those in FIG. 3 are designated by the same reference numerals and the description thereof will be omitted. The circuit of FIG. 11 has a first inductance L.
A dedicated capacitor Cr for forming a series resonance circuit together with r is connected in series to the first inductance Lr, and the others are configured similarly to FIG. Figure 1
Since the operation of the circuit of No. 1 is substantially the same as that of the circuit of FIG. 3, it is possible to obtain the same effect as the circuit of FIG.

[0023]

[Sixth Embodiment] Next, a DC-DC converter of a sixth embodiment will be described with reference to FIG. However, in FIG. 12, the same parts as those in FIG. 3 are designated by the same reference numerals and the description thereof will be omitted. The circuit of FIG. 12 is obtained by connecting a series resonance circuit of a resonance capacitor Cr and a first inductance Lr in parallel to a second switch Q2 via a primary winding N1, and otherwise the same as FIG. It is configured. 12
Since the operation of the circuit of is substantially the same as that of the circuit of FIG.
It is possible to obtain the same effect as that of the circuit of FIG.

[0024]

MODIFICATION The present invention is not limited to the above-mentioned embodiments, and the following modifications are possible. (1) The second inductance Lp can be connected in parallel with the secondary winding N2. Further, the second inductance Lp can be connected in parallel to the upper half and the lower half of the secondary winding N2 having the center tap. Further, the transformer T may be provided with a tertiary winding, and the second inductance Lp may be connected in parallel therewith. (2) The first inductance Lr and the primary winding N1 of the transformer T are wound on the same core, and the second inductance Lp and the primary winding N1 of the transformer T are wound on the same core. be able to. Further, the inductance component of the primary winding N1 can also be used as the first inductance Lr. Further, an intermediate tap may be provided on the first inductance Lr, and the upper end of the second inductance Lp may be connected to the intermediate tap. (3) In the circuits of FIGS. 8 and 10, a dedicated resonance capacitor Cr can be connected at the same position as in FIG. Further, the third and fourth capacitors C3 and C4 can also be used as capacitors for series resonance with the first inductance Lr. Also, the capacitors Cr and C
Any combination of 1 to C4 can be used for the first inductance L
It may be a capacitor for series resonance circuit with r. Also,
The stray capacitance between the drain and source of the first and second switches Q1 and Q2 can be used as the resonance capacitor Cr. (4) Instead of detecting the output voltage for voltage control by the control circuit 6a, the voltage of the power source 1 or the capacitors C1 and C2 may be detected and a control pulse may be formed to bring this to a desired value. (5) The power supply 1 is a full-wave rectifier connected to a sinusoidal AC power supply, and an unsmoothed pulsating current is input to obtain a power factor improving effect. (6) In each embodiment, the first and second switches Q
When 1 and Q2 are semiconductor switches having substantially no carrier storage effect, the first and second switches Q1 and Q2 are alternately turned on and off without providing a dead time between them. Can be controlled. (7) In each of the embodiments, the dead time width may be controlled according to the output voltage.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a conventional DC-DC converter.

FIG. 2 is a waveform diagram schematically showing a state of each part of the circuit of FIG.

FIG. 3 is a circuit diagram showing a DC-DC converter of a first embodiment of the present invention.

FIG. 4 is a waveform diagram schematically showing a state of each part of FIG.

5 is a circuit diagram equivalently showing a part of FIG. 3. FIG.

FIG. 6 is a waveform diagram schematically showing a state of each part of FIGS. 3 and 5.

FIG. 7 is a circuit diagram showing a DC-DC converter of a second embodiment.

FIG. 8 is a circuit diagram showing a DC-DC converter of a third embodiment.

9 is a waveform diagram schematically showing a state of each part of FIG.

FIG. 10 is a circuit diagram showing a DC-DC converter of a fourth embodiment.

FIG. 11 is a circuit diagram showing a DC-DC converter of a fifth embodiment.

FIG. 12 is a circuit diagram showing a DC-DC converter of a sixth embodiment.

[Explanation of symbols]

 Lr first inductance Lp second inductance

Claims (3)

[Claims] 1. An output transformer including a series resonance circuit of a first inductance and a capacitor between a DC power supply and a primary winding of the output transformer, and a switch for turning on / off the voltage of the DC power supply. In a resonance type DC-DC converter having a configuration in which a smoothing rectification circuit is connected to the secondary winding of the above, in parallel to the primary winding with or without the first inductance, or A second inductance connected in parallel to a winding electromagnetically coupled to the next winding and having an inductance value larger than the first inductance; and a control signal for turning on / off the switch. For generating the ON time width of the switch.
Is set longer than the half-wave period of the series resonance current flowing based on the series resonance circuit of the inductance and the capacitor, and the output voltage of the rectifying and smoothing circuit is adjusted by changing the ON / OFF cycle of the switch. And a control circuit configured as described above. 2. A series circuit of first and second capacitors connected between one end and the other end of a DC power supply, and first and second series circuits connected between one end and the other end of the DC power supply. A transformer connected in series via a first inductor between a series circuit of two switches and an interconnection midpoint of the first and second capacitors and an interconnection midpoint of the first and second switches. Rectifying and smoothing circuit connected to the primary winding of the transformer and the secondary winding of the transformer, third and fourth capacitors connected in parallel to the first and second switches, and the primary winding Is connected in parallel with or without the first inductance, or in parallel with a winding electromagnetically coupled to the primary winding, and is larger than the first inductance. A second inductance having an inductance value A circuit for generating a control signal for alternately turning on the first and second switches, wherein an on-time width of the first and second switches is actually equal to the first inductance. Is set longer than the half-wave period of the series resonance current flowing based on a series resonance circuit with a capacitor connected in series or equivalently, and the ON / OFF cycle of the first and second switches is set. And a control circuit configured to adjust the output voltage of the rectifying and smoothing circuit instead. 3. A first and a second capacitor connected between one end and the other end of the DC power supply, and a first and a second switch connected between one end and the other end of the DC power supply. And a primary winding of a transformer connected via a first inductance between an interconnection midpoint of the first and second capacitors and an interconnection midpoint of the first and second switches. A first rectifying and smoothing circuit connected to the first secondary winding of the transformer; a second rectifying and smoothing circuit connected to the second secondary winding of the transformer; the first and second A third and a fourth capacitor connected in parallel to the switch, and in parallel with the primary winding with or without the first inductance, or with respect to the primary winding. Is connected in parallel to the coupled windings and is connected to the first inductance A circuit for generating a control signal for alternately turning on the first and second switches, and one or a plurality of second inductances having a large inductance value. A series resonance current flowing based on a series resonance circuit with a capacitor that is actually or equivalently connected in series to the first inductance while the ON time widths of the two switches are set to be different from each other. A DC-DC converter comprising: a control circuit that is set to be longer than each half-wave period.