JPH07234270A - Receiving device for complex acoustic signal - Google Patents

Abstract

PURPOSE:To provide a receiving device of complex acoustic signals which provides a high dynamic range and excellent S/N by receiving broad-band acoustic signals in addition to normal acoustic signals. CONSTITUTION:In addition to three receivers 1, 2, 3 and AGC amplifiers 49, 50, 51 connected to the respective receivers, a nondirectional broad-band receiver 42, an AGC amplifier 43 connected to the receiver 42, and a band converter 45 are provided. Of them, the AGC amplifiers 50 and 51 are subjected to gain control based on the output signals of the nondirectional AGC amplifier 49. The band converter 45 converts the frequencies of broad-band acoustic signals into a higher frequency band than that of a complex signal consisting of three acoustic signals, and an adder 47 combines the broad-band complex signals into one broad-band complex signal, which is then transmitted from a transmitter 26.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method of transmitting an acoustic signal obtained from a small acoustic detector located at a remote place,
In particular, a directional acoustic signal obtained from a figure-8 directional receiver, and a received wave that is modulated by combining a compass signal indicating the direction on the magnetic azimuth axis of the acoustic wave sensor and an omnidirectional broadband acoustic signal. Regarding the device.

[0002]

2. Description of the Related Art Conventionally, as a device of this type, Japanese Patent Publication No.
-22038 was disclosed. About this, Fig. 10
~ It will be described with reference to FIG.

FIG. 10 is a block diagram showing a conventional transmitter. In the figure, this transmitter is composed of three receivers 1,
It has 2 and 3. The wave receiver 1 is omnidirectional, but the other wave receivers 2 and 3 both have directivity, and the directivities of both are orthogonal. As a specific configuration of these directional wave receivers 2 and 3, for example, that described in Japanese Patent Publication No. 58-51478 can be used. When these three receivers capture a sound wave, a weak electrical signal is generated based on the sound wave, and the receiver receives the amplified electric signal and inputs it to the corresponding amplifiers 4, 5 and 6, respectively, for amplification. .

Here, the three wave receivers 1 to 3 provided for direction finding are acoustically placed at the same position. Let A (t) be the acoustic signal at the output point 7 of the omnidirectional amplifier 4 as a function of time t, and let θ be the incident angle of the sound wave measured from the reference points of these receivers 1 to 3, and let When the directivity is cos θ and the directivity of the receiver 3 is sin θ, the acoustic signal at the output point 8 of the cosine amplifier 5 is represented by A (t) cos θ, and the acoustic signal at the output point 9 of the sine amplifier 6 is represented by Signal is A (t) sin
It is represented by θ.

The above-mentioned three receivers 1 to 3 and the three amplifiers 4 to 6 constitute a direction finder, and as will be described later, the reference points of the three receivers 1 to 3 are used. The information θ that defines the measured relative direction of the sound wave source is configured to be taken out by the receiver. Phase shifter 11, two modulators 12
And 13 together form a balanced modulator.

The subcarrier generator 10 generates a subcarrier cos ωt having an angular frequency ω, and the generated subcarrier co
s ωt is a phase shifter 11, a modulator 13 and a frequency divider 16 described later.
Given to. The angular frequency ω of the subcarrier is selected to be at least twice the maximum angular frequency of the acoustic signal A (t). The phase shifter 11 delays the phase of the subcarrier cos ωt by (π / 2), and is − (π / 2) phase shift subcarrier cos (ωt-
π / 2) is given to the modulator 12. The modulator 12 receives the acoustic signal A (t) cos θ from the cosine amplifier 5 and
12 is the acoustic signal A (t) cos θ- (π / 2) phase-shifted subcarrier cos
Modulate with (ωt-π / 2). On the other hand, the acoustic signal A (t) sin θ is given from the sine amplifier 6 to the modulator 13, and the modulator 13 modulates the acoustic signal A (t) sin θ with the subcarrier cos ωt. Therefore, at the output terminals 14 and 15 of the modulators 12 and 13, signals A (t) cos θcos (ωt-π / 2), A (t) sin θcosω, which are the product of the acoustic signal and the subcarrier, respectively.
yield t.

The frequency divider 16 calculates the frequency of the subcarrier cos ωt.
Divide into 1/2 and drive compass 17. This compass 17
Is, for example, the drive signal 2
It is of the flux gate type, in which the double frequency phase represents the azimuth angle of the magnetic field. The compass 17 is formed integrally with the above-mentioned wave receivers 1 to 3, and the output of the compass 17 represents the angle φ on the magnetic azimuth axis of the angle reference points of the directional wave receivers 2 and 3. Therefore, the signal cos (ωt-φ) is generated at the compass output point 18. Hereinafter, in this specification, this signal cos (ωt-φ) is referred to as a phase pilot signal. The angle φ is information necessary for obtaining the absolute direction θ + φ from the relative direction θ of the sound wave generation source.

Since the output of the compass 17 contains many high frequency components, a selection filter (not shown) is inserted in the circuit and the phase angle φ of the phase pilot signal is set to the directional receiver 2 and An adjustment function (not shown) is also provided for the purpose of accurately matching the reference point of the angle of 3. The output point 19 is also provided in the frequency divider 16 and the signal co obtained by dividing the frequency of the subcarrier cos ωt in half is used.
s (ωt / 2-γ) is also output. Here, γ is an indefinite phase angle due to frequency division. This signal cos (ωt / 2-γ) is called a frequency pilot signal.

The mixer 20 has an output point of the omnidirectional amplifier 1.
7. The signals at the output points 14 and 15 of the modulators 12 and 13, the output point 18 of the compass 17, and the output point 19 of the frequency divider 16 are input, and the mixer 20 adds these signals. Therefore,
The composite signal C 1 expressed by the following equation (1) is output to the output terminal 21 of the mixer 20.

C = A (t) + A (t) cos θcos (ωt-π / 2) + A (t) s
in θcos ωt + cos (ωt- φ) + cos (ωt / 2-γ) = A
(t) + A (t) cos (ωt-π / 2 + θ) + cos (ωt-φ) + cos
(Ωt / 2-γ) (1)

The frequency pilot signal cos (ωt / 2-γ) in the composite signal C is the phase pilot signal cos (ωt-φ) on the demodulator side when the frequency of the target sound is an extremely low frequency. ) Is a double sideband signal A (t) cos (ωt-
This is added in order to facilitate the design of the demodulator in view of the difficulty in extracting from between (π / 2 + θ).

FIG. 12 shows the frequency array of the composite signal C expressed by the above equation (1). In the figure, 22
Is an omnidirectional acoustic signal A (t), 23 is a directional acoustic signal A (t) cos (ωt-π / 2 + θ) modulated by a subcarrier, 24 is a phase pilot signal cos (ωt-φ), Reference numeral 25 represents a frequency pilot signal (ωt / 2-γ).

Reference numeral 26 in FIG. 10 denotes a transmitter, which constitutes the final stage of the transmitting device and transmits the above-mentioned composite signal to the central receiving device (not shown) via the antenna by the main carrier in the ultra-high frequency band. For the modulation on the main carrier, FM modulation is used to obtain a low distortion and a high SN ratio, and in particular, the frequency deviation of this FM modulation secures a sufficiently large value for normal audio signal levels. And, the SN ratio is raised. Usually, the frequency deviation is set to be several times as large as the highest frequency of the composite signal.

A plurality of main carrier frequencies, that is, channel frequencies are assigned to a transmission device which is a small acoustic detector in an ultra high frequency band to be transmitted, and a reception device selects and receives this channel. it can. The central receiver usually has multiple receivers,
Each can simultaneously receive radio signals on different channels. Therefore, it is possible to simultaneously receive the acoustic signals from the detectors located in different remote areas and having different channels. Note that these central receiving devices are used by many users.

FIG. 11 is a block diagram of a conventional receiver shown in Japanese Patent Publication No. 53-22038. In the figure, 27 is a receiver, which constitutes the first stage of the receiving device. This receiver 27 receives the radio signal by the main carrier in the ultra-high frequency band,
This is FM demodulated and the composite signal C is extracted and output terminal
Output from 28. The receiver 27 has a channel selection function, and by selecting an arbitrary channel, a detector located in a different remote place can be selected.

As is well known, the FM signal has a frequency band that is twice the sum of the frequency shift and the highest modulation frequency. Therefore, the frequency of each channel is set at a value that is twice the sum of the maximum allowable frequency deviation and the maximum modulation frequency, plus the frequency of the attenuation characteristic of the filter separating the channels, etc. Has been done. Also, the maximum frequency deviation is FM to prevent the frequency band from expanding due to an excessive modulation signal and interfering with adjacent channels.
This value is defined for the modulation function, and is the transmitter 26 of FIG.
This is realized by a level limiter (not shown) in the input section of the.

In FIG. 11, 29 is a phase locked loop circuit,
Phase pilot signal cos (ωt-φ) from the composite signal C
Subcarrier cos (ωt-φ) with the same frequency and phase as
To create a. Therefore, the output terminal 30 of the phase locked loop 29 becomes cos (ωt-φ), which is the balanced modulator 31.
It is a carrier wave input. The output of the phase synchronization circuit 29 is
At the same time, it is also supplied to the phase shifter 32, and the phase shifter 32 performs an operation of delaying the phase of the signal of the angular frequency ω by (π / 2). Therefore, the output of the phase shifter 32 is cos (ωt-φ-π / 2)
And this becomes the carrier input of the balanced modulator 33. Therefore, in FIG. 11, the output terminals 34 and 35 of the two balanced modulators 33 and 31 are represented by the following equations (2) and (3), respectively.

A (t) cos (ωt-φ-π / 2) + A (t) {cos (2
ωt-π + θ- φ) + cos (θ + φ)} / 2+ {cos (2 ωt-2
φ-π / 2)} / 2 (2)

A (t) cos (ωt-φ) + A (t) {cos (2 ωt-π
/ 2 + θ- φ) + sin (θ + φ)} / 2+ {cos (2 ωt-2 φ) +
1} / 2 ・ ・ ・ (3)

Reference numerals 36, 37, and 38 in FIG. 11 denote the same bandpass filters that pass only the frequency band of the acoustic signal A (t), and the time delay of the acoustic signal A (t) due to this is A (t ) → A '
Expressed as (t). The input of bandpass filter 36 is the composite signal C and the signal appearing at its output 39 is A '(t).
Becomes

According to the above equation (2), what can pass through the bandpass filter 37 is the second half component of the second term, that is,
Since A (t) {cos (θ + φ)} / 2, the signal appearing at the output terminal 40 of the bandpass filter 37 is A '(t) {cos (θ
+ φ)} / 2. Similarly, according to Eq. (3), the bandpass filter 38 can pass the latter half of the second term,
That is, since A (t) {sin (θ + φ)} / 2, the signal appearing at the output terminal 41 of the bandpass filter 38 is A '(t)
It becomes {sin (θ + φ)} / 2. Therefore, output terminals 39-41
Respectively, the omnidirectional acoustic signal A '(t) and the maximum sensitivity directions on the geomagnetic azimuth axis were north-south and east-west, respectively.
Figure 8 directional acoustic signal A '(t) cos (θ + φ), A' (t) sin
(Θ + φ) is output. The azimuth measurement using these signals can be performed, for example, using the configuration described in Japanese Patent Publication No. 55-28514.

[0022]

In the transmitting device having the above-mentioned configuration, the maximum frequency of the modulated signal for the main carrier, that is, the composite signal is 3 times or more the maximum frequency of the target acoustic signal from the frequency array of FIG. Become.

On the other hand, from the viewpoint of acoustic signal detection performance, it is required to transmit an acoustic signal having a frequency as high as possible, for example, 10 times or more that of the conventional frequency. In order to meet such demands, if the acoustic signal frequency is increased with the conventional configuration, many existing receivers and demodulators need to be extensively modified. In addition, since the occupied frequency bandwidth becomes wide, it is difficult to secure a sufficient number of channels in the allocated ultra-high frequency band.

In order to solve the above problems, in the present invention, a wideband omnidirectional acoustic signal having a maximum frequency of 10 times or more of the conventional acoustic signal is transmitted in addition to the conventional acoustic signal. , It is intended to be able to receive a conventional directional acoustic signal and a wideband omnidirectional acoustic signal at a high SN ratio without adding a large-scale modification to many existing receivers and demodulators.

[0025]

In order to achieve the above object, in the present invention, an omnidirectional receiver, a cosine directional receiver having directivity orthogonal to each other, and a sine directional receiver. And three amplifiers for amplifying the signals of these three receivers and outputting them to the balanced modulation configuration unit including the frequency pilot signal, respectively, and
Wideband receivers for frequencies up to frequencies higher than a single signal, wideband amplifiers for amplifying the signals, band converters for frequency-converting the amplified wideband signals, and receivers, amplifiers, and band converters A composite acoustic signal receiving device that frequency-divides the output signal of the balanced modulation component and the output signal of the band converter and mixes them to obtain a wideband composite signal.In addition to the above three amplifiers, as a wideband amplifier While using an amplifier with automatic gain control function,
From the output signal of the amplifier with the automatic gain control function for amplifying the signal of the omnidirectional receiver, of the amplifier with the automatic gain control function and the sine directional receiver for amplifying the signal of the cosine directional receiver A gain controller that forms a gain control signal for both the amplifier with the automatic gain control function that amplifies a signal is provided, and the band converter further maintains the original frequency spectrum shape of the wideband signal. The frequency is converted to a frequency band higher than the frequency band of the balanced modulation component, and the band converter is a 90-degree phase difference wave detector, two multipliers with subcarriers having different 90-degree phases, and an adder. It is configured to cancel the lower sideband or the upper sideband, and the frequency of the subcarrier used for the band conversion is an integral multiple of the frequency of the frequency pilot signal of the balanced modulation configuration unit. Those were.

[0026]

With the above construction, in the present invention, the wide band acoustic signal is converted into a high frequency band signal while maintaining the original frequency spectrum shape, and is higher than the conventional composite signal by an integral multiple of the frequency of the frequency pilot signal. An amplifier with an automatic gain control function is used as an amplifier for arranging in the frequency band, FM-modulating and transmitting, and amplifying the captured signal from each receiver, and the captured signal from the omnidirectional receiver is used. A gain control signal for an amplifier with automatic gain control function for cosine and sine directional acoustic signals is generated from an output signal of the amplifier with automatic gain control function for amplifying.

[0027]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS One of the preferred embodiments of the present invention will be described below with reference to the drawings.

FIG. 1 is a block diagram showing the configuration of a transmitting apparatus according to an embodiment of the present invention. In the drawings, the same constituents as those described in the section of the related art are given the same reference numerals as those in the corresponding drawings, and the description thereof will be omitted. In Figure 1, Figure 10
Compared with the conventional example, the omnidirectional broadband receiver 42 is added. The broadband receiver 42 captures a high-frequency sound wave having a frequency 10 times or more the conventional acoustic signal frequency, and outputs an electric signal corresponding to the captured sound wave. Broadband receiver 42
The electric signal output from the amplifier is input to and amplified by a wide band AGC amplifier 43 having an automatic gain control (hereinafter abbreviated as AGC) function. 44 is a wideband gain controller, which is a wideband AGC
The output level of the amplifier 43 is detected and integrated to output a gain control signal. Wideband based on this gain control signal
The AGC amplifier 43 keeps its output level constant. Reference numeral 45 denotes a band converter, which converts the frequency of the wideband acoustic signal into a frequency band higher than the maximum frequency of the conventional composite signal shown in FIG.

Reference numeral 46 is an output terminal of the band converter 45. 47 is a mixer similar to the mixer 20 shown in FIG. However,
Compared to the conventional mixer 20, the mixer 47 has
One has been added. The output signal from the output terminal 48 of the mixer 47 is input to the transmitter 26 as in FIG.

Further, as is clear from the comparison of FIG. 1 and FIG. 10, in this embodiment, each of the receivers 1, 2,
AGC amplifiers 49, 50, and 51 having an AGC function are used as amplifiers for amplifying the omnidirectional acoustic signal, the cosine directional acoustic signal, and the sine directional acoustic signal from 3 and 3. Therefore, the gain controller 52 is added. The gain controller 52 is an AG for omnidirectional audio signals.
Based on the acoustic signal level output from the C amplifier 49,
Generates gain control signals for the three AGC amplifiers 49-51.

In the following description, the above AGC amplifiers 49, 50 and 51 will be replaced by an omni AGC amplifier and a cosine AGC amplifier, respectively.
Amplifier, called sine AGC amplifier. As described above, the configuration requirements except for the configuration in which the wideband acoustic signal from the wideband receiver 42 is amplified, the band is converted and added, and the configuration in which the weak electric signals from the receivers 1 to 3 are amplified are Also, the transmitting apparatus of this embodiment may have the same configuration as the conventional one.

The operation of the characteristic part of this embodiment, which is composed of the wide band receiver 42, the wide band AGC amplifier 43, the wide band gain controller 44 and the band converter 45, will be described below. The output of the wide band AGC amplifier 43 is supplied to the band converter 45 of the next stage and also to the wide band gain controller 44. After being detected and integrated in the wide band gain controller 44, the output of the wide band AGC amplifier 43 is obtained. Return to 43 and configure the AGC loop. With this loop, the output level of the wide band AGC amplifier 43 is always kept constant. Wideband AGC amplifier 43 and omni AGC amplifier 49, cosine AGC amplifier described above
As the sine AGC amplifier 51 and the sine AGC amplifier 51, it is possible to use one having a similar configuration.

FIG. 2 shows a wide band AGC amplifier 43 and an omni AGC.
The configuration of an amplifier, a cosine AGC amplifier, and an AGC amplifier 53 used for a sine AGC amplifier is shown. Also, in FIG.
A characteristic curve of the voltage controlled resistance element 55 is shown in FIG. this
The AGC amplifier 53 is, for example, a non-inverting amplifier, and has an input resistance
At the input stage of the fixed amplification component consisting of operational amplifier 54 having R ₂ and feedback resistor R _{3, there} is a resistor R _{1 for} dividing the acoustic signal input (voltage input) and a voltage controlled resistance element 55.
Is provided. As shown in Fig. 3, the voltage-controlled resistance element 55 has a control input voltage (gain control signal) from the gain controller 44 that is a certain value or less, and the resistance value rapidly decreases from a very large value to a certain degree. When the resistance value decreases, an element having a characteristic in which the resistance value gradually decreases thereafter as the control input voltage increases is used. An example of such an element is, for example, a field effect transistor (FET).

Since the acoustic signal input is divided by the voltage control type resistance element 55 and the resistance element R1 and input to the amplification stage, the control input voltage is small and the resistance value of the voltage control type resistance element 55 is very small. When it is large, it is amplified with a large and almost fixed gain as a whole, and when the control input voltage is large and the resistance value of the voltage controlled resistance element 55 is small, it is amplified with a small gain as a whole. That is, the voltage-controlled resistance element 55 has a control input voltage (gain control signal).
Below a certain value, the gain remains almost fixed. That is, linear amplification is performed. On the other hand, as the control input voltage becomes higher, the gain of the AGC amplifier 53 is lowered. In this way, the amplification level is kept constant.

FIG. 4 shows a configuration example of the band converter 45. The operation of the band converter 45 will be described below. The band converter 45 converts the acoustic signal applied to the input terminal 56 into (π /
2) An acoustic signal having a phase difference of (π / 2) is obtained at its output terminals 58 and 59 by supplying it to the phase difference wave device 57. The two acoustic signals are applied to the multipliers 60 and 61, respectively, and the outputs of these multipliers are added by the adder 62 at a ratio of 1: 1. Reference numeral 63 is an original oscillator, which supplies a high-frequency oscillation signal, which will be described later, to the output terminal 64 and, at the same time, supplies it to the frequency divider 65. The frequency divider 65 frequency-divides the original oscillation signal and outputs the subcarrier −sin ω ₁ t of the angular frequency ω ₁ , and the phase shifter 66 and the multiplier
Supply to the other modulation input terminal of 61. Phase shifter 66 the phase of the subcarrier -sinω ₁ t (π / 2) to obtain only the advancing allowed to subcarrier cos ω ₁ t, and supplies it to good enough for the other input terminal of the multiplier 60.

By exchanging the signals as described above, assuming that the angular frequency of the broadband acoustic signal at the input terminal 56 is p, the acoustic signals at the output terminals 58 and 59 of the (π / 2) phase difference wave detector 57 are cos (pt ) And sin (pt), the output of the multiplier 60 is expressed by the following equation (4).

Cos (ω ₁ t) × cos (pt) = {cos (ω ₁ + p) t + co
s (ω ₁ -p) t} / 2 ・ ・ ・ (4)

Similarly, the output of the multiplier 61 is expressed by the following equation (5). -sin (ω ₁ t) × sin (pt) = {cos (ω ₁ + p) t-cos (ω ₁ -p)
t} / 2 ・ ・ ・ (5)

Since the adder 62 adds the signals represented by the above equations (4) and (5), its output becomes cos (ω ₁ + p) t, and the output of the adder 62 is the angular frequency of the subcarrier. Only the component (ω ₁ + p) of ω ₁ and the angular frequency p of the input acoustic signal appears. That is, the lower sideband, which is the difference component of the angular frequency, is canceled out to zero, and only the upper sideband of the sum component is obtained. The multipliers 60 and 61 are variable transconductance type analog multipliers. The reason for using this will be described later.

67 is a high-pass filter with a cutoff angular frequency ω ₁ , and 46 is an output terminal of the high-pass filter 67. This is also the output terminal of the band converter 45, so
Use the same reference number as 1. High pass filter 67
Is the difference between the angular frequency ω ₁ generated at the output of the adder 62 and the angular frequency p (ω _1- when the addition of Eqs. (4) and (5) does not follow the theory due to the effect of circuit deviation. It is provided to remove the component of p).

FIG. 5 is a diagram for explaining the frequency arrangement of the transmitter of the present invention and the band conversion operation in the band converter 45. FIG. 5 (a) shows a conventional composite signal as in FIG. 12, and the scale of the frequency axis in FIG. 5 is larger than that in FIG. FIG. 5 (b) shows the frequency spectrum of the broadband acoustic signal, which has a wider frequency band than the conventional composite signal, and its maximum frequency is represented as p _max . The acoustic signal shown in FIG. 5 (b) is input to the input terminal of FIG.

FIG. 5C shows the frequency spectrum of the wideband acoustic signal at the output of the adder 62. Figure 5
The part of the frequency lower than the angular frequency ω ₁ in (c) is (4) above.
The angular frequency component of the acoustic signal that satisfies (ω ₁ <p), which represents the component of the difference (ω ₁ -p) between the angular frequency ω _{1 of the} subcarrier in Eq. (5) and the angular frequency p of the broadband acoustic signal. Is
The frequency spectrum turns from DC and is canceled by the addition operation of the adder 62, and therefore is represented by a dotted line in the figure.

The high pass filter 67 shown in FIG.
As mentioned above, this is to remove the residual component of this angular frequency difference (ω ₁ -p), but the component folded from DC is the sum of the angular frequencies (ω ₁ + p) shown by the solid line. When the component of is reached, it becomes impossible to remove it. For this reason,
The value of the angular frequency ω ₁ is higher than 1/2 of the maximum value p _{max. Of} the angular frequency p of the wideband acoustic signal and higher than the maximum angular frequency of the conventional composite signal. It is set to a value that is an integral multiple of the angular frequency (ω / 2) of the signal.

FIG. 5D shows the above-mentioned unnecessary angular frequency difference (ω
_1- p) represents the frequency spectrum of the wideband acoustic signal at output port 46 with the component removed.
wideband acoustic signals expressed in (b) is the angular frequency as spectral shape has a shape of a raised cos (ω ₁ + p) t only omega _1. Here, the ω ₁ component represented by the dotted line is a leak component from the multipliers 60 and 61, but it is sufficiently suppressed, and since it is not superimposed on the conventional composite signal band, it does not cause interference. .

The original oscillator 63 shown in FIG. 4 has an angle which is the least common multiple of the angular frequency (ω / 2) of the frequency pilot, the angular frequency ω of the phase pilot and the angular frequency ω ₁ of the output of the frequency divider 65. Generates a sine or square wave of frequency. Therefore, the output terminal 64 of the original oscillator 63 is applied to the subcarrier generator 10 shown in FIG. 1, and the subcarrier generator 10 operates as a frequency divider and outputs a subcarrier of angular frequency ω.
That is, in FIG. 1, the subcarrier generator 10 is drawn independently from the similarity with the conventional device shown in FIG. 10, but in reality, it operates as a frequency divider and has a subfrequency of ω. Generating a carrier wave.

As a result, as will be described later, if the frequency pilot signal of the angular frequency (ω / 2) is extracted and multiplied on the receiving device side, the subcarriers of the angular frequencies ω and ω ₁ whose frequencies are completely matched are reproduced. can do. In FIG. 1, the operation of the characteristic portion including the omni AGC amplifier 49, the cosine AGC amplifier 50, the sine AGC amplifier 51, and the gain controller 52 will be described below. The omnidirectional acoustic signal from the receiver 1 is amplified with an amplification degree depending on its input level, and the amplified acoustic signal is output via the output point 7 to the mixer.
It is input to 47 and the gain controller 52. The gain controller 52 detects and integrates the amplified acoustic signal to generate a gain control signal. This gain control signal is input to the omni AGC amplifier 49. In this way, an AGC loop is formed and the level of the output acoustic signal from this amplifier 49 is controlled.

The gain controller 52 forms two gain control signals other than the gain control signal to be given to the omni AGC amplifier 49, and forms each gain control signal with a cosine AG.
C amplifier 50 and sine AGC amplifier 51 are supplied to maintain the gain relationship of these three amplifiers 49-51. As will be described later, the gain controller 52 is a cosine AGC amplifier 50.
In addition, the formation of the gain control signal applied to the sine AGC amplifier 51 maintains the linear relationship between the amplified output and the relative direction θ. Figure 6, as already mentioned,
6 is a block diagram showing the configuration of a gain controller 52 that forms three types of gain control signals. FIG.

The gain controller 52, as shown in FIG.
It is mainly composed of a detector 68 and an integrator 69.
The output acoustic signal from the Omni AGC amplifier 49 is a volume
The voltage is divided by V ₁ and input to the detector 68. This volume V ₁ has a function of adjusting the level of the acoustic signal to be supplied to the detector 68. Therefore, the volume V ₁
Is provided in consideration of variations in manufacturing quality of the detector 68 and the like. The detector 68 passes only the plus or minus component of the AC waveform to the integrator 69, and the integrator 69
Integrates the detected plus or minus signal over time over a long period of time. Therefore, by the functions of the detector 68 and the integrator 69, a value obtained by averaging the level of the acoustic signal over time for a long time can be obtained. Note that the integrator
Reference numeral 69 is a device whose output impedance is suppressed low in consideration of the configuration of the output stage described below. The function consisting of the volume V ₁ , the detector 68, and the integrator 69 is
The function is the same as that of the wideband gain controller 44 described above.

The DC voltage signal representing the acoustic signal level obtained as described above is the resistances r ₁ and r ₂
The voltage is divided by and output as a gain control signal for the omni-AG amplifier 49. The DC voltage signal representing the acoustic signal level is divided by the volume V ₂ and output as a gain control signal for the cosine AGC amplifier 50. Further, the DC voltage signal representing the acoustic signal level is divided by the volume V ₃ and output as a gain control signal for the sine AGC amplifier 51.

Therefore, when the omnidirectional acoustic signal level after amplification is large on average from the gain controller 52, a large gain control signal is output to each AGC amplifier 49-51 and amplified. If the subsequent omnidirectional sound signal level is low on average, a small gain control signal will be applied to each AGC amplifier 49.
It is output to ~ 51. The reason why the volumes V ₂ and V ₃ are used as the configuration for dividing the DC voltage signal representing the acoustic signal level to form the gain control signal for the cosine AGC amplifier 50 and the sine AGC amplifier 51 will be described later.

The above-mentioned Omni AGC amplifier 49, cosine AG
Figure 2 shows a C amplifier 50 and a sine AGC amplifier 51.
A configuration similar to that of the broadband AGC amplifier shown in can be adopted.
The two volumes V ₂ and V ₃ provided on the output side of the gain controller 52 described above adjust the variations of the two voltage-controlled resistance elements 55 in the cosine AGC amplifier 50 and the sine AGC amplifier 51, respectively. To do. In the characteristics shown in Fig. 3, the control input voltage whose resistance changes abruptly varies due to individual differences in the elements used.For example, the relative angle θ at which the cosine acoustic signal and the sine acoustic signal have the same value is (π / 4) and for the test input signal in which the AGC function is activated, the two volume V ₂ are set so that the output levels of the cosine AGC amplifier 50 and the sine AGC amplifier 51 become equal to each other. And adjust V ₃ . When making this adjustment,
The divided voltage by r ₁ and r ₂ is
Be equal to the average adjustment point for V ₂ and V ₃ .

FIG. 7 shows a characteristic using a decibel value in order to explain the adjustment method of the _three volumes V _{1 to} V ₃ in the gain controller 52 and the gain and input / output characteristics of each AGC amplifier 49 to 51. It is a curve figure. FIG. 7A shows the input / output characteristics of the omni AGC amplifier 49. An omnidirectional test input signal of a predetermined level x in the constant output level area (AGC area) is input to the omni AGC amplifier 49, and the amplified output signal at that time becomes a predetermined level a, Adjust the volume V ₁ of the gain controller 52. The wideband AGC amplifier described above
The output characteristic of 43 is similar to that of FIG. 7 (a).

In FIG. 7B, the relative direction θ of the sound wave source is (π
/ 4), the level of the 3-channel test input signal that commonly represents the omnidirectional acoustic signal, the cosine directional acoustic signal, and the sine directional acoustic signal from each of the receivers 1, 2, and 3. Is a representation of the output of the cosine AGC amplifier 50 for. The predetermined input level X in FIG. 7 (b) corresponds to the above-mentioned x in the omnidirectional acoustic signal level during three-channel input. The volume V of the gain controller 52 is adjusted so that the output level of the cosine AGC amplifier 50 becomes the predetermined level A for this three-channel input of the predetermined level X, that is, the substantially omnidirectional sound signal level x. Adjust ₂ . Also, in the same way, a sine AGC amplifier is connected to a 3-channel input of a predetermined level X.
The volume V ₃ of the gain controller 52 is adjusted so that the output level of 51 becomes the predetermined level A. The level Y in Fig. 7 (b) represents the input level at which the AGC function is not operating, and the output level at that time is represented by B.

FIG. 7 (c) shows the relationship between the relative direction θ of the sound source and the output from the cosine AGC amplifier 50 as a situation when the incoming sound level is a three-channel directional signal for testing. It is a thing. The straight line a represents the relative direction θ at a predetermined incoming sound wave level in the constant output level region.
And the output level of the cosine AGC amplifier 50.The straight line B shows the relative direction θ in the incoming sound level in the linear amplification region of the input level Y of FIG. 7 (b).
And the output level of the cosine AGC amplifier 50.

When the relative direction θ is (π / 4), FIG. 7 (b)
As is clear from the above, the output level in the AGC region is A, and the output level for the predetermined level Y in the linear amplification region is B. When the relative direction θ is 0, the relative direction θ
The input signal level to the cosine AGC amplifier 50 is increased by 3 dB compared to when is (π / 4). On the other hand, the output signal level from the cosine AGC amplifier 50 is also increased by 3 dB compared to when the relative direction θ is (π / 4).

This is because the gain controller 52 generates the gain control signal from the omnidirectional acoustic signal, and therefore the cosine AGC
The gain control signal to the amplifier 50 is constant regardless of the relative direction θ (however, it is affected by the omnidirectional acoustic signal level), and the cosine AGC amplifier 50 outputs the input acoustic signal increased by 3 dB to the relative direction θ. This is because amplification is performed with the same gain as when is (π / 4).

Such a linear relationship between the input level and the output level established between the two relative directions θ is also established for the other relative directions θ, and the relative direction θ becomes (π / 2). As it goes, the output level goes to zero. Similarly, the sine AGC amplifier 51 also has a linear relationship between the input level and the output level related to the relative direction θ. In other words, although each of the directional AGC amplifiers 50 and 51 has an automatic gain control type input / output characteristic, there is a linear relationship between the relative direction θ and the amplification output level, and the amplification output level is the relative direction θ. Can be said to specify. Therefore, AG
Even if C control is performed, the same orientation determination processing as in the past can be applied.

In FIG. 1, the omnidirectional, cosine directional, and sine directional acoustic signals that have undergone the above-described AGC control are supplied to the mixer 47, as in the conventional transmitter shown in FIG. Therefore, the frequency array of these three directional acoustic signals at the output terminal 48 of the mixer 47 is
It becomes the same as the frequency array of the composite signal shown in Fig. 12, and
(a) is a scaled representation of this scale. on the other hand,
As already mentioned, the frequency spectrum of the output terminal 46 of the band converter 45 is as shown in Fig. 5 (d), so the frequency array of the output terminal 48 in Fig. 1 is as shown in Fig. 5 (e). Thus, a conventional composite signal and a wideband acoustic signal are frequency-divided and arranged (hereinafter referred to as a wideband composite signal).

FIG. 8 is a block diagram showing an embodiment of a receiving apparatus. A part corresponding to the conventional receiving apparatus shown in FIG. 11 is surrounded by a broken line, and the same constituent elements as those of the conventional receiving apparatus are the same. Reference numerals are given. As shown in FIG. 8, in this embodiment, the conventional receiving device is used as it is, and an omnidirectional acoustic signal is output to the output terminal 39.
At 40 and 41, we obtain 8-shaped directional acoustic signals whose maximum sensitivity direction is north-south and east-west on the geomagnetic azimuth axis, respectively. For broadband acoustic signals, FM receiver
This is obtained by pulling the output of 27 from the output terminal 28.

FIG. 9 is an explanatory diagram of frequency conversion of a portion for obtaining a broadband acoustic signal in the receiving device shown in FIG.
Hereinafter, description will be given with reference to FIGS. 8 and 9 as appropriate. In FIG. 8, 70 is a single-peak filter and 71 is a high-pass filter. The inputs to both of these filters are FM receivers 27
Since it is connected to the output terminal 28 of, the frequency array of the wideband composite signal shown in FIG. 9 (a) is the same as the frequency array of the transmitter shown in FIG. 5 (e).

Of these, the single-peak filter 70 has a central angular frequency (ω / 2) and the pass band width is a narrow width corresponding to the frequency deviation of the original oscillator 63 in FIG. As a filter with a steep characteristic curve, a frequency pilot signal of angular frequency (ω / 2) is extracted from a wideband composite signal. The output of this unimodal filter 70 is two frequency multipliers 72 and 73.
Is supplied to. In this frequency multiplier 72, the angular frequency (ω
The angular frequency ω ₂ that is an integer multiple of / 2) is
Angular frequency (ω ₂ -ω ₁ ) that is an integral multiple of angular frequency (ω / 2)
Are reproduced as subcarriers.

On the other hand, the high-pass filter 71 is a high-pass filter with a cut-off frequency ω ₁ , and blocks the passage of a conventional composite signal and band-converts it into a high-frequency band wide-band composite signal cos (ω ₁ + p) t. Pass through. As a result, the frequency spectrum of the output of the high pass filter 71 has the shape shown in FIG. 9 (b).

Reference numeral 74 is a balanced modulator,
Since the output of the filter 71 is input and the output of the frequency multiplier 72 is input to the modulation input terminal, the output of the balanced modulator 74 is expressed by the following equation (6).

Cos (ω ₁ + p) t × cos ω ₂ t = {cos (ω ₂
+ ω ₁ + p) t + cos (ω ₂ -ω ₁ -p) t} / 2 ・ ・ ・ (6)

However, the angular frequency ω ₂ is ω ₂ >
(Ω ₁ + p _max. ) And an integer multiple of the angular frequency (ω / 2). FIG. 9 (c) shows the frequency spectrum of the output of the balanced modulator 74 expressed by the equation (6).

Reference numeral 75 is a low-pass filter having a cutoff angular frequency (ω ₂ -ω ₁ ), which is a balanced modulator represented by the equation (6).
Of the output of 74, only the component of the angular frequency difference (ω ₂ -ω ₁ -p) is passed and input to the balanced modulator 76.

The balanced modulator 76 has the same structure as the balanced modulator 74. A frequency multiplier is used as the modulation input of the balanced modulator 76.
Since the output of 73 is applied, if the output of the frequency multiplier 73 is cos {(ω ₂ -ω ₁ ) t + β} (β indicates an indefinite phase angle), the output of the balanced modulator 76 is , Is expressed by the following equation (7).

Cos {(ω ₂ -ω ₁ ) t + β} × cos (ω _2-
ω ₁ -p) t = [cos (pt + β) + cos {(2 ω ₂ -2 ω ₁ -p) t
+ β}] / 2 ... (7)

FIG. 9D shows the balanced modulator 76 expressed by the equation (7).
Represents the frequency spectrum of the output of. Reference numeral 77 is a low-pass filter with a cutoff angular frequency p _max. Of the outputs of the balanced modulator 76 expressed by Eq. (7), only the angular frequency difference p component is allowed to pass through and is shown by a broken line. Sum of angular frequencies (2
ω ₂ -2 ω ₁ -p) component is blocked.

Therefore, the output terminal of the low-pass filter 77
At 78, a broadband acoustic signal cos (pt + β) having an angular frequency p and a phase angle shifted by β is obtained. This is indicated by the solid line in FIG. 9 (d). Incidentally, FIG. 9 (d) on a subcarrier angular frequency (ω ₂ - ω ₁₎ the sum of the components as well as the angular frequency ₍₂ ω 2 -2ω
_{The 1-} p) components are suppressed to a sufficiently low level.

In the receiving apparatus described above, a conventional receiving apparatus is used to obtain omnidirectional and two-directional 8-shaped directivity, that is, a total of three types of directional acoustic signals, and the FM receiver Since the broadband acoustic signal is obtained only from the output signal, the broadband acoustic signal can be obtained by adding a simple adapter to the conventional receiving device. Further, the signal processing device that processes these three types of acoustic signals to measure the azimuth can be used as it is.

Next, the reason why such a transmission / reception system should be adopted will be described. The sound waves targeted by the receivers 1, 2 and 3 and the broadband receiver 42 shown in FIG. 1 are a mixture of sea noise and sound waves to be captured by wind waves, underwater creatures, navigation vessels and the like. Therefore, the level of the sound wave actually received varies greatly depending on the underwater noise and the situation of the sound wave source. In order to deal with such fluctuations, the conventional FM transceiver uses the margin of the channel spacing of the main carrier to make the FM frequency deviation several times that of the composite signal shown in Fig. 5 (a). By applying it, the required SN ratio was ensured even if the sound wave level decreased.

However, in order to detect the target by using all the acoustic signals that actually exist, a wideband acoustic signal having a frequency 10 times higher than that of the conventional one is received, and
If you try to perform transmission using an FM transceiver, the maximum frequency deviation that can be secured will be small due to the upper limit of the occupied frequency bandwidth of the FM signal, so even if the sound wave level decreases slightly, the SN ratio will decrease. Causes the inconvenience.

In order to improve this, in the present invention, AGC is applied to all acoustic signals to keep the acoustic signal output level at an optimum constant value for a wide range of input sound wave levels, and the SN ratio as a whole. The maximum frequency shift value, which is the most desirable state, is maintained. At the same time, the omnidirectional and cosine used to measure the orientation of the sound source.
AGC operation that maintains the azimuth relationship of incident sound waves is realized for acoustic signals with sign directivity.

Further, in the conventional transmitter shown in FIG. 10, the signals at the output terminals 7, 14, and 15 are mixed once, and this is shown in FIG.
It is also possible to supply AGC amplifier as shown in Fig. 2 to realize AGC operation, but in this case, balanced modulator 1
Since the acoustic signal levels at points 2 and 13 must be kept low, the carrier leak of the subcarrier becomes relatively large. Therefore, as shown in Fig. 1, AGC is applied to each acoustic signal of cosine and sine directivity before the balanced modulation.

The wideband composite signal according to the present invention is
As shown in FIG. 5 (e), since the conventional composite signal is left as it is, the conventional demodulator for reception and the processor for azimuth measurement can be used without modification. Also,
Since the wideband acoustic signal was arranged by converting the band to a higher frequency than the conventional composite signal while maintaining the original frequency spectrum shape, the acoustic frequency band of the existing FM receiver is shown in FIG.
Maximum frequency of the wideband composite signal shown in (e) (ω ₁ + p _max. )
Even if it is not extended to, the lower frequency part, which is the main part of the broadband acoustic signal, can be received.
Further, the 90 ° phase difference wave device 57 shown in FIG. 4 can be easily realized by continuously connecting an all-pass circuit including a resistor and a capacitor.

Further, in the block diagram of the transmitter of the present invention shown in FIG. 1, the outputs of the subcarrier generator 10, the phase shifter 11, the frequency divider 16 and the compass 17 which operate as frequency dividers are rectangular. The structure of the device becomes simpler if it is realized as a wave. In this case, the frequency pilot signal, the phase pilot signal shown in FIG. 12 and FIG. 5 (a), and the directional acoustic signal modulated by the subcarrier shown as 23 in FIG. 12 generate harmonics of odd multiples not shown. To do. For this reason, as it is, it interferes with the band-converted broadband acoustic signal. Therefore, although not shown in FIG. 1, output terminals 7, 14, 15, 1
The conventional composite signals output from 9 and 18 are mixed in a mixer, passed through a linear-phase low-pass filter to remove harmonic components, and then the band-converted broadband sound at output terminal 46. Mix with the signal.

Further, in the band converter shown in FIG. 4, the multiplier
As described above, analog multipliers are used instead of balanced modulators for 60 and 61. The reason is that the multiplication of the switching operation by the balanced modulator generates the harmonics of the subcarriers, and the phase difference of (π / 2) is not maintained between these harmonics. This is because the lower sideband of the frequency (3ω ₁ ) component is superposed on cos (ω ₁ + P) t shown in Fig. 5 (c) (d) and becomes a large disturbance.

In order to realize the band conversion operation by the balanced modulator, first, the angular frequency of the first subcarrier is set to the maximum angular frequency p max. Of the wideband acoustic signal or higher, and (4) ( Five)
The high-pass filter 67 is changed to a low-pass filter by connecting so as to obtain the difference in angular frequency, that is, only the lower sideband component by the addition operation using the formula. Next, prepare a second balanced modulator and a subcarrier, set the angular frequency of the second subcarrier to (angular frequency of the first subcarrier + ω1), and use the low-pass filter to remove the lower sideband component. If taken out, the same as in Fig. 5 (d),
A band-converted broadband acoustic signal is obtained. Whether to use the analog multiplier as shown in FIG. 4 or the two-stage balanced modulator as described above is a trade-off between the manufacturing cost of the device and the detailed performance.

[0080] Further, the frequency conversion of the receiver shown in FIG. 8 and FIG. 9, the angular frequency omega ₂ and (ω ₂ - ω ₁₎ and the was performed using two subcarriers, the angular frequency omega ₁ of Similar results are obtained with only one subcarrier. However,
As can be seen from Fig. 9, the angular frequency ω ₁ is superimposed on the angular frequency band of the original wideband acoustic signal, so the angular frequency ω 1
There is a drawback that the leak component of the sub-carrier of ₁ leaks into the broadband acoustic signal. Therefore, frequency conversion was performed using two subcarriers as in the present embodiment.

Further, in the band conversion explained in FIGS. 4 and 5, when the wide band acoustic signal is converted into the frequency band shown in FIG. 5 (d) by only one conversion operation for obtaining the lower sideband component of the difference. Since the shape of the frequency spectrum is reversed, there is a fatal defect that the low frequency part, which is the main part of the wideband acoustic signal, cannot be received due to the variation in the characteristics of the existing FM receiver.

[0082]

As described above, according to the present invention, a wideband acoustic signal is converted into a high frequency band while maintaining the original frequency spectrum shape and arranged in a higher frequency band than a conventional composite signal for FM transmission. In addition, an amplifier with an automatic gain control function is used as an amplifier that amplifies the captured signal from each receiver, and from the output signal of the amplifier with an automatic gain control function that amplifies the captured signal from the omnidirectional receiver. ,
Since the gain controller for forming the gain control signal for the amplifier with the automatic gain control function for the cosine and sine directional acoustic signals is provided, the conventional receiving device and the azimuth measuring processing device can be used without modification. The wideband acoustic signal can be received by adding a wideband acoustic signal demodulator to the output of the conventional FM receiver, and the SN ratio can be kept good even if the level of the input sound wave is lowered. Furthermore, automatic gain control can be performed while maintaining the azimuth relationship of the incident sound waves.
In addition, the main part of the broadband acoustic signal can be received even if the acoustic frequency band of the FM receiver does not extend to a sufficiently high frequency.

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of a transmission device.

FIG. 2 is an explanatory diagram showing an embodiment of an AGC amplifier.

FIG. 3 is an explanatory diagram showing a characteristic curve of a voltage controlled resistance element.

FIG. 4 is an explanatory diagram showing a configuration example of a band converter.

FIG. 5 is an explanatory diagram of frequency arrangement and band conversion operation.

FIG. 6 is a block diagram showing the configuration of a gain controller.

[Figure 7] Characteristic curve diagram of each AGC amplifier

FIG. 8 is a block diagram showing an embodiment of a receiving device.

FIG. 9 is an explanatory diagram of frequency conversion of the receiving device.

FIG. 10 is a block diagram showing a conventional transmitter.

FIG. 11 is a block diagram of a conventional receiving device.

FIG. 12 is an explanatory diagram showing a frequency array of a composite signal.

[Explanation of symbols]

 1 Omnidirectional receiver 2 Cosine directional receiver 3 Sine directional receiver 4 Amplifier 5 Amplifier 6 Amplifier 42 Wideband receiver 43 Wideband AGC amplifier 44 Wideband gain controller 45 Band converter 49 AGC amplifier 50 AGC Amplifier 51 AGC Amplifier 57 90 ° Phase difference wave device 60 Multiplier 61 Multiplier 62 Adder 66 Phase shifter

Claims (1)

[Claims] 1. An omnidirectional receiver, a cosine directional receiver and a sine directional receiver whose directivities are orthogonal to each other, and an omnidirectional receiver, a cosine directional receiver and a sine. An amplifier that is connected to the directional receiver and that amplifies the signal of each of the receivers and outputs the amplified signal to a balanced modulation configuration unit that includes a frequency pilot signal; A wideband receiver that targets frequencies higher than the signal, a wideband amplifier that amplifies the signal of the wideband receiver, a band converter that frequency-converts the signal amplified by the wideband amplifier, and the balanced modulation configuration described above. In the wave receiving device for a composite acoustic signal, the output signal of the section and the output signal of the band converter are frequency-divided and mixed to obtain a wide band composite signal, wherein each of the amplifiers has an automatic gain control function. Amplification While using, from the output signal of the wide omnidirectional amplifier, a gain control signal for an amplifier that amplifies the signal of the cosine directional receiver and an amplifier that amplifies the signal of the sine directional receiver, respectively. A gain controller is provided, and the band converter frequency-converts the wideband signal into a frequency band higher than the frequency band of the balanced modulation component while maintaining the frequency spectrum shape, and the band conversion Is a 90-degree phase difference wave multiplier, a multiplier that multiplies subcarriers having different 90-degree phases and the output of the 90-degree phase difference wave detector, and an adder to cancel the lower side band or the upper side band. And the frequency of the subcarrier used for the band converter is an integral multiple of the frequency of the frequency pilot signal of the balanced modulation component. , Reception device of the composite acoustic signal.