JPH07221675A - Rearrangement spread type communication system - Google Patents

Abstract

PURPOSE:To provide a communication system where an error characteristic is improved without lowering a frequency use ratio eta. CONSTITUTION:In a communication system where a transmitter transmits either one signal AT j (j=1, 2,...N) of N signals as a transmission signal to a transmission line 3 after the signal is assigned to a frame composed of a certain time width, a receiver receives the sum of the component ARj corresponding to the transmission signal and the noise eN added in the transmission line 3 and a demodulation deciding that the transmission signal is ATj from the received signal is performed, the receiver is provided with the rearrangement spread part corresponding to each ATj and N circuits 7 composed by an analysis decision part.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention
The present invention relates to a rearrangement / spreading communication system for detecting information by obtaining a desired frequency component under a high S / N ratio by diffusing a noise component mixed in a transmission signal during transmission in a communication system.

[0002]

2. Description of the Related Art A block diagram of a conventional digital modulation / demodulation communication system is shown in FIG. 1 is the baseband modulator of the transmitter, 2
Is a final stage modulator using a carrier wave (radio frequency), 3 is a transmission line (wired or wireless), 4 is a first stage demodulation circuit of a receiver, 5 is a baseband demodulator, 6 is a logical value determination circuit, and e _I is a transmission Power binary information (including multilevel information), e _T is a baseband transmission signal, e _TC is a carrier band transmission signal, e _N is noise mixed in the transmission path, e _RC is a carrier band reception signal, e _R Is a baseband received signal, e _D is a demodulated signal, e _o is a reception determination output (binary information), f _c and f _c ′ are carriers used in the carrier band modulation / demodulation process, and (S / N) _R and (S / N ) _D is the reception SN ratio and the demodulation output SN ratio. In the baseband modulator 1 of the transmitter, for example, frequency modulation is performed by e _I to obtain a baseband modulated wave e _T. The output of e _T becomes e _TC by modulating f _c in the final-stage modulator 2, and this is sent to the transmission line 3, and e _TC is usually attenuated and e _N is added during the transmission process, and e _RC ^* (= E _RC + e _N ). The e _RC is demodulated by the local carrier f _c ′ (= f _c ) on the receiving side in the first stage demodulation circuit 4 and becomes a baseband signal e _R ^* (= e _R + e _N ) including noise, and e _R is a baseband demodulation. It is added to the vessel 5. Baseband demodulator 5
, E _R is baseband demodulated to e _D. e _D
Is a signal obtained by adding e _N to the transmission information e _I. This e _D
Is compared with the threshold value eth in the first-stage demodulation circuit 4, a binary logical value is determined, and the result is e _o .
If e _I and e _o match, there is no error, and if they do not match, an error judgment is made. Here, the SN ratio (S /
N) _R is a value determined by the transmission path length, noise environment, etc., and the decision SN ratio (S / N) _D of the demodulation output is a value that depends on the modulation / demodulation method and directly affects the error rate of the decision circuit output e _o. It is an important method evaluation factor. (The final-stage demodulator 2 and the first-stage demodulator circuit 4 may include an amplification / modulation function of an intermediate frequency stage, but the description thereof is omitted here.) As the above-mentioned baseband modulation method, a normal ASK (amplitude) is used. Modulation), FSK (frequency modulation), and PSK (phase modulation) are used. In addition, QPSK (4 phase modulation) and QAM
If (quadrature amplitude modulation) is used, multilevel transmission can be realized with one piece of information. In case of ASK, (S / N) _D and (S /
N) _R is equal, but FSK, PSK, QPSK, QAM
Then, by taking the exclusive band B of the transmission line wider than the bit rate f _b of e _I , (S / N) _{D becomes} (S / N) _R
You can increase more. Therefore, the error rate can be improved.

However, since the improvement of the error characteristic is achieved by using a wide frequency band on the transmission line, there is a drawback that the frequency utilization efficiency η = f _b / B (1) is lowered. Here, FIG. 2A shows an example of the signals e _I , e _T , and e _R described in FIG. In this example, FSK is used. In FSK, two frequencies f _i and f _o are associated with a sequence of binary logical values “1” and “0”. T _D is a frame period and is an occupied time of one piece of transmission information. Here, for simplification, distortion and noise of the transmission line are ignored, and it is shown as e _R = e _T. Also in the following description, noise is separately added, and its influence is examined. Further, the influence of distortion can be eliminated by incorporating an equalization function in the receiver, and therefore its explanation is omitted here.

FIG. 3 is a block diagram of a conventional digital modulation / demodulation communication system using a spread spectrum technique. The baseband modulator 1 portion of FIG. 1 is composed of 1 _A and 1 _B , and
The part of is composed of 5 _A and 5 _B. 1 _A is a spread modulation circuit, 1 _B is the same baseband modulator as 1 in FIG. 1, 5
_B is 5 and the same base band demodulation circuit in FIG. 1, 5 _A despreading demodulation circuit, [M], [M ' ] is equal to each other code sequence with a spreading code. Further, e _IS is a spread signal and e _DS is a demodulated spread signal. FIG. 2B is a time waveform of each part of FIG. In the system shown in FIG. 3, the binary information e _I is transmitted after being spread by the spreading code [M]. A time length for transmitting one piece of information is set as a frame, and one spreading code is sent in one frame.
Specifically, spread code [M] corresponding to "1" and "0"
And its inversion sign

[0005]

[Outer 1] (In the specification excluding the drawings, the inverted code of [M] is expressed as [M] ^# hereinafter). This is called a direct spread spectrum spread system. (In addition, there is a frequency hop type spread spectrum system, where the description thereof is omitted) when the code length of M and N _M, the clock rate f _IS of e _IS is, f _IS = N _M f _b Becomes

Therefore, the occupied band of the transmission signal e _{T obtained} by subjecting e _IS to FSK, for example, is further increased than f _IS .
In this system, the demodulated SN ratio (S /
N) _D is equal to (S / N) _D in FIG. 1, but the value (S / N) _P after despreading by 5 _A is less than (S / N) _D. Code spreading gain G _P = 10log _{It is} improved by ₁₀ N _M (3) and the error rate is reduced accordingly.

However, the transmission path band B has a drawback that it is N _M times as large as that in the case of FIG. As described above, when transmitting information in a poor noise environment due to a long transmission distance or low transmission power, in all of the conventional techniques, it is necessary to increase the transmission path band B in order to prevent a decrease in error rate. It was This has the drawback of lowering the frequency utilization efficiency η.

[0008]

An object of the present invention is to eliminate the drawbacks of the conventional communication system as described above, and to provide a communication system capable of improving the error characteristics without lowering the frequency utilization efficiency η. The purpose is to provide.

[0009]

SUMMARY OF THE INVENTION In order to achieve the above-mentioned object, the present invention provides a transmitter in which any one of the N signals A _Tj (j = 1,
2, ... N) is assigned to a frame having a certain time width, and is transmitted as a transmission signal to the transmission path, and the receiver receives the component A _Rj corresponding to the transmission signal and the noise e _N added on the transmission path. In a communication system for demodulating such that the sum of the received signals is received and the transmitted signal is determined to be A _Tj from the received signal, the receiver is configured by a rearrangement diffusion unit corresponding to each A _Tj and an analysis determination unit. N circuits are provided, the original sample value of the received signal is sampled for each received frame in the rearrangement spreader, and the time waveform of the transmission signal corresponding component A _Rj in the original sample value is stored, and To change the time waveform of the noise e _N ,
A time-position conversion process is performed to create a rearrangement destination time position and a rearrangement sample value based on the original sample value, and an enlarged frame in which the time width of the received frame is enlarged is formed. By analyzing the expanded frame by using a correlation sequence created in advance based on the components A _Tj and A _Rj related to the transmission signal, a large output can be obtained only when the transmission signal is A _Tj. The present invention relates to a rearrangement / diffusion type communication system characterized by the above.

[0010]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention will be described below in detail with reference to the drawings showing the embodiments. 1. FSK-DFT analysis method 1.1 Waveform unit rearrangement method FIG. 4A is a diagram showing a configuration of an embodiment of the present invention.
1, 2, 3, 4, 6 have the same functions as in FIG. Reference numeral 7 denotes a rearrangement / spreading processing circuit (rearrangement demodulation circuit), which has a function different from that of 5 in FIG. 1 and is a portion that realizes a main function characterizing the present invention. FIG. 4B is a detailed configuration diagram of 7,
7 _A is a sampling circuit, 7 _B is a sample value rearrangement / expansion frame generation circuit (conversion circuit), and 7 _C is a feature analysis circuit.

FIG. 5 is an auxiliary explanatory view of FIG. Now, as shown in FIG. 2, the binary information "1" and "0" are set to the frequency f _i.
(I = 1,0) of the sine wave _{e I (f 1), e} I (f o)
Let's say you sent it in correspondence with. Hereinafter, the general representation of the frequency of the sine wave is f _1, and the value of f ₁ or f ₀ is taken. Figure 5
Is a time waveform, and shows an example of e _I (f _i ). Here, 1
The case where four frames of f _i are included in the frame period T _D (f _i = 4f _D , f _D = 1 / T _D ) is shown. here,
The receiver needs to identify in advance the frequency and phase of the carrier wave f _c , as well as the phase (time position) of the frame period T _D. Since this purpose can be dealt with by using well-known carrier wave reproduction technology and frame synchronization technology, the following description will be made on the premise that such synchronization is ensured.
However, the present invention can also be applied to the synchronization technique, which will be described later.

For such FSK transmission, 7 _B is f ₁
Two conversion circuits for detecting and f _o are prepared. Here, the reception waveform corresponding to the transmission waveform e _T for explaining the conversion circuit for f ₁ is e _R. Then, e _R * is a received waveform including noise e _N , and corresponding to "1", e _T = e _I (f ₁ ) = A sin2πf ₁ t 0 <t <T _D (3) e _R * = e _I Let (f ₁ ) + e _N = e _R + e _N (4). Here, since attenuation and distortion in the transmission process can be corrected by the equalization technique, they are ignored for simplification, and instead, noise having power equivalent to that of the signal is considered. Therefore, it is generally expressed by the following equation. e _R = e _T = e ₁ (f ₁ ) (5) For easier understanding now, consider the case of a single frequency sine wave as the noise e _N e _N = Asin2πf _N t (0 ≦ t <T
Let's say _D ). Further, FIG. 5 shows an example in the case of f _N = f _D.

[0013] 7 _A in FIG. 4 (b), to sample over one frame demodulation input e _R ^*. The sampling rate f _S is 7 _c
In order to keep the accuracy of the DFT analysis performed in 1. sufficiently high, it is usual to set f _S >> f _i , f _D (6). Sampled value for one frame e _F (= e _I (f ₁ ) +
e _N ) is sent to 7 _B. The "1" conversion circuit in 7 _B performs the rearrangement process on e _{F as} shown in FIG. That is, four cycles of e _I (f _i ) are set in the area ABCD.
In this case, e _I (f _i ) does not change even if the arrangement order is changed for each waveform region. However, if the noise waveform included in the frame does not match the f _i, which is influenced, it changes the original noise waveform. FIG. 5 is an example in the case of f _N = f _D , and the area order of the first frame: A, B, C, D → e _N0 The area order of the second frame: B, D, C, A → e _N1 Area order of 3 frames: B, A, A, C → e _N2 Area order of 4th frame: C ^* , D, B, A → e _N3 Only noise waveforms are shown as e _N0 to e _N3 , These cascade arrangements are shown as an enlarged frame signal e _EF . Actually four periods of the transmitted waveform e _I (f _i) in which is weight e _EF.

The first frame of the enlarged frames is the incoming original frame e _ND in the case of the figure, but a relocation frame, e.g. e _ND ^' , can be used instead. The area A appears twice in the order of the third frame. When constructing a frame, the region sample value of the original waveform can generally be used 0 to multiple times. Also, in the order of the fourth frame,

[0015]

[Outside 2] (Hereinafter, in the specification excluding the drawings, this symbol is expressed as C ^{* #} ). This shows an operation in which the polarities of the sample values in the region C are reversed and the temporal order is reversed. Even if this operation is performed, the waveform of e _I (f _i ) is kept unchanged. (Note that for such region division, a technique such as increasing the number of divisions and producing regions of different sizes can be adopted. Further, for the sample value of each region, there is also an amplitude axis conversion method described later in Section 4. That is, this processing is a means for changing only the noise waveform (noise diffusion) without changing the waveform of the signal wave to be identified (signal storage).

FIG. 6 is an explanatory diagram of the rearrangement process performed by the "0" conversion circuit in 7 _A of FIG. 4B. Since noise undergoes such changes in the time waveform, its frequency spectrum naturally spreads. FIG. 7 shows a time waveform of e _KF when f _N = 5.1 f _D. Further, FIG. 8 shows a spread spectrum obtained by performing DFT analysis on the waveform of FIG. In the example of FIG. 5, since expanded on the basis of the original signal for one frame is incoming to the four frames, in response to the expansion cycle 4T _D, DF of 7 _C in FIG. 4
The output of the T analysis is in steps of (f _D / 4), and the spectrum is spread up to f _S. Therefore, if the sample point of one frame is N _fo = f _o / f _D and the ratio of the expanded frame to the original frame is generally N _E , the number N _f of spectrum generation points and the frequency interval Δf of the DFT analysis output are It is given by _a following equation. N _f = N _E N _fo = N _E f _S / f _D (7) Δf _a = f _D / N _E (8) Therefore, if ideal uniform spreading is performed, the incoming noise power is Should be distributed over 1 / N _f . However, it is difficult to achieve a completely uniform distribution, and in practice, it is slightly concentrated in a certain frequency band.

From this, the SN ratio of the output e _A of 7 _c in FIG. 4 is given by the following equation. (S / N) _A = (S / N) _R + (S / N) _E (9) (S / N) _E = 10log _1o (αN _f ) (10) α = m / (m + σ) (11) Formula (10) (S / N) _E is the SN improvement amount by the method of the present invention. Here, m is the average value of the power of each spectrum component when dispersed into N _f spectra, and is given by the following equation when the noise power is P _N. m = P _N / N _f (12) σ is the standard deviation of the power of each spread spectrum. Therefore, when σ = 0, the maximum (S / N)
_E is obtained.

When N _E = 1 is set, the conventional method is used, and even if N _f is increased, (S / N) _E = 0 dB. That is, this corresponds to α = 1 / N _f . Also, if only f _s is simply increased, N _f will increase but α will decrease,
(S / N) _E does not increase. The reason for this will be explained below. FIG. 9 shows a signal e obtained by rearranging and spreading the signal and noise components.
_{This is} a modeled characteristic of analysis output power by _EF DFT. Signal components f ₁ and f ₁ ′ (=
corresponding to a point and expansion frame period of f _S -f _i), (f
_D / N _E) spectral component f ^f ₁ to a point each, f ^f _2, f ^f ₃
Occurs. W and W'are the envelopes of power components due to noise. When a strong correlation remains between the frames created by the rearrangement, a part without power remains and α becomes small as shown in FIG. However, if the rearrangement is performed sufficiently randomly, outputs are generated for all the components as shown in FIGS.

[F ₊ ] [F ₋ ] on the frequency axis indicates positive and negative frequency regions, and when DFT analysis is performed, to be precise, a component generated in the negative frequency region is generated on the left side of f _S. . Since the power spectra in the positive and negative frequency regions are equal to each other, f
₁ and f ₁ ′ and W and W ′ are symmetric with respect to the point of f _S / 2. Since there is no DC component in the transmission signal or noise, the DC component of W can be considered to be 0. On the other hand, the value of the upper limit fmax of W depends on the level change between sample points. That is,
It corresponds to the waveform of e _EF shown in FIG. Generally, the higher the fmax, the lower the m and the stronger the diffusion. The main frequency component of the original waveform of f ₁ [FIG. 5] can be eliminated by adding a filter that limits the band to (f ₁ ± f _D ) to remove noise in other frequency bands. Therefore, when N _E = 1, fmax is (f ₁ + f _D ). Now, let τ be the minimum time width of the abrupt waveform change due to the sample point rearrangement performed at 7 _B in FIG. 4B. In the example of FIG. 5, both τ = T _D / 4 and π = 1 / f _s are mixed. The latter corresponds to the large transitions that occur at the boundaries of the region. Generally, when there is a transition of the time waveform every τ seconds, the effective frequency band Fr is given by the following equation. τ F _r ≧ 0.5 (13) In practice, the right side may be assumed to be 0.5 to 1.

In addition to the above-mentioned relation, considering the effect of the waveform change (modulation) due to the transmission information generated in each frame period T _D , fmax ≈ (γ / τ) + f _D (14) γ = 0.5 to 1 (15), and in the case of FIG. 5, if _{τ = T D / 4 (γf} i
+ Given by f _D), also matches the case of f _max = 5f _D next N _E = 1 as gamma = 1. Here, the number in the case of τ = 1 / f _s in FIG. 5 is small, and the effect cannot be sufficiently expressed. In fact, looking at FIG. 8 which shows the spectrum corresponding to FIG. 5, f _max ≠ 20 f _D.

However, if the frequency of occurrence when τ = f _s can be increased, fmax can be increased and diffusion can be strengthened. Now, if rearrangement is performed so that the transition of the time waveform occurs at a high frequency with one sample point as a unit, then fmax ≈ (γf _S + f _D ) (16), and even if γ = 0.5, W and W'slightly overlap as shown in FIG. 9 (b). Further, if γ = 1, then FIG.
As shown in (c), W spreads in the region of approximately 0 to f _S ,
W overlaps with W'in the whole area, and extremely strong diffusion can be realized. Assuming that the rearrangement in sample value units is sufficient for randomization, α → 1 in the equation (10), and strong noise diffusion corresponding to N _f can be performed. Conventional method (N _E =
If the received signal in 1) and the DFT analysis, there frequencies capable of noise becomes a discrete value for each f _D, noise generated assuming limited to the range of the reception band (f ₁ ± f _D) The number of points N _fo is at most three. In the method of the present invention, the number of noise generation points is N _f in equation (7). From this, the SN improvement amount of the method of the present invention compared to the conventional FSK method is

[0022]

[Equation 1] Becomes Although the above equation is an ideal model, it is characterized in that the improvement of the (S / N) ratio can be realized infinitely by increasing N _E and f _S.

1.2 Equal Level Point Rearrangement Method The signal waveform can be kept unchanged even when rearrangement is performed between sample points where the absolute values of the sample values of the transmission signal are equal to each other. FIG. 10 shows an equi-level point of a sine wave and an example of rearrangement thereof. (A) is the same received signal e _R as the transmitted signal e _I (f ₁ ).
(F ₁ = 4f _D ), e _{R1 in} (b) represents one cycle of e _R. ai (i = 1, 2, 3, 4) constitutes a group Ga with sample points belonging to the equal absolute value level v _a of e _R1 .
b _i constitutes G _b with sample points belonging to the same level point v _b . (C) the e _R1 'is an example of a sample value substituted in the same group, v inverts the polarity for _a and -v _a sample point mutual conversion belonging to, as a _i → a _i ^# Relocating (Equivalent level points may be used instead of absolute level points, but the relocation destination is limited accordingly).
Although the signal waveform is kept unchanged, the noise waveform e _N present at each sampling point is greatly changed, and its frequency component f _{N is changed.}
Is distributed to a plurality of other frequency components, so that the peak value of the power spectrum is reduced. This is called a diffusion effect. Note that a _i ^# shown above as a code representing the state in which the polarity of a _i is determined is, to be exact,

[0024]

[Outside 3] However, due to the limitation of usable codes, it is expressed as a _i ^# (only in the specification).

Table 1 shows one method of the random arrangement conversion sequence.

[0026]

[Table 1] This is the exponent part of the N _DFT-th order DFT matrix, consisting of N _L columns × N _L rows (N _L = N _DFT −1), and when N _DFT is a prime number, the y _j arrays in the j-th row are completely mutually Table 1 (a), which is a different combination, shows the case where N _DFT = 5. Using this table, the transformations y _j (a _i → x _ij ) and y _j ′ (b _i → x _ij ′) can be applied to create a new frame. The table shows the polarity conversion

[0027]

[Outside 4] (Hereinafter, in the specification excluding the drawings, it is expressed as X _ij ^# )
Displayed in. In FIG. 10C, a _i and b _i indicate an example of conversion using the row of j = in Table 1 respectively.
The number N _L of equal absolute level points in e _R and the size N _DFT of the _DFT correspond to the number N _c of cycles of the signal in T _D : N _L = 4N _c (18) N _DFT = 4N _c +1 ( 19). From this, 4N _c different y _j arrays are created. Table 1 (b) shows an example of the arrangement for N _c = 4. On the other hand, the number N _G of groups having equal absolute value level points such as v _a and v _b is given by the following equation. N _G = f _S / (4f ₁ ) (20) When one rearrangement frame is created, the rearrangement is performed by giving different row arrangements y _j to each of the N _G groups as much as possible. It can realize various conversions.

Next, a conversion circuit for the sine wave signal e _R of FIG. 10 (a) is made by using the rearrangement method of Table 1 (b), and the diffusion effect on the noise e _N weighted by e _R is investigated. See.
First, the method parameters are determined as follows. (A) f ₁ = 4f _D , f _S = 112 f _D , N _E = 4, total number of sample points 112 = 448 From this, the following parameters are derived. (B) N _c = 4, N _L = 16, N _G = 8 Four rearranged frames F _k (k = 1,
Table 2 shows an example of the configuration of 2, 3, 4).

[0029]

[Table 2] Here, the values in Table 1 (b) are used for y _j [shown by ∘ ...] in Table 2. In this case, f ₁ = 4f
Make a rearrangement algorithm in _D and use e _R as input
_{(F 1 = 4f D),} e N (f N = 3.5 f D, 3.75f D)
Figure 1 shows the result of computer simulation for
1 to 13. In this case, the minimum interval of waveform change due to rearrangement is τ = T _S = 1 / f _S. Since the frequency is high, the power of e _N spreads every (f _D / 4) from 0 to f _S as shown in the figure. The increment of the SN ratio in FIG.
In (0), (S / N) _E > 13 dB, and remarkably S
Improves N characteristics. In FIG. 12, since f _N is close to f ₁ , the f ₁ component remains, but it can be reduced by increasing N _E. It is also possible to improve by the method of item 5 described later.

1.3 Function Conversion Method In the method of section 1.2, there is a constraint that the sample points whose arrangement can be changed are equal absolute value levels. This restriction is an element that prevents improvement in characteristics. Here, a method of enabling the arrangement change between arbitrary sample points will be described. FIG. 14 shows the received signal waveform e _R (t). Now, the time t _1, t the phase angle of the _two signals to be converted to _{θ 1 = 2πft 1, θ 2} = 2πft 2
And sample values at these points are a ₁ = A cos θ
₁ , a ₂ = Acos θ ₂ , θ ₂ = θ ₁ + Δθ. Then, in order to save and convert the waveform of e _R , use the complex sine wave conversion here.

[Equation 2] Considering, the following equation holds.

[0031]

[Equation 3] From these relational expressions, when the converted value corresponding to the value of the expression (23) is shown with a dash, a ₂ ′ = a ₁ c−b ₁ d (25) b ₁ ′ = a ₁ d + b ₁ c (26 ) B ₁ = Asin θ ₁ = Acos (θ ₁ −π / 2) (27) is obtained. That is, from the sample values (a ₁ , b ₁ ) of the two original frames shown in FIG. 14, using the complex numbers (c, d) corresponding to the phase difference, the transformed sample values (a ₂ ′, b) of the rearrangement destination. ₂
′) Is obtained. b ₂ is a sample value at the position of (θ ₂ + π / 2). For simplicity, (a ₁ , b ₁ ) and (c,
It may be considered that only a ₂ ′ is obtained from d), (b ₂ ′ is not particularly used), and this is converted (a ₁ → a ₂ ′). By this method, the sample value at t ₁ can be converted into the sample value at an arbitrary time t ₂ and the time position can be moved.

When the above method is used, the degree of freedom in designing becomes extremely large when manufacturing different rearrangement patterns, and the diffusion effect can be enhanced. This method corresponds to the case of N _G = 1 in the method of the above-mentioned item (b), and the number N _{L of} sample points that can be freely replaced is given by the following expression. _{N L = f S / f D} (28) Therefore, the number of all different arrangement patterns of the components is (N _L +1) If prime, the N _L pieces together. When defining the number of frames as N _E * which can now while maintaining a high diffusion efficiency (while maintaining high values of alpha) expansion, since N _E * become ≒ N _L (29), on the N _E of the formula (7) Letting N _f * be the total number of sample points / enlarged frame obtained by substituting N _E * in the equation, N _f * = N _E * f _S / f _D = (f _S / f _D ) ² (30) N _f * is the total number of sample points when an enlarged frame is created under the constraint of keeping the value of α high, and from this and equation (10), the SN improvement amount of the following equation is obtained.

(S / N) * _E = 10log ₁₀ (αN _f *) = 10log ₁₀ [α (f _S / f _D ) ² ] (31) Even if N _E is increased and N _f is increased, the rearrangement pattern The higher the mutual similarity, the lower the value of α. That is, the result is that power is concentrated at a specific frequency. However, in this method, α can be kept high up to the limits of equations (29) and (30).
From the equation (31), if it is necessary to further improve the SN, it suffices to increase f _S , and a high SN ratio can be realized relatively easily. f
If there is no particular restriction on the increase of _S , this method can increase the (S / N) ratio infinitely.

Although the complex sine wave is used in the above description, a
To obtain ₂ ′, two sample values (a
_The same conversion can be performed by using ( ₁ , b ₁ ) and selecting the phase difference θab between a ₁ and b ₁ to a value generally shown by the following equation. θab = π / 2 ± Nπ (32) Here, N is an integer. More generalized, θab is
It is possible to derive a ₂ ′ by using a general value of θab ≠ 0, but in this case, the conversion formula of the following formula is used. That is, when a ₂ ′ is obtained from a ₁ = Acos θ ₁ (33) b ₁ = Acos (θ ₁ −θab) (34), a ₂ ′ = a ₁ cos Δθ− [b ₁ −a ₁ cos θab ] Sin Δθ / sin θab (35) b ₂ ′ = [b ₁ −a ₁ cos θab] cos Δθ / sin θab + a ₁ sin Δθ (36) is obtained. Therefore, by giving θab and Δθ, it is possible to obtain (a ₂ ′, b ₂ ′) from the original sample values (a ₁ , b ₁ ). Therefore, this method can also be used for diffusion, and the value of θab can be changed when obtaining the sample value of each relocation destination. That is, for example, if θab is determined by the method of Table 1 or a random number, then a ₁
While widely changing the time position of b ₁ used corresponding to
It is possible to sequentially find a ₂ ′ (or a ₂ ′ and b ₂ ′), which makes it possible to realize more ideal diffusion.

In the above description, two original sample values (a
_{Although the} rearrangement destination sample values (a ₂ ′, b ₂ ′) are obtained from ₁ , ₁ b ₁ ), it is also possible to obtain _one original sample value a ₁ to a ₂ ′. The method will be described below. a ₁ = A cos θ ₁ (37) a ₂ = A cos (θ ₁ + Δθ) (38) From the converted sample value a ₂ ′,

[0036]

[Equation 4] Is required as. In this method, considering that noise is also included in a ₁ when θ ₁ = π / 2, as it can be a ₁ ≠ 0, this case becomes a ₂ '→ ∞. Generally ε <<
For example, it is necessary to avoid using original sample values in the range of | cos θ ₁ | <θ ε (40).

As another method, the root mean square value of the sample values of the original frame in the past

[0038]

[Outside 5] (Hereinafter referred to as A ^{# in the} specification excluding the drawings), and ignoring the influence of noise,

[0039]

[Outside 6] (Hereinafter, in the specification excluding the drawings, it is expressed as A≈√ (2) A ^# ), so that A ^# is obtained. In this case a
_{Never 2} 'is excessively large, the conversion sample values is determined. In this way, rearrangement diffusion can be realized by converting a ₁ → a ₂ ′.

FIG. 15 shows one of the results of computer simulation according to this method. 1.4 Multilevel Transmission Method In the above description, the transmission method in which the binary information on the transmission side is associated with different frequency groups f ₁ and f ₀ is used. This is called BFSK (Binary Frequency Shift Keying). Using the principle of the present invention, multivalued information can be transmitted in association with different frequencies fi (i = 0, 1, 2, ... N-1). Correspondingly, N frequency detection circuits are required, but the amount of information that can be sent in one frame increases from 1 bit to I = log ₂ N (bit) (42). Limit value of N is given by the interval △ fa of the output frequency f _a post DFT analysis of equation (8). That is, it is necessary to select f _{i + 1} ≧ f _i + Δf _a (43). If the frequency interval is reduced below this limit, the detection output leaks to adjacent frequencies, resulting in deterioration of the SN ratio.

Even if it is selected under the constraint of equation (30),
Let us consider a case where a frequency whose cycle is not completed in one frame is selected as the transmission frequency such that f _i ≠ kf _D (44) holds when k is a positive integer. When the frames in this case are made continuous, as shown in FIG. 6, at the frame boundaries,
A waveform discontinuity indicated by e _j occurs. Therefore, even in the absence of noise, the output frequency f _a by DFT analysis
Includes the frequency components before and after that, and is as follows. Here, h is a positive integer. This phenomenon is caused by _a decrease in the output of the f _i component in f _a and the occurrence of components other than f _i .
(S / N) Decrease _E. This phenomenon can be avoided by extending or shortening the frame period.

In FIG. 16, the frequency corresponding to "1" is f
Received waveform e _R (f ₁ ) when selected to be 3.75 times _D
And e _R (f ₁ )
Indicates _RE (f ₁ ). e _RE (f ₁ ) cycle is from T _D to T _DE
Since it is extended to 2, it can accommodate exactly 2π × 4 cycles of f ₁ . Thus, a DFT analysis over a period of T _DE will detect a single sine wave when the noise is not weighted. This is called a period extension method. In this case, the last sample value S _{2 of} e _RE (f ₁ ) is e _R (f ₁ )
, _One of the other parts of the same sinusoidal waveform of e _R (f ₁ ) that is not present in S ₂ , such as S ₁ in the figure, and
Also used as. That is, S ₁ is used twice.

In the figure, it is also possible to delete a part S ₃ of e _R (f ₁ ) and use just 2π × 3 cycles of f ₁ . This is a cycle shortening method. Since the portion where the signal exists is discarded, it is slightly disadvantageous in terms of SN ratio compared to the above-described expansion method. Such processing can be easily realized by temporarily storing the sample values of the original frame in the memory and then rearranging them. In any of the methods described above, the signal e _RE (f ₁ ) of an integral multiple can be easily obtained by using the signal of a non-integral multiple of f _D. 4.1 (a) for this e _RE (f ₁ )
If the rearrangement diffusion processing is performed according to the methods of (c),
A similar improvement in SN ratio is obtained. By applying this method, the frequency interval that can be used on the transmission side can be narrowed to Δf _{a in} equation (8). FIG. 2 already described is a time waveform when binary information is represented by ON / OFF pulses and a frequency corresponding thereto is set.

FIG. 17 shows each FS when the modulation shown in FIG. 2 is performed.
The spectrum of K system is shown. In (a), since f ₀ and f ₁ can be separately detected in the spectrum of the conventional BFSK system, f _SP = f _D is widely set when the interval between them is f _SP . Next, (b) is a spectrum when the present invention is applied, and f _SP can be set to Δf _a in Expression (8). (C) is a spectrum in the case of 8-level FSK. Conventional FS
The transmission path occupied band in the case of K and the method of the present invention is given by the following equation, where μ is the multilevel value. (Conventional FSK) B ₁ = (μ-1) f _D + 2f _D (46) (FSK of the present invention) B ₂ = (μ-1) Δf _a + 2f _D (47) Based on the frequency efficiency of the conventional FSK Then, the relative efficiency η _r of the method of the present invention becomes η _r = 1.0 f _D / Δf _a = 1.0 N _E (μ >> 2) (48) for a large μ. Therefore, there is an advantage that the frequency utilization efficiency can be improved by increasing N _E.

2 PSK-DFT method Since the DFT analysis output can detect not only the frequency component of the signal but also the phase component, phase modulation is used on the transmitting side and demodulated on the receiving side by the PSK method in accordance with item 1. The technology already described can be applied. Since the above-mentioned rearrangement method has a signal storage-noise spreading function, when the frequency of the incoming signal is the same and the phase is different, the waveform of the transmission changes in any of the above two methods except for the function conversion. To do. Therefore, it is generally applicable not only to binary phase modulation but also to multiphase modulation. Consider a case where the polyphase modulation is performed by the method of 1.2. Now, two signals e _a and e _{b having a} phase difference φ with each other.
think of. _{e a (t) = Acos 2πf} t (49) e b (t) = Acos (2πf t -φ) (50) Fig. 18 is a case of adding the e _b (t) to the detection circuit of e _a (t) FIG. E _a corresponding to the equal level v now
Suppose that the rearrangement is performed on four sample points at time t _i (i = 1,2,3,4) in (t). Even if a ₁ is moved to any of t ₂ to t ₄ , e _a (t) does not change if only polarity conversion is performed. This method is applied to e _b (t), and b _{1 is set} to t ₂ ~
If placed at t ₄ , the absolute values of their sampled values b ₂ ′ to b ₄ ′ are equal to b ₁ . The actual values b ₃ and b ₃ ′ match but otherwise do not. Let this value be Δb.
For example, at the t ₂ point, Δb = b ₂ ′ −b ₂ = A {cos (2πft ₂ −φ) −cos (2πft ₁ −φ)} = 2A sin 2πft ₂ sin φ≈ (Asin 2πft ₂ ) × 2φ (51 ) From this result, when the rearrangement is made to the half of the rearrangement destinations in the equi-level conversion, the wave that does not match the reference phase undergoes a waveform change, so that the frequency component is diffused. Therefore, when only e _b (t) is input, the DFT output has frequency f
The phase amount with respect to the output of e _a (t) differs not only by φ, but also the absolute value of f decreases and spreads to other frequency components, which facilitates identification. This means that the phase difference can be detected even if the value of φ is designed to be small, so QP
There is an advantage that SK and a multi-phase transmission method with more phases can be realized.

From the above description, the present invention can be applied to the determination of the phase difference when using the ASK or QAM system as the modulation system on the transmission side. In the case of QAM, it is necessary to identify the reference amplitude of the received signal. That is, it is necessary to have a function of determining the magnitude of the amplitude of the DFT output based on the reference level. This purpose can be realized by a method of occasionally transmitting a pilot signal. Although the present invention can reduce the phase difference and increase the number of multi-levels as described above, a plurality of code words such as an orthogonal code and a pseudo-orthogonal code are made to correspond to multi-levels and the code "0", " 1 "," 2 "
, Can be made to correspond to different phases, multilevel transmission can be performed.

FIG. 19 (a) shows a method of transmitting a fourth-order Hadamard matrix in correspondence with four values. Each frame is composed of 4 chips, each chip consists of 1 cycle, and "1", "" of the Hadamard matrix are used. In this example, 0 "corresponds to phases 0 and π. Since these four rows H _i i = 0,1,2,3) are orthogonal to each other, a normal phase demodulator corresponding to the transmission waveform e _Ti (the reception waveform is e _Ri = e _Ti ) is used. The transmission information can be determined by preparing four pieces. However, the SN ratio cannot be improved. With the principle of relocation diffusion of the present invention, 7 of FIG. 4 _A
, The i-th circuit converts the signal waveform by inverting the polarity of the sample value of 4 chips,
e _Ti → e _To can be realized. That is, if the predetermined signals are added to the outputs of the four circuits, they all have the same waveform as e _TO . Thereafter, rearranges diffusion described in item 1 by 7 _B of FIG. By this process, 7 _C in FIG. 4 outputs the frequency and phase components of e _{TO with} a high SN ratio. Here, since the output obtained by inverting the sign of H _i can also be used, transmission of 8-value (3 bits) / 1 frame can be realized by this method. FIG. 19B shows the Hadamard matrix

[0048]

[Outside 7] (Hereinafter, in the specification excluding the drawings, it is expressed as (-) ^# ),

[0049]

[Outside 8] Four transmission waveforms in the case where phases (−π / 2) and (π / 2) are made to correspond to (hereinafter, expressed as (+) ^{# in the} specification excluding the drawings) are shown. If both (a) and (b) are used and the case of the inverted code is included, 16-value transmission is performed. In this case, when the normal phase demodulation is performed, the phase difference is smaller than that in the case of FIG.
(S / N) _D decreases. However, in the method of the present invention,
First figure (b) (-) by reversing the polarity of the original sample values ^# chip waveform, (+) ^# (hereinafter, in the specification, except the drawing, (+) ^# and expressed) chip Smoothly connect waveforms. As a result, all outputs match e _TO ′, and a high signal-to-noise ratio can be obtained by the rearrangement diffusion method described above. In general, the phases for (−) ^# and (+) ^# are made to correspond to −θ _j and (−θ _j + π). Here ^'the difference thought _{△ θ j = θ j - θ} j' different values theta _j to theta _j by the above-mentioned phase identification performance and using the principle of even select smaller present invention, to obtain a high SN ratio be able to. Now
Choose Δθ _j = π / q and set the order of the Hadamard matrix to N _H
(This corresponds to the number of cycles / frame. It is equal to N _C mentioned above.)
Then, the amount of information transmission / frame is I = log ₂ (2qN _H ) (bit) (52).

As described above, in the reception demodulation circuit of the communication system for transmitting the discrete information by the predetermined modulation, the sampling value is extracted from the reception signal including the target waveform having the repetition period T _i = 1 / f _i , and it is necessary. Then, after performing the rearrangement conversion processing or the processing by the filter, the waveform is transformed into a special waveform by performing the nonlinear processing on the amplitude axis or the time axis on the sample value, and as a result, the integer of f ₁ is obtained. Double frequency component k
f ₁ (k = 1, 2 ...) Is generated, the noise component in the output is spread widely in the frequency region that does not match kf ₁ by this nonlinear processing, and these outputs are rearranged and transformed if necessary. After the processing is performed, it is possible to identify the target waveform in addition to the analysis unit.

3 Chirp Modulation (CM) -DFT Method (Time Axis Conversion Prefix Method) The present invention provides an effective reception method even when the chirp modulation (CM) is used as the transmission side modulation method.
FIG. 20 is an explanatory diagram of the principle of the CM-DFT method. Figure C
₁ , ₁ and C ₀ are chirp modulation characteristics corresponding to "1" and "0", which are frequency-time characteristics. t _S indicates the start point of the chirp. e _T1 (t) and e _TO (t) are transmit chirp waveforms,

[0052]

[Outside 9] (Hereinafter, in the specification excluding the drawings, expressed as C ₁ ^# , C ₀ ^# ) is a time axis companding characteristic showing a change of λ = T _S ′ / T _S with time t, where T _S is the reception Unit time of signal, T _S ′
Is the unit time after companding, t ₁ and t ₀ are C ₁ ^# and C ₀ ^# where λ = 1
Is the point. e _F1 (t) and e _F0 (t) are time waveforms of the frame signal after the time axis conversion, and both waveforms match. .

[0053] When the CM-DFT scheme, 7 _A of FIG. 4 converts the input chirp signal into a single sinusoidal signal has a function of performing the sampling. When the chirp modulation shown by C ₁ and C ₀ in FIG. 14 is performed, if the time axis of the sampled value is compressed or expanded on the receiving side as shown by C ₁ ^# and C ₀ ^# in the figure, e
_F1 (t) and e _F0 (t) are the chirp signal frequencies f ₁ and t _{1 at} times t ₁ and t ₀ at which C ₁ ^# and C ₀ ^# are 1 respectively.
The waveform of f ₀ is continuous. (here,

[0054]

[Outside 10] By appropriately selecting (hereinafter referred to as C ₀ ^# in the specification excluding the drawings), e _F0 (t) can be set to another frequency, for example, f ₁ . ) By applying the method described in section 1 to e _F1 (t) and e _F0 (t), the noise component that is superimposed on this signal component and that comes in can be diffused.
It is possible to sufficiently increase the determination SN ratio after the FT process.

In the systems described in the section 1, if a noise component having the same frequency as the transmission signal, for example, f _N = f ₁ and the same phase is added, such noise is spread by the rearrangement spreading method described above. It will not be possible. As a result, even when the signal of f ₁ is not transmitted from the transmitting side, the receiving side erroneously determines that f ₁ is transmitted. CM-DFT
Then, since the noise of the single frequency f ₁ is diffused by the time axis companding process, the output after the time axis companding is f
_The probability of matching ₀ and f ₁ is extremely small. It is also possible to use a gradient different from the above-mentioned C ₁ and C ₀ for the chirp modulation method. The starting point t _S can be set freely.
Further, if a high-order characteristic such as a quadratic equation is used as the chirp characteristic, a large number of chirp waveforms can be used, so that multilevel transmission can be easily realized.

4 Non-sinusoidal Waveform Modulation-DFT System [a] Amplitude Axis Conversion System In Sections 1 and 2, an example of application to a modulation system based on sinusoidal wave modulation has been described, but an arbitrary waveform other than a sinusoidal wave is used. The present invention can also be applied to the case of transmission by transmission. Although it is not a sine wave by nature, such as a memory read output or a medical machine diagnostic output other than a communication system, but the type of its waveform is limited, and its position and waveform can be expected on the receiving side, The present invention provides an effective reception detection means. Figure 21
[Fig. 6] is an explanatory diagram of reception determination of a signal transmitted by non-sinusoidal waveform modulation. e _T1 and e _TO are transmission waveforms corresponding to "1" and "0", and e _F1 and e _F0 are amplitude axis conversion output waveforms, which are sine waveforms.

In the circuit configuration of FIG. 4 for detecting e _T1 , 7 _A takes a sample value of the incoming e _T1 and converts it to the sample value using the amplitude axis conversion function ξ (t) below. Get the sample value. To determine ξ a (t), Expressed sine wave e _F1 corresponding thereto and portions e _T11 corresponding to the time 0 to t ₁ in the waveform of e _T1 in continuous function, e _T11 (t) = At _{/ t 1 (0 s t <} t 1) (53) e F1 = Asin 2πf since ₁ becomes t (54), the amplitude axis transformation function of this _{section, ξ (t) = t 1} sin2πf 1 t / t ( 55) given in. Similarly, the functions of other sections can be obtained. If the noise waveform e _N (t) = A _N sin (2πf ₁ t + φ _N ) (56) is superposed on e _T11 , the converted output is

[0058]

[Equation 7] Therefore, the noise component mainly diffuses to a value other than f ₁ regardless of the value of the frequency component of the incoming noise, including the case of f _N = f ₁ and φ _N.

Therefore, when e _R * shown in equation (5) comes in, the sampled value of e _T1 (t) corresponding to e _I (f ₁ ) is converted into the sampled value of the sine wave of equation (54). Since e _N is mainly frequency components other than f ₁ , this converted output

[0060]

[Outside 11] (Hereinafter, in the specification excluding the drawings, it is expressed as e _R ^{* #} )
By sequentially applied to 7 _B and 7 _C in FIG. 4 the sample values, if Hodokose relocation diffusion process described in item 1 can detect f ₁ component at a high SN ratio by the principle described above, the transmission signal e _T1
Can be determined.

For the circuit for detecting e _{TO in} FIG. 21, if a function of performing amplitude conversion of e _TO into the illustrated output e _F0 (t) = A sin 2πf ₀ t is prepared, e _{TO is operated on the} same principle. Can be determined with a high SN ratio. Generally, as a modulation waveform on the transmission side, if a large number of waveforms are selected from the waveforms having a small cross-correlation value and used, the equation (3
It is possible to realize a multilevel transmission method in which the information amount shown in 7) is transmitted for each frame.

[B] Original waveform storage-correlation analysis method When the transmission waveform e _T1 of FIG. 21 is received, this can be used as the original waveform and the original sample value can be rearranged while the original waveform is stored. That is, the principles of the terms 1.2 and 1.3 can be applied to the case of such a non-sinusoidal wave. FIG. 22
(A) is an explanatory view of a relocation method of equal absolute value level points.
time _{t i (i = 1,2, ···} 8) which e _T1 takes a constant amplitude V _a when inverting the polarity when the sample value ai in the required t _i
You can move to any other time in. Rearrangement diffusion processing is performed by this method. FIG. 22B is an explanatory diagram by the function conversion method, where a ₁ ,

[0063]

[Outside 12] It is shown that the sample value of a6 can be calculated by using the sample value of (hereinafter, referred to as a ₁ ^# in the specification excluding the drawings) and the waveform property of e _T1 . That is, generally, the sample value at the time ti in the i-th time domain shown in the figure is given by a _i = α _i A [γ _i + β _i (t _i / t ₀ −i + 1)] (58) from the waveform of e _T1. To be The values of the symbols α _i , β _i , γ _i are shown in the figure. When the calculated value of a ₆ is obtained from the two points of a ₁ ,

[0064]

[Equation 8] Given in. Here, t _p is a unit of the time axis. Generally at time t _j ,

[0065]

[Outside 13] (Hereinafter, in the specification excluding the drawings, it is expressed as t _j ^# )
The sampled value of a _j ,

[0066]

[Outside 14] (Hereinafter, in the specification excluding the drawings, expressed as a _j ^# ), the converted sample value a ′ _k of the relocation destination t _k of a _j is _obtained , and the values are C ₁ ^# and C _0. Given by ^# . Where F is e
It is a function similar to equation (58) determined by the waveform of _T1 . Thus, the two original sample values to calculate the converted sample values relocation destination based on, by arranging the converted sample values in the relocation destination, can make a larger frame in 7 _B in FIG. This method has no restrictions on the relocation destination.

The frame formed by the conversion sample values obtained by the conversion method of FIG. 22 stores the original waveform consisting of triangular waves, but the waveform of other noise e _N is different from the original waveform and its spectrum is spread. . The magnified frame thus created is applied to 7 _C via 7 _B in FIG. When the DFT analysis is performed at 7 _C , a plurality of frequency components corresponding to the triangular wave are detected as large values. This set of frequency components is different for e _T1 and e _TO , so that the transmitted waveform can be determined. Generally, when a large number of types of waveforms e _Ti (i = 0, 1, 2, ... N-1) are used as the transmission waveform, DF
Instead of the T analysis, the analysis waveform e _Ai derived from e _Ti is shown in FIG.
Was prepared in 7 _C, performs correlation calculation between FIG 23 (a) larger frame and e _Ai. That is, in the correlation calculation with N kinds of waveforms, when the correlation value with the j-th waveform is strong, an output as shown in the figure is obtained, and from this, it is determined that the transmission waveform is e _Tj . Here, the analysis waveform A _i is the time-axis sampled value sequence of the i-th transmission waveform e _Ti [S _0i , S _1i , ... S
Assuming that _N−1 , _i ], the transmission matrix [S] is as shown in FIG. From this, the analysis matrix corresponding to the analysis waveform e _Ai is the complex co-transpose matrix of [S]

[0068]

[Outside 15] (Hereinafter referred to as [ts] ^{# in the} specification excluding the drawings). The aforementioned DFT analysis is a kind of such correlation calculation. Using this analysis technique, the selection range of the transmitted signal is significantly expanded. However, the smaller the cross-correlation between the waveforms of different e _Ti, the easier the determination. Further, in this method, the time width of the expanded frame does not have to be an integral multiple of the transmission frame, and if e _Ai is selected as a waveform corresponding to the time width of the expanded frame, analysis determination can be performed. Furthermore, it is not necessary to always keep the time width of the transmission frame constant.

FIG. 24 shows a minimum type of PSK.
The transmission waveform by shift keying (MSK) is shown.
In this figure, the phases 0 and π are associated with "1" and "0". This modulation method has an advantage that the occupied band of the transmission line can be narrowed because the sine waveform within the frame period is weighted. The two types of methods described above can be applied to such a modulation method. F _{A in} FIG. 24 is an amplitude axis conversion function,
It shows the time change of the ratio p = x '/ x of the amplitude scale of the input / output signals x and x', and is given by F _A (t) = (sin πf _D t) ^-1 (61). Although the value pm is exceeded in the vicinity of the frame boundary, an interpolated value of adjacent sample values is used for this portion. This is because if there is a large amount of noise in this part, the converted output becomes abnormally large. In the figure, e _D (t) is an output waveform when the input is only a signal component, and is a PSK waveform with a constant amplitude. If the rearrangement diffusion processing described above is performed on this waveform, it is possible to perform identification with a high SN ratio.

5 Code Sequence Modulation Spreading System This system is one in which the present invention is applied to the spread spectrum (SS) transmission system described in FIG. 3, and the code of Formula (3), which is a feature of the conventional SS system. With the diffusion processing gain G _p ,
It is possible to obtain the SN ratio improving effect by the SN improvement amount (S / N) _E of the equation (10). FIG. 25 shows an embodiment of the present invention, in which a code spreading function is added to the structure shown in FIG. (A) is a modulator on the transmission side, 1 _A is a modulator for performing FSK with information e _I , 1 _M is _M with output e _S of 1 _A
A spread modulator 2 for modulating a sequence is a carrier band modulator 2 for performing FM on a carrier f _C , for example, with an output e _T of 1 _M. The output e _TR of 2 is transmitted to the transmission path.
(B) is a receiving side demodulation unit, 4 is a carrier band demodulator for demodulating a received signal e _RC * including noise by a carrier wave f _C ′, 7 _M
Is a demodulator that despreads the output e _R * of 4 with the M sequence, 7 _A
Is a circuit for sampling a spread output e _S * (including noise) of 7 _M to obtain a sampled value sequence of the original frame, and 7 _B and 7 _C are circuits having the same function as in FIG. FIG. 26 is an example of the time waveform e _T (t) of e _T in FIG. 25, which is a part of the input sine that is a part of e _S (e _S ^* excluding noise and equal to the output of the transmitting side I _M ). One cycle of the wave changes as shown when the 7-chip M series is excluded. This corresponds to the waveform e _R (t) obtained by removing noise from the received signal e _R *.

Since the operation of this system is clear from the description of the SS system of FIG. 3 and the FSK-DFT system of FIG. 4, detailed description will be omitted. In the transmission line, noise biased to a part of the band or noise of a single frequency component is easily mixed, but in the present invention, such noise component is 7 _{M in} FIG.
It is spread by the despreading function. This spread output is
The probability of matching the frequency component f _i (i = 0, 1, ...) Used as information is extremely small and can be practically ignored. Accordingly, an advantage that by relocation diffusion performed by the 7 _B in FIG. 25, the noise diffusing function is almost ideally performed. This system has a powerful noise diffusion function like the systems of the 3rd and 4th terms. Here, FIG. 27 shows the result of the computer simulation. This is obtained by applying the modulation method of FIG. 25 to the method shown in FIG. 12 with M sequence length [M] = 7, and a great improvement can be obtained compared to FIG.

6 Rearrangement Spreading Iteration Method FIG. 28 is a configuration diagram in which the receiving circuit of FIG. 4B is modified. Reference numeral 8 in the drawing is a filter, and other symbols are the same as those in FIG. The rearrangement spreading iterative method _applies the expanded frame output e _EF1 after the rearrangement spreading processing performed in 7 _B1 to the filter 8 ₁ . 8 is a constituent component of a sine waveform or a general waveform to be detected here [e _R component in e _R * of the equation (5)]
It is a bandpass filter that mainly passes only. The filter 8 is composed of a digital filter or an analog filter. In the latter case, the output is resampled. The output e _F2 of 8 ₁ generally includes not only the e _R component but also the noise e _N unless the length of the expanded frame is ∞. e _F2 is added to 7 _B2 again, rearranged and diffused,
The output is applied to the next filter 8 ₂ . This is repeated. The figure shows the output e of 7 _B3 after three relocation diffusion processes.
_{The case where EF3} is obtained and added to the analysis / decision circuit 7 _C is shown. Since the three rearrangement spreading circuits repeatedly spread except the desired signal component e _R , each (S
/ N) _E1 and (S / N) _E2 (S / N) _E3 improve the SN ratio. Noise component in the signal e _F2 and e _F3 is because they contain only very close component on a frequency component of the detected signal, in the 7 _B2, 7 _B3 to spread by further rearrange them,
It is particularly effective to adopt a rearrangement algorithm that takes the time difference between the incoming sample value and the rearrangement destination as long as possible. In this case, since the output of 7 _{B in} the final stage substantially includes only the frequency component to be detected, it can be determined by simply detecting the magnitude of power without performing DFT analysis. That is, the filter 8 ₃ and the power discriminating circuit may be used instead of 7 _C.

7 Sample Number Increasing Method As described in the equations (30) and (31), the sampling frequency f _S
Directly affects the increase in the S / N ratio improvement amount. But,
The sampling frequency that can be realized by an actual sampling circuit has a technical upper limit fsm. However, the sample value between adjacent sample points can be determined more accurately by the sampling theorem. Now, assume that the band of the input signal e _R * of 7 _A in FIG. 4 is the frequency component f _{R of the} signal e _R (t).
(Actually, f _R + kf _D is required as described later)
It can be limited to: FIG. 29 shows that when such f _R is given, when sampling is performed while preserving components below f _R ,
It is a figure explaining the relationship between the band limitation characteristic G (jω) of an input signal (ω = 2πf) and the sampling frequency. Now, let us assume that f _R is selected as the value of the following equation for a plurality of frequency components f _i of the transmission signal waveform. _{f R = (f i) max} (62) relative to the f _R, the sampling frequency placed and _{f SJ = 2f R (63)} , cut-off frequency characteristics [G (j indicated by G _j in FIG. 29
ω) / T], this is the case for the ideal filtering characteristic. The impulse response in this case is given by the sampling function of the following equation from the inverse Fourier transform of G (jω).

G _j (t) = (sin πt / T) / (πt / T) (64) T = 1 / f _SJ (65) The i-th sample value sampled every sampling period T _S is a _i
Then, the original received signal waveform before sampling is a _i and g _j (t)
Using,

[0075]

[Equation 10] Can be restored from. In the above equation, the actually required integration range corresponds to the accuracy of the restored waveform. However, g _j (t) is 1 /
Since the vibration waveform is attenuated at t, the range of i to be added is remarkably large. However, normally, f _S >> f _R can be selected, so that using the G _H cutoff characteristic of FIG.

[0076]

[Equation 11] q = Δf _H / (f _H −Δf _H ) (68) T = 1/2 (f _H −Δf _H ) (69) G _H is the cosine descending characteristic, and G _H (jω) / T = 1
It is bilaterally symmetric with respect to the frequency (f _H −Δf _H ) that is / 2. In this case, g _H (t) is attenuated by 1 / t ³ , and therefore, when e _R * (t) is obtained from the equation (66), the range of i to be added has an actually required accuracy. Also depends, but is significantly smaller than with g _j (t). Since the frequency range of G _H (jω) is f _H ,
The sampling frequency f _{SH in} this case is f _SH >> 2f _H (= 6f _R ) (70) from the sampling theorem, but the case where f _SH = 6f _R is shown in the figure. Here, since e _R is a waveform in which the sine wave is limited to the section of the frame period T _D , f _R is insufficient as the bandwidth actually required to transmit the waveforms at the beginning and the end of this section with sufficient accuracy. Thus, f _R + kf _D (where k> 2). Instead of f _R for transmission now sufficiently accurate for example f _R ^'= f _R
If + 10f _D is selected, a sufficiently accurate restored waveform can be obtained.
In this way, the original waveform can be easily restored by calculation using the sampled values a _i and g _H (t). Therefore, a large number of interpolated sample values existing between the adjacent sample values a _i and a _{i + 1} can be obtained from this original waveform. If the calculation accuracy is high enough, the number of interpolated sample values can be increased infinitely. Therefore, the SN ratio improvement amount can be increased infinitely.

8 Synchronization Method In describing the various methods of the present invention, the description has been given on the premise that the receiving side can secure the synchronization regarding the carrier and the frame position. It is essential for realizing the above-mentioned rearrangement spreading function that the receiving side can provide accurate frame position information of the received signal e _R * (t) including noise. The present invention can also be applied to secure this synchronization. FIG. 30A shows an embodiment when the present invention is applied to the synchronization system. f _T0 is a transmission reference frequency, 9 is a frequency divider, 10 is the transmission circuit (1 and 2) of FIG. 4, 11 is an addition circuit, 3 is a transmission line, 12 and 13 are filters,
Reference numeral 14 is a synchronizing signal extraction circuit, 15 is a multiplication / divider, and 16 is a receiving circuit for demodulating and identifying a multi-level FSK similar to that shown in FIG. (B)
Is a detailed view of 14 in (a), and 7 _{A to} 7 _C and 6 are the same as those in FIG. f _R0 is a reception reference frequency, and 17 is a voltage controlled oscillator.

(A) will be described. 9 frequency fi necessary for transmission of information e _I by dividing the f _T0 (i
= 0, 1, ... N-1), the carrier frequency f _C and the synchronization reference frequency f _y are created. Signal e _y (f _y ) of frequency f _y
And the information transmission signal e _TC is added at 11 and by modulating the carrier wave f _c ,

[0079]

[Outside 16] (Hereinafter, in the specification excluding the drawings, it is expressed as _eTC ^# )
And sent to 3. Here, it is assumed that the frequency bands of e _TC and e _y do not overlap each other. Received signal

[0080]

[Outside 17] (Hereinafter, in the specification excluding the drawings, it is expressed as e _RC ^# )
It is one in which noise e _N is applied to the e _TC ^#. e _RC ^# is separated by the filters 12 and 13 and becomes the two components of the following equation. e _y * = e _y + e _Ny (71) e _RC * = e _RC + e _NRC (72) e _Ny and e _NRC are noise components. The e _Ny in the e _y * is removed by the circuits 7 _{A to} 7 _{C in} FIG. Therefore, the output of the frequency f _y should appear as an extremely large value in the analysis output e _A as compared with the outputs of other frequencies. However, the frame period T _D of the sampled value of the received signal sampled at 7 _A is determined by f _R0 .
However, at the time of starting the operation, since f _R0 ≠ f _T0 (73), the frame period of the transmitting side and that of the receiving side do not normally match, so e _A produces outputs at many frequency points other than f _y. Become. 6 to identify and f _R0 <f _T0 or f _R0> f _T0 by examining the components of such output e _A, to create a control output e _B. Controls 17 built-in oscillators with e _B. Due to the function of this phase control oscillation loop, f _R0 → f _T0 , and if f _y = f _D is selected in advance, as a result, the transmission / reception frame period can be made to exactly match.
In this case, the function of the 7 _B rearrangement diffusion realizes accurate negative feedback control under a high SN ratio, so that the transmission / reception frame time widths can be exactly matched. The reference frequency f _R0 thus obtained is supplied to 15. 1
5 multiplies and divides f _y to reproduce _N signal frequencies f ₀ ^′ , f ₁ ^′ , ... F _N−1 ^′ and a carrier frequency f _c ^′ synchronized with the frame frequency f _D. These, including the phase, match the frequencies f ₁ , f ₂ , ... F _N-1 , f _c on the transmitting side.
With these frequencies 16 can demodulate the incoming signal e _RC ^* and determine the logical value.

If the transceiver moves relatively, e _RC ^# is affected by the Doppler shift, so that the reception reference frequency changes by the Doppler frequency f _d . f _R0 = f _T0 + f _d (74) However, since similar Doppler shift occurs between the transmission / reception signals e _TC and e _RC , the reception operation is normally performed. That is, the synchronization signal e _y is added to the information signal e _RC and transmitted, as shown in FIG.
The method of 0 has an advantage that the influence of Doppler shift can be avoided.

From the above description, it has been clarified that the frame period (time width) of transmission and reception can be matched. But,
When f _y > f _D is selected, the frame boundary (time position) is also necessary for the receiving operation. This boundary is identified by a known technique (frame synchronization technique) in which the transmitting side periodically inserts a synchronization frame into the information signal and the receiving side detects it. Incidentally, created using e _y process the frame information, for example, PSK instructing the boundaries of the product, which is harmless if in the e _y, detects a time position of a frame in e _y * analysis process on the reception side It is also possible.

9. Overlapping Transmission Method Multilevel transmission has been described in the second and third sections. That is, the type of transmission signal is e _Ti (i = 0, 1, 2, ... N-1)
Then, the amount of information given by equation (42) can be sent for each frame. FIG. 31 is an explanatory diagram of a method in which such multilevel transmission is further generalized. Where I _j (j = 1, 2 ...
*) Indicates that the type of information sent using one frame is the j-th. The figure (a) shows the perfect duplication transmission method which sends two frames in duplicate in the frame period T _D , and the figure (b) is followed by the next frame with a delay of ΔT = T _D / 3 seconds, The partial overlap transmission method in which three frames are redundantly sent is shown.

In the complete redundant transmission method, if the number of redundant frames is generally k, the amount of information that can be sent for each frame is I _a = log ₂ ( _N C _k ) (bit) (75). On the other hand, the amount of information that can be sent for each frame by the partial overlap transmission method is given by I _b = hlog ₂ N (76) h = T _D / ΔT (77). Compared to the method of sending one frame, the required power of these methods is k or h times, but there is an advantage that the transmission speed can be increased.

When such duplication is performed in a normal transmission method, the (S / N) _R on the receiving side shown in FIG. As a result, it generally decreases significantly. Therefore, the number of errors increases,
It will be difficult to realize. However, if the information of the rearrangement diffusion of the present invention is used, even if the information is duplicated, if the information is not the same, then FIG.
Since the (S / N) _A can be remarkably increased, the judgment S / N ratio is not impaired. Therefore, a reliable transmission method can be realized.

10 Phase Shift Spreading Method Consider the case where the difference Δf = f _N −f ₁ (78) between the transmission frequencies f ₁ and f _N corresponding to the logical value “1” is small in the FSK modulation method described above. See. When brought close to the logical value "1" f ₀ to f _1, or the frequency f ₁ near the other transmitting station to f ₁ ', f _0' when communicating with another station using f _0, f ₁ ^', f
Any ₀ ^'and the like is also approaching the f _1. These are generally expressed as f _N , and the case of Δf → 0 in Expression (78) is considered. Δf
→ By setting to 0, the frequency utilization efficiency is improved, but f
_It becomes difficult to distinguish between ₁ and f _N. As described above, when the noise frequency component extremely close to f ₁ is rearranged and diffused by the above-described method, the same output as f ₁ is obtained as the DFT output unless N _E is made sufficiently large. This causes a misjudgment. The equal-level point rearrangement method of the first and second terms is such that even if the frequency of the noise signal is close to the signal frequency f ₁ to be detected or f _N = f ₁ , if the phase difference between them is sufficiently large, the noise Has a characteristic that can be strongly diffused to other frequency components. An embodiment utilizing this characteristic is shown in FIG.

FIG. 32 shows an embodiment of the phase shift type spread system, in which the received signal e _R * = e _R + e _N containing noise is added to N _T phase shift filters (T-FIL) and its output depends on the frequency. After giving a delay, a rearrangement spreading process is performed, and N _T or more of the delay dispersion frames thus created are concatenated to form an expanded frame. The arrangement of the sampled values is changed under the condition that the signal waveform e _R is stored between the delay dispersion frames which are the constituent elements of this enlarged frame, and the output is analyzed and judged. FIG. 32 shows an example when N _T = 4. The kth T-FIL provides delay times τ _Rk and τ _Nk for e _R and e _N. 33 (a), (b)
[Fig. 3] is a diagram showing characteristics regarding a phase (θ) and an amplitude (P) of a typical filter, respectively. When an appropriate attenuation characteristic is given, the phase characteristic greatly changes, and in the figure, the delay time is shown by a value obtained by differentiating θ in the figure with a large difference. Delay time τ for frequencies 0.5 and 1 kHz, for example
Considering _0.5 and τ ₁ as τ _0.5 → τ _Rk and τ ₁ → τ _Nk , respectively, the relative delay time is Δτ _k = τ _Rk − τ _Nk = τ _0.5 − τ ₁ (79), and this Δτ _k is It is changed by k.

On the other hand, when a delay dispersion frame is _created , a constant delay τ _Ak that does not depend on frequency is applied to the frame signal, and the sum is a constant value τ _c + (k−1) T _D = τ _Rk + τ _Ak (80) To select τ _Ak . Then, a common relative delay amount τ _c is given to the signal e _R having the frequency component f ₁ in all delay dispersion frames. Where (k-
1) T _D is a delay that does not overlap with each other when these outputs are added to the next stage to create an enlarged frame. In each delay-dispersion frame constituting the expanded frame created in this way, the phases of the e _R components therein are equal to each other, and the phase of the e _N components is Δτ _k is different depending on k,
Different from each other. In this expanded frame, in the delay-dispersed inter-frame rearrangement stage of FIG. 32, the sampled value of each delay-dispersed frame is repositioned to another delay-dispersed frame (such as the position of the signal) under the principle of signal waveform preservation. If relocated to the level point), e _N undergoes a large waveform change.

The frequency-dependent delay as shown in FIG. 33 can be easily realized by a digital filter or an analog filter (in this case, the filter output is sampled). In particular, if the Q of the filter is increased, the delay time difference can be increased even if Δf is small, so Δf → 0 can be dealt with. Further, although FIG. 33B shows the case of the reduced pass type filter characteristic, a band pass type filter characteristic centered on the frequency f ₁ to be detected can be used. Further, in order to filter the constant delay τ _Ak or (k-1) T _D , the sample value of the output of T-FIL can be temporarily stored in the memory and read after a desired time.

In the circuit of FIG. 32, equation (80)
Instead of, τ = τ _Rk + τ _Ak (81), the phases of the e _R components in each delay dispersion frame are made equal to each other, and the sampled values at equal absolute value level points are set between these delay dispersion frames. Are mutually exchanged (or rearranged), and then (k−
1) An extended frame may be created by adding and adding a delay of T _D. In other words, this is a method in which the function of rearrangement between delayed dispersed frames is moved immediately after the stage of creating a delayed dispersed frame, and a function equivalent to the above-described method can be realized. It should be noted that although the spreading effect is reduced, the rearrangement process can be performed by limiting the sample value to within the delay dispersion frame in the above-mentioned two methods.

In this way, by rearranging the absolute value equal-level sample values between the delay dispersion frames, the waveform of the e _N component changes greatly, and efficient spreading can be realized. Now, e _R = Asin 2πf ₁ and _{t (82) e N Asin (} 2πf N t + θ N) (83), 4 for detecting the f _1, a f _N added to the circuit of Figure 32, the frequency generated in the DFT analysis output Let e _A (f ₁ ) = A′sin (2πf ₁ t + θ _N ′) (84) be the component e _A (f ₁ ) equal to f ₁ among the components. Then, the SN ratio focused on f ₁ is given by (SN) _Ef1 = 201 _og10 (A / A ′) (85). The SN in the above equation is a value when the SN improvement amount according to the method of the present invention is limited to the same frequency component as the noise f ₁ in the equation (10). FIG. 34 shows the result of computer simulation in which this value is obtained for the incoming noise component f _N = (3 to 5) f _D close to f ₁ . Figure is a result when the limited relocation destination within the delay dispersion frame but, f ₁
It is shown that an extremely large SN improvement amount can be obtained by using this method even in the vicinity of. Further, as e _N , not only the general noise frequency component f _N but also the adjacent frequency components f ₀ and f ₁ ^{′ of} other information signals and interference signals can be strongly spread. Therefore, even if Δf is small, it is possible to cope with the situation, so that the frequency utilization efficiency can be remarkably enhanced. Although the present invention is applied to the communication system as an example,
The present invention is not limited to this, and can be applied to a transmission system in a high noise environment.

The application objects other than the communication system are shown below. (1) Distance measurement (high-accuracy distance measurement using ultrasonic waves or X-rays, diagnostic equipment, radar, etc.) (2) Reading recorded information (identification of minute read output of magnetic tape, magnetic disk, optical disk, etc.) (3) Electronic Microscope (identification of minute detection output) (4) Measuring instrument (accurate frequency characteristic measurement, detection of pure sine wave) (5) Atomic physics measurement (measurement of lattice vibration frequency of atoms) (6) Space electromagnetic wave Measurement (measurement of electromagnetic waves and gravitational waves emitted from the celestial body) (7) Digital broadcasting (narrow bandwidth of digital video, low power consumption)

[0093]

Since the present invention is configured as described above, the received signal e _R waveform corresponding to the transmitted signal e _T is stored for the received signal e _R * containing noise, and the noise waveform is stored. By spreading, the determination SN ratio of discrete information can be significantly increased. The signal waveform saving and noise waveform changing functions described above enable signal spectrum saving and noise spectrum spreading. Therefore, the analysis and determination of the rearrangement / spreading process output becomes extremely easy. The relocation diffusion process described above
This is equivalent to encoding the received signal of the original frame, and is a function of generating the sample value of the enlarged frame using the sample value of the original frame. Here, the sample value of the original frame can be used any number of times. Also, generally, the original sample value is used as it is, or the converted sample value is obtained based on the original sample value,
These can be used to determine the placement sample value and the placement time position. The noise spreading function can be further enhanced by performing rearrangement after giving different delays corresponding to the slight frequency difference between e _R and e _N. Such a rearrangement / diffusion function does not exist in the past, and provides a completely new principle. This function has the ability to diffuse noise virtually infinitely and expand the noise component overlapping with the signal component to 0 infinitely by increasing the scale of the enlarged frame and the number of original sample values. Therefore, the determined S / N ratio approaches infinity. The effect of increasing the SN ratio enables the occupied band of the transmission path signal to be reduced, so that the transmission efficiency (transmission capacity / Hz) of the equation (2) is obtained.
There is also an effect that can significantly increase. As described above, the effect of the present invention is remarkable.

[Brief description of drawings]

FIG. 1 is a block diagram of a conventional digital modulation / demodulation communication system.

2A and 2B are diagrams showing an example of the signal of FIG. 1 and a time waveform diagram of each part of FIG.

FIG. 3 is a block diagram of a conventional digital modulation / demodulation communication system using spread spectrum technology.

4A and 4B are diagrams showing the configuration of an embodiment of the present invention.

FIG. 5 is a supplementary drawing of FIG. 4, showing a time waveform corresponding to a logical value “1”.

FIG. 6 is a diagram showing a time waveform corresponding to a logical value “0”, similar to FIG. 5.

FIG. 7 is a diagram showing a time waveform by rearrangement.

FIG. 8 is a diagram showing a result of computer simulation using a spread spectrum by rearranging waveform units.

9 (a), (b) and (c) are DFs of rearranged and spread signals.
The figure which shows the modeled characteristic of the analysis output electric power by T.

10 (a), (b) and (c) are diagrams showing a sine wave equi-level point and an example of rearrangement thereof.

FIG. 11 is a diagram showing a result of computer simulation with a DFT output when a signal to be detected is rearranged by equal level conversion.

FIG. 12 is a diagram showing a result of computer simulation with DFT output when noise is rearranged by equal level conversion.

FIG. 13 is a diagram showing a result of the same computer simulation as in FIG.

FIG. 14 is a diagram showing a rearrangement example by functional conversion of a sample value of a received signal waveform e _R (t).

FIG. 15 is a DF when noise is rearranged by function conversion.
The figure which shows the result of computer simulation by T output.

FIG. 16 is a diagram showing a received waveform e _R (f ₁ ) and a decompressed frame e _RE (f ₁ ) created based on the received waveform e _R (f ₁ ).

17 (a), (b) and (c) are diagrams showing spectra when the conventional binary FSK system, the binary FSK of the present invention and the same eight-level FSK modulation are respectively performed.

FIG. 18 is an explanatory diagram of a rearrangement effect regarding signals having different phases.

19 (a) and 19 (b) are explanatory views of the case where the phase of a sine wave is transmitted in association with a fourth-order Hadamard matrix.

FIG. 20 is an explanatory diagram of the principle of the chirp modulation-DFT method.

FIG. 21 is an explanatory diagram in the case of receiving determination of a signal transmitted by non-sinusoidal waveform modulation.

22 (a) and 22 (b) are explanatory views when the equal absolute value level point rearrangement method and the function conversion method are applied to a general waveform.

23 (a) and 23 (b) are views for explaining analysis and determination by correlation calculation.

Figure 24: PSK with minimum shift keying
Explanatory drawing for converting a transmission waveform into normal PSK.

25A and 25B are diagrams showing the configuration of another embodiment of the present invention in which a code spreading function is added to the configuration shown in FIG.

26 shows an example of a time waveform e _T of e _T in Fig. 25 (t).

27 is a diagram of D corresponding to noise added in the transmission line of FIG.
The figure which shows the result of computer simulation by FT output.

FIG. 28 is a block diagram of a rearrangement spreading iterative method.

FIG. 29 is a diagram illustrating a relationship between a band limiting characteristic G (jω) of an input signal and a sampling frequency.

30 (a) and 30 (b) are diagrams showing an example in which the present invention is applied to a synchronization system.

FIGS. 31 (a) and 31 (b) are explanatory views of a complete and partial overlap transmission system.

FIG. 32 is a diagram showing a configuration example of a phase shift type diffusion system.

33A and 33B are diagrams showing phase and amplitude characteristics of a phase shift filter.

[Explanation of symbols]

1 ... Baseband modulator of transmitter 2 ... Final stage modulator by carrier wave (radio frequency) 3 ... Transmission line (wired, wireless) 4 ... First stage demodulation circuit of receiver 5 ... Baseband demodulator 6 ... Logical value determination circuit 7 ... Rearrangement spread processing circuit (rearrangement demodulation circuit) 8 ... Filter 9 ... Divider 10 ... Transmitter circuit (1 in FIG. 4 And 2) 11 ... Adder circuit 12, 13 ... Filter 14 ... Sync signal extraction circuit 15 ... Multiplier / divider 16 ... Receiver circuit (7 _{A to} 7 _C ) of FIG.

[Equation 5]

[Equation 6]

[Equation 9]

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Naoki Suehiro 3-6-305-103 Takezono, Tsukuba, Ibaraki Prefecture (72) Inventor Toshikatsu Naito 2-1-1 Kotani, Samukawa-cho, Takaza-gun, Kanagawa Toyo Communication Equipment Co., Ltd. In the company

Claims (15)

[Claims] 1. The transmitter allocates any one of the N signals A _Tj (j = 1, 2, ... N) to a frame having a certain time width, and then assigns it as a transmission signal. Send it to the transmission line,
The receiver receives the sum of the component A _Rj corresponding to the transmission signal and the noise e _N added on the transmission path, and performs demodulation to determine from the received signal that the transmission signal is A _Tj. In, the receiver is equipped with N circuits composed of a rearrangement spreader corresponding to each _ATj and an analysis determination unit, and the rearrangement spreader collects original sample values of the received signal for each received frame. , The time waveform of the transmission signal corresponding component A _{Rj in} the original sample value is stored, and the time waveform of the noise e _N is changed so that the rearrangement destination time position and the rearrangement destination time position are changed based on the original sample value. A time-position conversion process for creating an arrangement sample value is performed, and an enlarged frame in which the time width of a received frame is enlarged is formed. The analysis / determination unit determines the enlarged frame to include components A _Tj and A _Rj related to a transmission signal. Correlation system created in advance based on By analysis using the transmission signal A _Tj
A rearrangement diffusion type communication system characterized in that a large output is obtained only when 2. After sampling the original sample value of the received signal, the sampling period T _S of the original sample value and the received transmission signal corresponding component A _Rj
The sampling position is determined by using the sampling function determined by the occupancy band and the interpolated sample value between two adjacent original sample values is calculated, and the interpolated sample value is added to the original sample value, and then the time position conversion processing is performed. The rearrangement / diffusion type communication system according to claim 1, wherein 3. The amplitude of the waveform of the transmission signal corresponding component A _Rj is i
(I = 1, 2, ... M) Based on the sample values of the original sequence S _i0 composed of N _i sample values on the received frame corresponding to the absolute value-th level equal level V _i , By _performing a time position rearrangement conversion process of moving a certain sampled value of S _{i0 to} a time position of another sampled value of S _i0 , k (k = 1, 2) whose rearrangement patterns are different from each other including no conversion.・ ・ ・ N
_E) th give conversion and level sequence S _ik, as A _Rj does not _{change, S 1k, S 2k, ···} S MK give relocation frame sequence S _K by placing a different k S 3. The rearrangement spreading type communication system according to claim 1, wherein a sampled value sequence of an expanded frame having a length N _E times the received frame length is obtained by sequentially arranging _k . 4. The frequency f in the process in which the rearrangement / spreading unit creates the sample value of the conversion sequence by moving the original sample value of the received signal for each reception frame period to another sample point.
Phase angle point θ ₁ (= 2πft) corresponding to time t ₁ of the original sample sequence of the reception-side sine wave signal corresponding to
₁₎ by using the original sample values b ₁ = cos [theta] _'1 of the original sample value a ₁ = cos [theta] ₁ and θab radian phase delayed angle point _{θ' 1 = θ 1 -θab (} = 2πft 1 ') of similar Relocation destination phase angle point set θ ₂ = θ ₁ + Δθ = 2πft ₂ , θ ′ ₂ = θ ₂ −θab
= 2πft ₂ ′ converted sample value a ₁ = cos θ ₂
And a function conversion method for obtaining b ₂ = cos θ ₂ ',
Generally, original sample values (a ₁ , b ₁ ) at arbitrary times (t ₁ , t ₁ ′)
) Is converted to a conversion sample value (a ₂ ′ b ₂ ′) at an arbitrary relocation destination time (t ₂ , t ₂ ′), and the time (t ₂ , t _2
2 ′), the conversion sequence different from the received sample value sequence of the original frame is obtained without changing the waveform of the transmitted sine wave signal, and the expanded frame is obtained by using the frames composed of such different conversion sequences. 4. The rearrangement diffusion type communication system according to claim 3, wherein 5. A filter means for removing only the component A _Rj corresponding to the transmission signal A _Tj of the received signal, and a rearrangement diffusion process for the output of the filter means. 2. The rearrangement diffusion type communication system according to claim 1, wherein an expanded frame is obtained by arranging a plurality of frames. 6. When the transmission signal A _Tj is a sine wave and the number of cycles X included in one frame is a non-integer, a part of the sampled values in the reception frame is used in duplicate or a part of the sampled values is used. 6. The rearrangement diffusion unit according to claim 1, wherein an expansion or contraction frame including a cycle of an integer multiple is configured by deleting the sampled value, and then input to the rearrangement diffusion unit. Form communication system. 7. The output of the rearrangement / spreading unit is set to a transmission signal A _Tj.
Is applied mainly to the filter means for removing other components, and its output is applied to the second rearrangement diffusion section having the same function as the rearrangement diffusion section, The output obtained by repeating the application of the output of the rearrangement diffusion unit to the second filter unit having the same function as the filter unit a plurality of times is applied to the analysis determination unit. 7. The rearrangement diffusion type communication system according to items 1 to 6. 8. A received frame signal (A _R containing a noise component
In + e _N) process constituting the relocation spreading frame the original sample values rearranged diffused to the original, (A _R + e _N) previously _{k (= 1,2, ··· N T} ) th phase shift in addition to the filter, a _R and e _N giving the respective delay times tau _RK and tau _NK to a constant value irrespective of the delay time k for a _R tau _C
As a, A _R + e _N both in the correction delay tau _AK of (= tau
_{C −} τ _RK ) composed of the corrected output sample value series k
Form the th delay-dispersed frame, and generally the relative delay Δτ
_k = τ _RK −τ _NK is selected as a different value depending on the value of k, an expanded frame is configured by using a plurality of the delay spread spectrum frames, and a sample value at an arbitrary time position of the expanded frame is used as the received signal. The rearrangement diffusion according to any one of claims 1 to 7, characterized in that a rearranged image at another time position of the enlarged frame is applied to the analysis / determination unit under the condition that the time waveform of (1) is stored. Form communication system. 9. The received frame is divided into a plurality of time regions S _i.
The original sampled value is divided into (i = 1,2 ...), and the sampled value of the region S _i is subjected to amplitude conversion, polarity inversion, and time axis inversion processing to obtain another region S _i ^{′ having} a different _i . A rearrangement frame is created by converting the sampled value into the sampled value and arranging the converted sampled value at the time position of the area S _i ^' , and a plurality of similar relocated frames having different conversion patterns are created, and an enlarged frame is created using these. The rearrangement / diffusion type communication system according to claim 1, wherein 10. The sample value series of the expanded frame is subjected to DFT (spread Fourier transform) analysis in the analysis determination unit, and a transmission corresponding frequency component output of the DFT output corresponding to a transmission signal A _Tj is output. And the transmission signal is determined to be _ATj based on the ratio of the frequency component corresponding to the noise mixed in the transmission process to the noise component having the same frequency as the transmission corresponding frequency. Claim 1
9. The rearrangement diffusion type communication system described in 9 above. 11. A reception side demodulation circuit of a communication system for transmitting a signal in which discrete information is modulated by a predetermined modulation system, wherein a reception signal to be detected is converted into a frame signal composed of a sine wave of equal amplitude and a single frequency. A rearrangement diffusion process and an analysis determination process are performed after performing a conversion process on the amplitude axis or the time axis on the sample value of the incoming signal containing noise. 11. The rearrangement diffusion type communication system according to any one of items 1 to 10. 12. A plurality of transmission signals A having different time waveforms.
_{12. The} rearrangement spread type according to claim 1, wherein transmission of multi-bit discrete information per unit time is made possible by sending _Tj simultaneously or with a time shift from a transmitter to a transmission line. Communication method. 13. A spread spectrum (SS) communication system in which discrete information is made to correspond to a time waveform, the time waveform is multiplied by a spreading code sequence to be transmitted, and the received signal is multiplied by the spreading sequence to despread. 13. The rearrangement spread communication system according to claim 1, wherein an original sample value is sampled from the spread signal, and the rearrangement conversion process and the expansion frame are created based on the original sample value. 14. In a reception demodulation circuit of a communication system for transmitting discrete information by predetermined modulation, the repetition cycle is T _i.
= 1 / f _i A sample value is extracted from the received signal, and if necessary, rearrangement conversion processing or filter processing is performed, and then nonlinear processing on the amplitude axis or time axis is performed on the sample value. by applying, by modifying the waveform to a particular waveform, an integral multiple of the frequency components of the results f _i kf _i (k
= 1, 2) generates a noise component in the output widely allowed diffused into the frequency domain that does not agree with kf _i This nonlinear processing, subjected to rearrangement conversion process if necessary these output 14. The rearrangement spread type communication system according to claim 1, wherein a target waveform is identified later in addition to the analysis unit. 15. In the process of constructing the expanded frame, a sample value at an arbitrary time position in the expanded frame is re-created at another time position in the expanded frame under the condition that the waveform of the received signal A _Rj is stored. The rearrangement / diffusion type communication system according to claim 1, characterized in that the arrangement process is performed.