JPH0685859A - Burst demodulator - Google Patents

Abstract

PURPOSE:To allow the demodulator to cope with high speed transmission processing by adapting the storage simultaneous demodulation system by a low speed sample, thereby relieving the load without deteriorating estimated accuracy. CONSTITUTION:An A/D converter 2 samples and quantizes a burst mode modulation wave subject to quasi-synchronization detection by a quasisynchronization detector 1 from a digital phase modulation burst reception signal 11. A carrier estimate means 4 estimates a carrier to correct a frequency and a phase to a sample point data signal 12 stored once for each burst in a buffer memory 3. An interpolation timing generating means 5 applies arithmetic operation to the corrected data signal 13 to estimate a Nyquist point, and generated interpolation timing information sets 14, 15 are given to a data correction means 6, which applies interpolation calculation to the correction data signal 13 to estimate Nyquist point data and to generate tentative demodulation data series 16, 17. A timing decision means 7 compares, discriminates and selects the accumulated the absolute values or the squared absolute values of data levels of the tentative demodulation data series 16, 17 for each burst, and the resulting demodulation data series 18 is discriminated by a data discrimination means 8, which outputs a demodulated data signal 19.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a burst demodulator for improving the characteristics of a collective batch demodulation system for demodulating a digital phase modulation burst reception signal.

[0002]

2. Description of the Related Art For example, literature (Honda et al .: Study on computational demodulation method of PSK signal, IEICE Technical Report, CS87-10).
9, 1987), the conventional burst demodulator shown in FIG.
As described above, the quasi-synchronous detector 1 performs quasi-synchronous detection on the digital phase modulation burst reception signal 11 with an asynchronous reference carrier having a carrier approximate frequency. The analog / digital converter 2 uses a high-speed clock (for example, about 16 samples or more per symbol period) for the burst mode modulated wave after detection from the quasi-synchronous detector 1 and the reference carrier difference frequency.
Sampling and quantizing operations are performed in. The buffer memory 3 temporarily stores all sample point data for each burst after waveform shaping by the reception filter from the analog / digital converter 2. The digital signal processing demodulator 10 extracts the optimum phase point of the sample point data signal 12 from the buffer memory 3 to determine and output the demodulated data signal 14.

The above-mentioned conventional burst demodulation device adopts a collective storage demodulation method for extracting an optimum phase point by high-speed sampling in order to reduce a phase estimation error.

As shown in FIG. 10, the digital signal processing type demodulator 10 first estimates the carrier wave from the data sequence of the sample point data signal 12 from the buffer memory by the carrier wave estimating means 4 and corrects the frequency and phase of the data. To do. Next, only the data of the optimum phase point (estimated sample point most recent to the Nyquist point) estimated by the timing estimating means 7a is extracted from the data signal whose frequency and phase have been corrected by the sampler 7b. Further, it is judged by the data judging means 8 and outputted as a demodulated data signal 19. The timing estimating means 7a estimates the optimum phase point as follows. First, in the case of two-phase phase shift keying (hereinafter referred to as BPSK), FIG.
When the BPSK received signal is sampled 1 time per symbol as described above (the solid line shows the waveform of the received signal with the amplitude a and the symbol period T. d represents transmission data (-1 or 1), where 1, 1 , -1, 1, a series of transmission data consisting of y, y represents a sampled signal, only y ₁ and y ₂ are shown here, and the sampling point is τ (0 ≦ τ) from the original Nyquist point. The sampled signal y is represented by the sum of the signal component and the intersymbol interference component at that time, assuming that they are shifted by <T). If there is an influence, the impulse response of the reception filter can be expressed as follows: h (τ) = sin (πτ / T) ・ cos (πατ / T) / {(πτ / T) ・ (1-2ατ / T )} alpha: roll-off rate for example y ₁ of the receive filter is y ₁ = d ₁ a _{(Τ) + d 2 ah (} τ + T) + ... + d
_{k + 1} ah (τ + kT) + d ₀ ah (τ−T) + ... + d
_{-k + 1 ah (τ-kT} ) becomes, m th sampled signal y _m in the same manner as is generally _{y m = Σ l d m +} 1 ah (τ + l T) -k ≦ l ≦ k τ = It can be expressed as (n-1) T / X. Further, when the BPSK received signal is X-sampled per symbol as shown in FIG. 12, if the initial phase difference is set to 0 for simplification, then n (1 ≦ n
≦ X) th received signal y _m is generally _{y m = Σ l d m +} 1 ah {(n-1) T / X + l T} -
It can be expressed as k≤l≤k. Now, assuming that the number of symbols in one burst is L, the n-th cumulative value Pn obtained by accumulating the absolute square value of the amplitude of the n-th sample in each symbol is Pn = Σ _m | Σ _l d _{m + 1} ah (τ + l T ) | ² 1 ≤ m ≤ L, -k ≤ l ≤ k. Here, d _j (j ≦ 0, L + 1 ≦ j) does not exist, but d _j (j ≦ 0, L + 1 ≦
Let j) be a pseudo noise (PN) pattern. Next, in the case of four-phase phase shift keying (hereinafter referred to as QPSK), since there are two series of in-phase and DC components and they are independent of each other, Pn obtained in the same way as in the case of BPSK is Pn = Σ _m [{Σ _{l d m + 1 ah (τ} + l T)} 2 + {Σ l d ■ m + 1 ah (τ + l T)} 2] 1 ≦ m ≦ L, the -k ≦ l ≦ k. Here, d and d ■ represent data series of in-phase and quadrature components, respectively. And amplitude a = 1, when L is sufficiently large data is random, the above equation simply ^{_{Pn = 2LΣ l h 2 (τ}} + l T) = 2LΣ l h 2 {(n-1) T / X + l T} -k ≦ l
≤k. In the above equation, Pn is maximum when τ = 0 (n = 1), so Pn is maximum when the Nyquist point is sampled, but the initial phase difference is not always 0 in actual operation. The estimated sample point closest to the Nyquist point is estimated as the optimum phase point.

[0005]

In the conventional burst demodulation device as described above, the accumulated batch demodulation for extracting the optimum phase point by high-speed sampling (for example, about 16 samples or more per symbol period) in order to reduce the phase estimation error. Since the system is adopted, the analog / digital converter needs to operate at high speed when the transmission speed becomes high, the buffer memory has an enormous amount of accumulated data, and the digital signal processing type demodulator has a problem that the amount of calculation becomes large. .

An object of the present invention is to provide a batch accumulation demodulation system for extracting an optimum phase point by low-speed sampling so that the burst demodulator can reduce the load and reduce the phase estimation error. .

[0007]

In order to solve the above-mentioned problems, a burst demodulation device of the present invention is an analog / digital converter for converting a burst phase modulated wave obtained by quasi-coherent detection of a digital phase modulation burst reception signal by a quasi-coherent detector. Sampling and quantizing operations are performed with, and the optimum phase point is extracted by the digital signal processing type demodulator from the sample point data signal that has been temporarily stored for each burst in the buffer memory, and the demodulated data signal is judged and output. The digital signal processing type demodulator is provided with the following means, and is characterized by adopting a storage batch demodulation method for extracting an optimum phase point by low-speed sampling.

The carrier wave estimating means estimates the carrier wave from the data series of the sample point data signal from the buffer memory and corrects the frequency and phase of the data.

The interpolation timing generating means corresponds to the carrier power to noise power ratio information measured directly or by providing the carrier power to noise power ratio measuring means and obtaining the average value and fractional value of the correction data signal from the carrier estimating means. Then, the correction data signal from the carrier wave estimating means or the sample point data signal from the buffer memory is operated to estimate the Nyquist point and generate a plurality of interpolation timing information.

The data correction means performs an interpolation operation on the correction data signal from the carrier wave estimation means with a plurality of interpolation timing information from the interpolation timing generation means to estimate the Nyquist point data and generate it as a plurality of temporary demodulation data series.

The timing determining means, for each of the plurality of temporary demodulation data series from the data correcting means, the cumulative value of the absolute value or absolute square value of the data amplitude, the absolute variance value of the data amplitude, and its known pattern. A demodulation data sequence is selected by comparing and determining the correlation value between the part and a predetermined known pattern or the correlation value between the soft decision data sequence subjected to the soft decision and the hard decision and the hard decision data sequence.

The data determining means determines the demodulated data sequence from the timing determining means and outputs it as a demodulated data signal.

[0013]

With the above means, the burst demodulator of the present invention corrects the frequency and phase of the data by estimating the carrier from the data sequence of the sample point data signal generated from the digital phase modulation burst reception signal. Next, the corrected data signal is calculated to estimate the Nyquist point and generate a plurality of interpolation timing information. Alternatively, a sample point data signal from the buffer memory may be used instead of the correction data signal. A correction data signal or sample point data signal corresponding to the carrier power to noise power ratio measured from the average value and the variance value of the correction data signal may be used.
Further, for each of a plurality of provisional demodulation data sequences generated by performing interpolation calculation on the correction data signal with a plurality of interpolation timing information and estimating Nyquist point data, the cumulative addition value of the absolute value or absolute square value of the data amplitude, The absolute variance value of the data amplitude, the correlation value between the known pattern portion and a predetermined known pattern, or the correlation value between the soft decision data series and the hard decision data series subjected to the soft decision and the hard decision is compared and determined. The selected demodulated data sequence is determined and output as a demodulated data signal.

[0014]

BEST MODE FOR CARRYING OUT THE INVENTION As shown in FIG. 1, a burst demodulating apparatus according to an embodiment of the present invention includes a quasi-synchronous detector 1, an analog / digital converter 2, a buffer memory 3, a carrier wave estimating means 4 and a data judging means 8. It corresponds to FIG. 9 and FIG. 10 of the above conventional example. The interpolation timing generation means 5 performs an operation on the correction data signal 13 from the carrier wave estimation means 4 to estimate the Nyquist point, and generates interpolation timing information 14 and 15. The data correction means 6 interpolates the correction data signal 13 from the carrier wave estimation means 4 with the interpolation timing information 14 and 15 from the interpolation timing generation means 5 to estimate the Nyquist point data, and the temporary demodulation data series 16 and 17 are obtained. Generate as. The timing determining means 7 compares and determines the value obtained by cumulatively adding the absolute value or absolute square value of the data amplitude of each of the temporary demodulated data series 16 and 17 from the data correction means 6 for each burst, and demodulated data series 18 Is selected and output to the data determination means 8.

The burst demodulation device of the above-mentioned embodiment adopts the batch accumulation demodulation method for extracting the optimum phase point by low-speed sampling.

The interpolation timing generation means 5 is, for example, QP.
At SK, as shown in FIG. 2, the cumulative addition value Pn obtained by performing the calculation on the correction data signal 13 from the carrier wave estimating means 4 is a function of deviation τ (−T / 2 ≦ τ ≦ T / 2) from the Nyquist point. The Nyquist point is estimated with respect to the cumulative addition value P (τ) that is normalized and the interpolation timing information 14 and 15 is generated. P (τ) = 2LΣ _L h ² (τ + LT) / P (0) -k
≤L ≤k For example, 1 sampled by 2 samples per symbol period T in FIG.
The ratio K between the second and second cumulative addition values P1 and P2 is P (3
T / 16) and P (11T / 16) and P (13T /
16) and P (21T / 16) are indistinguishable, so they are outside P1 and P2, and outside P2 by 5T / 16 from the timing point of P2 Inside P1 and P2, and timing of P1 From the point, the Nyquist point is estimated with 3T / 16 on the P2 side, and the interpolation timing information δ ₁
And δ ₂ 14 and 15, δ ₁ = 5/8 and δ ₂ = 3/8 are generated.

As shown in FIG. 3, the data correction means 6 uses δ ₁ and δ ₂ from the interpolation timing generation means 5 to perform linear interpolation (for example, internal division) on the correction data signal from the carrier wave estimation means 4 as shown in FIG. Nyquist point data is estimated by performing a quadratic interpolation (for example, Lagrange method) operation, and is generated as temporary demodulation data series 16 and 17. For example, in FIG. 3, the second of the (m-1) th symbol and the first of the mth symbol
Th sample point data y ₂ (m-1) and y ₁ (m)
The interval is internally divided by δ ₁ : (1−δ ₁ ), and the temporary demodulation data u ₁
(M) is obtained. Similarly, the first and second sample point data y ₁ (m-1) and y ₂ (m
-1) is internally divided by δ ₂ : (1-δ ₂ ) to obtain provisional demodulation data u ₂ (m). Data sequences consisting of u ₁ (m) and u ₂ (m) are generated as temporary demodulation data sequences 16 and 17, respectively. u ₁ (m) = (1-δ ₁ ) × y ₂ (m-1) + δ ₁ × y ₁ (m) u ₂ (m) = (1-δ ₂ ) × y ₁ (m-1) + δ ₂ × y ₂ (m-1)

As shown in FIG. 4, the timing deciding means 7 uses a cumulative adder 71 for each of the temporary demodulated data sequences 16 and 17 from the data correcting means 6 to calculate the absolute value or absolute square value of the data amplitude for each burst. Value S ₁ cumulatively added to
The comparator 72 determines that the larger value of S ₂ and S ₂ is the correct series, and the selector 73 selects one of the temporary demodulation data series 16 and 17 as the demodulation data series 18, and outputs it to the data judging means 8. S ₁ = Σ _m u ₁ (m) or Σ _m | u ₁ (m) | ²
1 ≦ m ≦ L S ₂ = Σ _m u ₂ (m) or Σ _m | u ₂ (m) | ²
1 ≦ m ≦ L

In the above embodiment, the timing determining means 7 has a dispersion calculator 71 instead of the cumulative adder 71 as shown in FIG.
Alternatively, the correlation value D ₁ between the absolute variance values V ₁ and V _{2 of} the data amplitude calculated by the pattern correlator 71 or the known pattern portion of the temporary demodulated data series 16 and 17 and a predetermined known pattern.
The comparator 72 may determine that the smaller value or the larger value of D ₂ and D ₂ is the correct series.

In the above embodiment, the timing decision means 7 has a soft-decision data sequence in which the soft-decision unit 74 and the hard-decision unit 75 perform soft-decision and hard-decision instead of the cumulative adder 71 as shown in FIG. the u _1} s and {u _2} s and the hard decision data sequence {u _1} h and {u _2} larger value by a comparator 72 correlation values C ₁ and C ₂ calculated in the correlator 76 h You may judge as a correct series. Here, the soft decision data sequence is a data sequence in which the amplitude of the temporary demodulation data sequence is quantized with a plurality of bits, and the hard decision data sequence is a data sequence in the case of 1-bit quantization.

Further, in the above embodiment, the interpolation timing generating means 5 is a carrier power to noise power ratio measuring means 9 as shown in FIG.
Is provided, the carrier power (C) is an average value of the correction data signal from the carrier wave estimating means 4, and the noise power (N) is a dispersion value.
Respectively, the carrier power to noise power ratio (C / N) information 20 is measured, the Nyquist point is estimated with respect to the cumulative addition value P (τ) corresponding to the C / N information 20, and the interpolation timing information 14 and 15 are obtained. May be generated. For example, in BPSK, the shape of P (τ) changes according to the value of C / N as shown in FIG. 7. Therefore, it is better to estimate the Nyquist point from P (τ) corresponding to C / N with respect to the noise superimposed signal. This has the effect of reducing the phase estimation error.

In the above embodiment, the interpolation timing generating means 5 is described as performing the calculation on the correction data signal 13 from the carrier wave estimating means 4, but as shown in FIG. 8, the sampling point data signal 12 from the buffer memory 3 is converted. You may perform calculation. Since the calculation uses sample point data on the complex plane, the cumulative addition value P (τ) is the same even if there is a residual frequency due to quasi-coherent detection.

In the above embodiment, the waveform of the output of the analog / digital converter 2 is shaped.
It goes without saying that analog / digital conversion may be performed after waveform shaping as long as it is a reception filter considering folding or the like.

[0024]

In the burst demodulator of the present invention as described above, low-speed samples (for example, 2 per symbol period) are used.
Since the collective batch demodulation method for extracting the optimum phase point by (sample) is adopted, the load can be reduced compared to the conventional method using high-speed samples, and high-speed transmission can be coped with. Further, the introduction of the interpolation timing generation means has the effect of reducing the phase estimation error by reducing the number of samples.

[Brief description of drawings]

FIG. 1 is a functional block diagram of a burst demodulator according to an embodiment of the present invention.

FIG. 2 is a diagram for explaining the operation of the interpolation timing generation means shown in FIG.

FIG. 3 is a diagram for explaining the operation of the data correction means shown in FIG.

FIG. 4 is a functional block diagram of the timing determining means shown in FIG.

5 is a functional block diagram of another embodiment of the timing determining means shown in FIG.

FIG. 6 is a functional block diagram of another embodiment including carrier power to noise power ratio measuring means according to the present invention.

FIG. 7 is a diagram for explaining the operation of the carrier power to noise power ratio measuring means shown in FIG.

FIG. 8 is a functional block diagram of another embodiment showing the present invention.

FIG. 9 is a functional block diagram of a conventional burst demodulator.

10 is a functional block diagram of the digital signal processing demodulator shown in FIG.

FIG. 11 is a diagram for explaining one sample operation per symbol of the timing estimation means shown in FIG.

FIG. 12 is a diagram for explaining an X sample operation per symbol of the timing estimating means shown in FIG.

[Explanation of symbols]

 1 Quasi-Synchronous Detector 2 Analog / Digital Converter 3 Buffer Memory 4 Carrier Estimating Means 5 Interpolation Timing Generating Means 6 Data Correcting Means 7 Timing Determining Means 8 Data Judging Means 9 Carrier Power to Noise Power Ratio Measuring Means 11 Received Signals 12 Sample Points Data signal 13 Corrected data signal 14, 15 Interpolation timing information 16, 17 Temporary demodulated data sequence 18 Demodulated data sequence 19 Demodulated data signal

Claims (4)

[Claims] 1. A quasi-synchronous detector and analog for temporarily storing all sample point data signals for each burst by subjecting a burst mode modulated wave obtained by quasi-coherent detection from a digital phase modulation burst reception signal to sampling and quantization operations. A digital signal processing type demodulator for extracting a optimum phase point of the sample point data signal from the buffer memory and judging and outputting a demodulated data signal from the buffer memory; The processing type demodulator estimates carrier wave from the data sequence of the sample point data signal from the buffer memory and corrects the frequency and phase of the data, and calculates the corrected data signal from the carrier wave estimating means. Interpolation timing for applying multiple Nyquist points and generating multiple interpolation timing information Generating means and data correcting means for performing interpolation calculation on the correction data signal from the carrier wave estimating means with the plurality of interpolation timing information from the interpolation timing generating means to estimate Nyquist point data and generating as a plurality of temporary demodulation data series. Timing determination means for comparing and judging the cumulative addition value of the absolute value or absolute square value of the data amplitude of each of the plurality of temporary demodulated data series from the data correction means, and selecting the demodulated data series; A burst demodulation device, comprising: data determining means for determining the demodulated data sequence from the determining means and outputting it as a demodulated data signal. 2. An absolute variance value, a correlation value between a known pattern portion and a predetermined known pattern, or a soft decision and a hard decision for each of a plurality of temporary demodulation data sequences from the data correction means by the timing decision means. The demodulation data sequence is selected by comparing and determining correlation values between the soft decision data sequence and the hard decision data sequence subjected to.
The burst demodulator described. 3. A carrier power to noise power ratio measuring means for calculating an average value and a variance value of the correction data signal from the carrier wave estimating means in the interpolation timing generating means and measuring a carrier power to noise power ratio is provided. The burst demodulator according to claim 1 or 2, wherein an operation is performed on an input signal corresponding to the noise power ratio information. 4. The burst demodulator according to claim 1, 2 or 3, wherein the interpolation timing generation means calculates the sampling point data signal from the buffer memory instead of the correction data signal from the carrier wave estimating means. .