JPH06178136A - Resonance type power source circuit - Google Patents

Abstract

PURPOSE:To obtain a resonance type power source circuit which prevents an unnecessary voltage pulse to generate for the period from the termination of a damper period to the next switch-on by a simple circuit. CONSTITUTION:A driving power source 3 is connected with one end side of the primary coil 2 of a flyback transformer 1, a damper diode 5 and MOS FET 11 are connected with the other end side of the primary coil 2 and the series circuit of a resonance capacitor and a clamp diode 12 is connected with the primary coil 2 in parallel. The clamp diode 12 is connected with the transistor 14 in parallel. A drive circuit 22 performs the switch driving of a MOS FET 11 by the drive signal where the pulse width of ON is widened as the drop amount of high pressure output voltage grows larger. A transistor 14 is turned on at the time of turning off the MOS FET 11 and is turned off within a damper period.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance type power supply circuit in which a voltage pulse is generated by a resonance operation of a primary side of a transformer, the voltage pulse is boosted by the transformer and output from a secondary side of the transformer. .

[0002]

2. Description of the Related Art A resonance type power supply circuit used in a television receiver or a display device incorporates a stabilizing circuit for stabilizing a high voltage output voltage applied from a flyback transformer to a cathode ray tube.

This type of stabilizing circuit is of a type that controls the power supply voltage of the drive power source on the primary side of the flyback transformer, and the correction voltage on the secondary side of the flyback transformer for the amount of drop in the high voltage output voltage. There is a secondary side control system that adds

The method for controlling the power supply voltage on the primary side is as follows:
There is a problem that the control response is extremely poor, and the secondary side control system has a problem that the correction width is narrow and the circuit becomes complicated.

Therefore, recently, in order to solve such a problem as much as possible, a method of directly controlling the primary side current of the flyback transformer is disclosed in Japanese Patent Laid-Open No. 2-2.
No. 22374 is proposed. However, this proposed example has an advantage that the control range of the high-voltage output voltage can be widened, but has a large power loss because an unnecessary return current flows, and noise is generated when the switching element is switched during the scanning period. There is a problem of causing.

[0006]

The inventors of the present invention have focused on a basic circuit of a resonance type power supply circuit as shown in FIG. 8 as a circuit of a system for directly controlling the primary side current, and have been working on improvement of its circuit characteristics. I'm out.

In the figure, the positive side of the drive power source 3 is connected to one end side of the primary coil 2 of the flyback transformer 1, and the negative side of the drive power source 3 is connected to the ground.
A transistor 4 as a switching element is connected in series to the other end of the primary coil 2, and a damper diode 5 and a resonance capacitor 6 are connected in parallel to the transistor 4.

The high voltage end of the secondary coil 7 of the flyback transformer 1 is connected to the anode of the cathode ray tube 10 via a high voltage rectifying diode 8.

In this type of circuit, during the ON period of the transistor 4 (transistor period), the current shown in FIG. 9B flows from the drive power supply 3 to the transistor 4 through the primary coil 2 and the primary coil 2 Electromagnetic energy is applied to. Next, when the transistor 4 is turned off, the primary coil 2
And the resonance capacitor 6 start series resonance, the electromagnetic energy stored on the primary coil 2 side is converted into electrostatic energy of the resonance capacitor 6, and as shown in FIG. Pulse) is generated. The flyback pulse has a peak when all the electromagnetic energy of the primary coil 2 is converted into the electrostatic energy of the resonance capacitor 6.

When the flyback pulse reaches its peak,
This time, the electrostatic energy of the resonance capacitor 6 is converted back into the electromagnetic energy of the primary coil 2, and as a result, the voltage of the flyback pulse gradually decreases. Then, when the pulse voltage is zero, that is, when the voltage at the point A of the circuit of FIG. 8 becomes zero, the damper diode 5 is turned on and a reverse current flows from the ground side to the primary coil 2 side, and the voltage at the point A becomes Drive power supply 3
When the power supply voltage is restored, the damper diode 5
Turns off. Then, when the transistor 4 is turned on again, the first circuit operation state is set, and by repeating this operation, the circuit operation is continued. The flyback pulse generated on the primary coil 2 side is boosted by the flyback transformer 1 and applied to the anode of the cathode ray tube 10 via the high voltage rectifying diode 8.

By the way, the pulse voltage V _c of the flyback pulse generated on the primary side of the flyback transformer 1 is
It is represented by the formula of V _c = E _B + r _a sin (ωt−φ _a ).
However, t is time, E _B is the supply voltage of the drive power source 3, and

R _a = √ {E _B ² + (I ₀ / Cω) ² }

Ω = (LC) ^-1/2

Φ _a = tan ⁻¹ (E _B Cω / I ₀ ).

Here, I ₀ is the current flowing through the primary coil 2, C is the capacitance of the resonance capacitor 6, and L is the inductance of the primary coil 2.

As is clear from the above equation of V _c , the voltage peak of the flyback pulse is when ωt−φ _a = π / 2, and at this time, the peak voltage is V _c = E _B + √ {E _B
A _{^{2 + (I 0 / Cω)}} 2}.

The current I ₀ flowing in the primary coil is I ₀
= (E _B / L) t _on + I ₁ . Here, t _on is the time when the transistor 4 is on, and I ₁ is zero, so that I ₀ is proportional to the on period of the transistor 4. Therefore, the flyback pulse crest value can be controlled by controlling the ON period of the transistor 4, and thus the high voltage output voltage output from the secondary coil 7 can be controlled.

In this way, when the ON period of the transistor 4 is controlled to stabilize the high voltage output voltage, the damper circuit (the ON period of the damper diode 5) is used from the ground side in the circuit of FIG. 8 as it is. When a reverse current flows through the diode 5 to the primary coil 2 side and the voltage at the point A is restored to the power supply voltage due to the flow of the reverse current and the damper diode 5 is turned off, the transistor 4 is already turned off. A current passing from the power supply 3 side through the primary coil 2 starts to flow through the resonance capacitor 6 to the ground side,
Series resonance occurs between the primary coil 2 and the resonance capacitor 6, and as shown in (a) of FIG. 9, an unnecessary pulse P is generated in the interval from the end of the damper period to the next turning on of the transistor 4. _W occurs.

This P _W causes noise and adversely affects the circuit operation. Therefore, in this type of circuit, in order to prevent current from flowing from the primary coil 2 side to the resonance capacitor 6 at the end of the damper period, the drive power supply 3 passes through the primary coil 2 and the resonance capacitor 6 or the transistor 4 to the ground. A switch element for cutting off the current is provided in the path leading to, or the start side of the ON period of the transistor 4 is overlapped with the end side of the damper period as indicated by the broken line in FIG. 9C. A new circuit design is required so that the current from the drive power supply 3 side flows to the transistor 4 side.

However, since the ON period of the transistor 4 overlaps the damper period, the output of the transistor 4 cannot be controlled by the ON timing. Further, even in the case where the switch element for cutting off the current is provided in the path from the drive power source 3 to the ground via the primary coil 2 and the resonance capacitor 6 or the transistor 4, a complicated control circuit for performing the operation timing of the switch element. Therefore, there is a problem that the number of components is increased, the entire circuit configuration is complicated, and the circuit cost is also increased.

The present invention has been made to solve the above problems, and an object thereof is to prevent the generation of an unnecessary pulse from the end of the damper period to the ON period of the transistor with a simple circuit configuration. Moreover, it is another object of the present invention to provide a resonance type power supply circuit which does not cause power loss due to the flow of an unnecessary return current as in the conventional primary side current control method.

[0022]

In order to achieve the above object, the present invention is constructed as follows. That is, according to the present invention, a driving power source is connected in series to a primary coil of a transformer, and a first switch element for controlling on / off of a current flowing through the primary coil on the primary coil side, and the first switch. A resonance capacitor that generates a voltage pulse by series resonance with the primary coil when the element is off, and a damper diode that causes a current flowing in a direction opposite to the direction of the current flowing through the first switch element to flow through the primary coil within a damper period are connected. In this resonance type power supply circuit, a clamp diode in which the direction of the current flowing from the primary coil to the resonance capacitor is in the forward direction is connected in series to the resonance capacitor, and the series circuit of the resonance capacitor and the clamp diode is a clamp diode. It is connected in parallel to the primary coil with the cathode side of the It is configured as characterized in that the second switching element is turned off in the damper period and turned on after the time of off of the first switch element is connected in parallel to the diode.

[0023]

In the present invention having the above structure, when the first switch element is turned on, a current flows from the drive power source side through the primary coil and the first switch element, and electromagnetic energy is stored in the primary coil. When the first switch element is turned off, a voltage pulse is generated due to series resonance of the resonance capacitor and the primary coil. After this voltage pulse is generated, the damper period starts, a reverse current flows from the damper diode side to the primary coil side, and after the damper diode turns off at the end point of the damper period, the voltage across the resonant capacitor is The clamp diode and the second switch element clamp the power supply voltage of the drive power supply. Due to this clamping action, the drive power supply and both ends of the resonance capacitor are at the same potential, so that no current flows from the drive power supply side to the resonance capacitor side through the primary coil, and the next period from the end of the damper period By the time the switch element No. 1 is turned on, circuit loss due to unnecessary return current is eliminated, and unnecessary voltage pulse causing noise is not generated.

[0024]

Embodiments of the present invention will be described below with reference to the drawings. In the following description of the embodiments, the same circuit parts as those in the conventional example are designated by the same reference numerals, and the duplicated description will be omitted. FIG. 1 shows the circuit configuration of a first embodiment of a resonant power supply circuit according to the present invention. In the figure, the drive power source 3 is connected to one end side (for example, winding start end side) of the primary coil 2 of the flyback transformer 1, and the first switch is connected to the other end side (winding end end side) of the primary coil 2. The drain side of a MOS FET (field effect transistor) 11 as an element is connected, and the source side of the MOS FET 11 is connected to the ground. A damper diode 5 is connected in parallel to the MOS FET 11 in the direction opposite to the direction of the current of the MOS FET 11. The damper diode 5 may be connected to an external diode as an electronic component. However, when the MOS FET 11 is used as a switch element, the MOS FET 11 itself has a reverse diode characteristic, so that the external diode is not used. By omitting the diode component, the diode characteristic of the MOS FET 11 can be made to function as the damper diode 5.

Further, one end side of the resonance capacitor 6 is connected to the winding end end side of the primary coil 2, the anode side of the clamp diode 12 is connected to the other end side of the resonance capacitor 6, and the cathode side of the clamp diode 12 is the primary side. It is connected to the connection between the coil 2 and the driving power supply 3. A transistor 14 as a second switch element is connected in parallel to the clamp diode 12.

One end of a series circuit of voltage dividing resistors 15 and 16 is connected to the high voltage end side of the secondary coil 7 of the flyback transformer 1, and the voltage dividing resistors 15 and 16 are resistance-divided.
High voltage output voltage is detected. Then, this detection voltage is applied to the non-inverting input terminal of the operational amplifier 17. A reference voltage is applied from the reference power supply 18 to the inverting input terminal side of the operational amplifier 17, and the operational amplifier 17 compares the detected voltage of the high voltage output voltage with the reference voltage and compares the signal corresponding to the drop amount of the high voltage output voltage with the comparator. Add to inverting input terminal of 20.
On the other hand, the signal from the waveform shaping circuit 21 is applied to the non-inverting input terminal of the comparator 20.

The waveform shaping circuit 21 integrates a horizontal drive signal (HD signal) shown in FIG. 2 (a) synchronized with a horizontal deflection output circuit (not shown) to generate a ramp as shown in FIG. 2 (b). A waveform is created and this ramp waveform signal is applied to the non-inverting input terminal of the comparator 20. The comparator 20 compares the ramp waveform signal with the signal from the operational amplifier 17, and as shown in FIGS. 2B and 2C, rises at the intersection of the output of the operational amplifier and the ramp waveform, and falls of the ramp waveform. That is, a drive signal that falls at the fall of the HD signal is produced. When the drop amount of the high voltage output voltage increases, the output level of the operational amplifier also decreases, and as a result, the pulse width of the drive signal increases. The comparator 20 produces a drive signal having a wider pulse width as the amount of drop of the high voltage output voltage increases, and applies the drive signal to the drive circuit 22. The drive circuit 22 performs switch-on driving of the MOS FET 11 according to the ON pulse width of the drive signal. As shown in FIG. 2F, the transistor 14 is supplied with a control signal which rises when the drive signal is off and turns off within the damper period.

This embodiment is constructed as described above, and its operation will now be described with reference to the circuit of FIG. 1 and the time chart of FIG. First, at t ₀ , MOS FE
When T11 is turned on, a current flows from the drive power source 3 side to the primary coil 2 and further to the ground side through the MOS FET 11. The current flowing through the primary coil 2 increases with time as shown in FIG. 2 (e), and electromagnetic current is stored in the primary coil 2 due to the current flow.

Next, when the MOS FET 11 is turned off at t ₁ , a current flows from the primary coil 2 toward the drive power source 3 side along the route passing through the resonance capacitor 6 and the clamp diode 12, and the inductance of the primary coil 2 and the resonance capacitor 6
The LC series resonance with the capacitance of is started and a flyback pulse (voltage pulse) is generated. This flyback pulse becomes maximum when all the electromagnetic energy on the primary coil 2 side is converted into the electrostatic energy of the resonance capacitor 6. Since the transistor 14 has been turned on since the MOS FET 11 was turned off, after all the electromagnetic energy of the primary coil 2 has been transferred to the resonance capacitor 6, this time from the drive power source 3 side, the transistor 14, resonance capacitor 6 and primary coil 2 are turned on. A reverse current flows through a route that sequentially passes through, and the electrostatic energy of the resonance capacitor 6 is inversely converted into the electromagnetic energy of the primary coil 2.

At t ₂ when the flyback pulse is completed, the voltage at the point A in the circuit of FIG. 1 becomes zero. At this time, the damper diode 5 turns on, and the primary side passes from the ground side through the damper diode 5 A current flows on the coil 2 side. This reverse current flow causes the voltage at point A to rise and t
_When the potential becomes equal to the power source voltage E _B of the driving power source 3 at 3, the damper diode 5 is turned off. At this time, MOS F
Since ET11 is off, current tends to flow from the drive power supply 3 side to the resonance capacitor 6 side. However, in this embodiment, the clamp diode 12 is provided, so that the voltage across the resonance capacitor 6 (point A , And the voltage at the point B) are both clamped to the power supply voltage E _B of the driving power supply 3 and held at the same potential as E _B , so that no current flows from the primary coil 2 side to the resonance capacitor 6 side. The unnecessary pulse voltage P _W that causes noise as shown in FIG. 9A is not generated.

Next, at time t ₄ , when the MOS FET 11 is turned on, the point A is grounded, and the drive power source 3
The current flowing from the primary coil 2 to the ground side through the MOS FET 11 coincides with the initial state of t ₀ . The circuit operation is continued by repeating the operations from t ₀ to t ₄ .

In the present embodiment, as the high-voltage output voltage drops, the on-time of the MOS FET 11 becomes longer, which increases the electromagnetic energy stored in the primary coil 2 and also increases the peak value of the flyback pulse generated. Therefore, the high voltage output voltage is effectively stabilized. Moreover, since the switch element for controlling the high-voltage output voltage is composed of one MOS FET element, the number of parts is very small and the circuit configuration is simple.

From the end of the damper period, the MOS
During the period when the FET 11 is turned on, the voltage across the resonance capacitor 6 is clamped to the same potential as the power supply voltage of the drive power supply 3, so that a current may flow from the drive power supply 3 to the resonance capacitor 6 through the primary coil 2. Therefore, it is possible to prevent the generation of unnecessary pulse voltage P _W that causes noise during this period. Further, since no return current is generated, there is no power loss (circuit loss) due to this return current, and the efficiency of circuit drive can be improved.

Moreover, the circuit for preventing the voltage pulse P _W has an extremely simple circuit configuration using only one clamp diode 12 and one transistor 14, and is a switch for blocking the flow of current as in the conventional case. It is not necessary to provide a complicated circuit to control the element and its switch element,
As a result, the circuit configuration can be simplified and the circuit cost can be significantly reduced.

Further, in this embodiment, the resonance capacitor 6
Since the voltage across both ends is clamped to the drive power supply voltage, the unnecessary voltage pulse P _W is prevented from occurring.
As shown by the broken line in FIG. 9C, the MOS FE
There is no restriction that the ON period of T11 overlaps with the damper period, whereby the pulse width of the drive signal for switching the MOS FET11 can be expanded to the pulse width of the horizontal drive signal to the maximum, and it is extremely wide. Therefore, voltage control becomes possible.

Further, in the circuit of this embodiment, the deflection cycle is 1
Since it is a resonance type circuit that charges and discharges every cycle, the response of stabilization of the high voltage output voltage is extremely good, and the control performance of the high voltage stabilization can be significantly improved.

Further, in this embodiment, since the transistor 14 that turns on the switch from the time when the MOS FET 11 is off to the damper period is connected in parallel to the clamp diode 12, the waveform of the flyback pulse generated is ideally symmetrical. Waveform can be set. That is, when the transistor 14 is not provided, when the flyback pulse is generated, when the MOS FET 11 is turned off and the electromagnetic energy of the primary coil 2 is converted into the electrostatic energy of the resonance capacitor 6, the current is the drive power source 3 side. Flows from,
Next, the electrostatic energy of the resonance capacitor 6 is transferred to the primary coil 2
The current does not flow from the drive power source 3 side because the clamp diode 12 is in the opposite direction when the electromagnetic wave energy is converted back into the electromagnetic energy. Therefore, the power supply voltage E _B content of the step of driving the power supply 3 to the peak value of the left half of the waveform and the right half is produced, which is a factor to deteriorate the high-pressure regulation characteristic of the flyback transformer 1.

On the other hand, in this embodiment, since the transistor 14 is turned on during the generation of the flyback pulse, even when the electrostatic energy of the resonance capacitor 6 is inversely converted into the electromagnetic energy of the primary coil 2, the current is converted. Is flowing from the drive power source 3 side, the peak value of the left half and the peak value of the right half of the waveform are the same, and the ideal flyback pulse shown in (b) of FIG. Therefore, it is possible to prevent deterioration of the high voltage regulation characteristic of the flyback transformer 1.

FIG. 4 shows a second embodiment of the present invention. In this embodiment, a series circuit of a deflection yoke D _Y and an S-shaped correction capacitor C _S is connected in parallel with the resonance capacitor 6 to form an integrated type circuit configuration for high voltage generation and deflection drive, and also a comparator 20 and a drive circuit. A pulse width limiter 23 is provided between the circuit 22 and the circuit 22, and the other configuration is the same as that of the first embodiment.

Generally, in a multi-scan type circuit capable of driving deflection over a wide range from low frequency to high frequency, the upper limit voltage of the crest value of the flyback pulse is set on the side where the deflection frequency is high frequency. There is. In the circuit of this embodiment, the pulse width of the drive signal can be widened up to the width of the maximum horizontal drive signal. Therefore, when performing multi-scan drive, the pulse width of the drive signal reaches the HD signal at the low frequency drive. When spread, MOS
The ON period of the FET 11 is much longer than that in the high frequency drive, and the current flowing through the primary coil 2 is also large. As a result, the peak value of the generated flyback pulse is much larger than that in the high frequency drive. There is a problem that the peak value of the flyback pulse exceeds the upper limit voltage, that is, the designed upper limit voltage. In this embodiment, in order to avoid this, a pulse width limiter 23 is provided, and the pulse width of the drive signal is limited by limiting the pulse width of the drive signal so as not to exceed the upper limit voltage set on the basis of the high frequency drive even for the low frequency deflection drive. The multi-scan drive in a wide frequency range from high frequency to high frequency can be performed without any trouble.

As a circuit for this multi-scan drive,
As shown in FIG. 5, the resonance capacitor may be configured by a series circuit of 6a and 6b, and the switch 19 may switch the resonance capacitance between low frequency driving and high frequency driving.

The present invention is not limited to the above-mentioned embodiments, and various embodiments can be adopted. In the circuit operation of each of the above-mentioned embodiments, while the next MOS FET 11 is turned on from the end of the damper period, the current which is about to flow from the drive power source 3 through the primary coil 2 is caused by the clamp operation of the clamp diode 12. Since there is no escape area for the flow of the current, there is a point t ₃ to t _{4 at the} point A, etc. of the circuit due to the parasitic capacitance built in the MOS FET 11 as shown in FIG.
Noise of the vibration component is generated in the section. This noise does not particularly cause damage when driving the cathode ray tube, but in order to achieve more perfect circuit performance, for example, as shown in FIG. 7A, the base side of the MOS FET 11, etc. Connect the saturable core 25 at an appropriate position, or connect the diode 26 in series to the source side of the MOS FET 11 as shown in FIG.
The built-in capacitance of the FET 11 and the built-in capacitance of the diode 26 are connected in series so as to reduce the apparent parasitic capacitance of the MOS FET 11, and further, as shown in FIG. May be provided to remove the noise of the vibration component between t _{3 and} t ₄ .

Further, in the circuits of the above respective embodiments, the smoothing capacitor 9 is provided on the high voltage end side of the secondary coil 7 as shown by the chain line, and the speed-up capacitor 28 for enhancing the response of high voltage stabilization is provided. It may be a thing. Also,
The circuit of each embodiment has a wide control range for high voltage stabilization, and
Since the response is good, it is generally not necessary to connect the choke coil for improving the regulation in parallel with the primary coil of the flyback transformer 1, but of course, this choke coil is connected in parallel with the primary coil 2. You may.

Furthermore, in each of the above embodiments, the series circuit of the voltage dividing resistors 15 and 16 is connected to one end of the secondary coil 7 in order to take out the high voltage output voltage. Since a resistance circuit for extracting the focus voltage and the screen voltage is connected to the secondary coil side, the high voltage output voltage may be detected using this resistance circuit.

Further, in each of the above embodiments, the MOS FE is
Although the high voltage output voltage is stabilized by the switch control of T11, the high voltage is stabilized by controlling the power source voltage of the drive power source 3 according to the drop amount of the high voltage output voltage, as in the conventional general circuit. May be performed.

Further, in the above embodiment, the first switch element is composed of the MOS FET 11, but
The switch element can be configured by using another switch element such as a bipolar transistor. Although the second switch element is composed of the transistor 24, it can be composed of another switch element such as a MOS FET.

Furthermore, in each of the above-described embodiments, the resonance type power supply circuit for generating a high voltage has been described, but the circuit of the present invention is also applicable as a resonance type power supply circuit for a low voltage.

[0048]

When the circuit of the present invention is used as a power supply circuit for generating high voltage, the first to the next from the end of the damper period.
The unnecessary voltage pulse that causes noise that is likely to occur until the switching element of the above is turned on can be prevented only by providing the second switching element in parallel with the clamp diode, and the second switching element is Since the control can be performed so that the first switch element is turned on when the first switch element is turned off and turned off within the damper period, the control circuit can be simply configured, and a complicated control circuit as in the past is not required.
The number of parts is significantly reduced, and the circuit cost can be significantly reduced.

Moreover, since the clamp action of the clamp diode and the second switch element prevents the generation of the return current of the circuit during the period when the first switch element is turned on next from the end of the damper period, this There is no circuit loss due to the return current, and the efficiency of circuit driving can be improved.

Further, since the circuit of the present invention is of the resonance type in which the resonance capacitor is charged and discharged in each cycle of deflection, the high-voltage stabilization response is excellent.

Furthermore, the circuit of the present invention has a configuration in which the clamp diode is used to clamp the voltage across the resonance capacitor to the power supply voltage of the driving power supply to prevent the generation of the unnecessary voltage pulse. Since there is no restriction that the ON period of the switch element must overlap with the damper period, the ON pulse width of the first switch element can be expanded to the pulse width of the horizontal drive signal, and the voltage control width can be significantly improved compared to the conventional one. It is possible to widen the range, and it is also suitable as a high voltage power supply for multi-scan.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a first embodiment of a resonant power supply circuit according to the present invention.

FIG. 2 is a time chart showing the circuit operation of the embodiment.

FIG. 3 is an explanatory diagram of an effect by the operation of the transistor 12 in the circuit of the embodiment.

FIG. 4 is a circuit diagram showing a second embodiment of the present invention.

FIG. 5 is an explanatory diagram of a circuit provided with a resonance capacitance switching unit as a modified example of the embodiment.

FIG. 6 is an explanatory diagram of vibration component noise generated from the end of the damper period to the next switch-on period.

FIG. 7 is an explanatory diagram of various circuit examples for removing vibration component noise.

FIG. 8 is a basic circuit diagram of a conventional resonant power supply circuit.

9 is a time chart showing the operation of the circuit of FIG.

[Explanation of symbols]

 1 Flyback Transformer 2 Primary Coil 3 Driving Power Supply 5 Damper Diode 6, 6a, 6b Resonant Capacitor 11 MOS FET (First Switch Element) 12 Clamp Diode 14 Transistor (Second Switch Element)

Claims (1)

[Claims] 1. A drive power source is connected in series to a primary coil of a transformer, and a first switch element for controlling on / off of a current flowing through the primary coil is provided on the primary coil side.
A resonance capacitor that generates a voltage pulse by series resonance with the primary coil when the first switch element is off;
In a resonance type power supply circuit connected to a damper diode that causes a current flowing in a direction opposite to that of a current flowing through the first switch element to flow in a primary coil within a damper period, the resonant capacitor flows from the primary coil to the resonant capacitor. The clamp diode with the direction of the current in the forward direction is connected in series, and the series circuit of this resonant capacitor and the clamp diode is connected in parallel to the primary coil with the cathode side of the clamp diode as the drive power supply side.
A resonance type power supply circuit characterized in that a second switch element which is turned on after the first switch element is turned off and turned off within a damper period is connected in parallel to the clamp diode.