JPH0584819U - Optical displacement detector - Google Patents

Abstract

(57) [Summary] [Structure] A light emitting current control means for controlling the light emitting current of the light emitting element 118 is provided so that the amplitude of the sine wave voltage output from the light receiving element 120 is always constant. Is a differential amplifier circuit 170 for obtaining an A ′ phase that is π-phase-shifted from the A-phase and a B ′ phase that is π-phase-shifted from the B-phase, and A-phase and A ′.
Phase, B-phase, and B′-phase differential phase amplitude amplified outputs are compared by half-wave rectification with a reference voltage to obtain a difference current, and the emission current of the light emitting element 118 is inversely proportional to the difference voltage. And a light emission current control circuit 200 for performing the optical displacement detection device. According to the optical displacement detection device of the present invention, the light emission current supply amount to the light receiving element 120 is controlled so that the amplitude of the voltage waveform from the light receiving element 120 is constant, and thus the encoder It is possible to reduce the size and cost, and the circuit configuration is simple, and the displacement amount detection accuracy can be improved.

Description

[Detailed description of the device]

[0001]

[Industrial applications]

 The present invention relates to an optical displacement detection device, and more particularly, to improvement of a control mechanism of a current flowing to a light source thereof.

[0002]

[Prior Art]

 Various measuring devices, machine tools, and more recently various information machines use various displacement detectors to detect the amount of displacement of two members that move relative to each other. Optical displacement detection devices are widely used. The optical displacement detection device includes a plate provided on each of the two members that move relative to each other, and a light emitting element and a light receiving element for detecting the overlapping of the gratings. As such a conventional optical displacement detection device, in addition to the usual encoder for detecting the overlap of two gratings, the displacement amount is detected by the change in the overlapping of three gratings as shown in FIG. The so-called three-lattice system is well known (Journal of the optical society of America, 1965, vol.55, No.4, PP373-381).

In FIG. 13, the encoder 10 includes a light emitting side grating 12 and a detection grating 14 arranged in parallel, a reference grating 16 arranged in parallel between the two gratings 12 and 14 so as to be relatively movable, and the light emitting. It includes a light emitting element 18 arranged on the left side of the side grating 12 in the figure, and a light receiving element 20 arranged on the right side of the detection grating 14 in the figure. Then, the light emitted from the light emitting element 18 reaches the light receiving element 20 via the light emitting side grating 12, the reference grating 16, and the detection grating 14, and the light receiving element 20 is irradiated by the respective gratings 12, 14, 16 limited. The light is photoelectrically converted and further amplified by the preamplifier 22 to obtain a detection signal s.

Here, when the reference grating 16 moves relative to the light-emitting side grating 12 and the detection grating 14 in the direction of arrow x, for example, the amount of light radiated from the light-emitting element 18 which is blocked by the gratings 12, 16, and 14 is increased. It gradually changes, and the detection signal s is output as a substantially sine wave. The pitch P1 of the reference grating 16 and the wavelength of the detection signal s correspond to each other, and the relative movement amount of the reference grating 16 is measured from the wave number of the detection signal s and its division value. FIG. 14 shows a vertical sectional view of such a reflective linear encoder of the three-grating system, and FIG. 15 shows a sectional view taken along the line I-I of FIG. The parts corresponding to those in FIG. 13 are designated by the same reference numerals and the description thereof is omitted. The linear encoder 10 shown in the figure is of a reflective type and has an index scale 30 and a main scale 32.

On the upper surface of the index scale 30 in FIG. 14, a detection unit 34 including one light emitting element 18 and four light receiving elements 20a, 20b, 20c, 20d is arranged. The lead wires of the light emitting element 18 and each light receiving element 20 are fixed to the print substrate 24. Further, the index scale 30 is provided with a frame 36 surrounding the light emitting element 18 and each light receiving element 20. Further, a reference grid 16 is provided on the main scale 32 arranged to face the index scale 30, and the reference grid 16 is formed with vertical striped scales as shown in FIG.

On the other hand, as shown in FIG. 17, the index scale 30 includes a light emitting side grating 12 corresponding to the light emitting element 18 and detection gratings 14 a, 14 b, 14 c, 14 d corresponding to the light receiving element 20. Are provided, and vertical stripe-shaped scales are formed on each grid. The pitches of the vertical striped scales of the detection gratings 14a ... 14d are formed so that their phases are shifted by 90 degrees from each other, and from each of the light receiving elements 20a ... 20d, an a phase whose phase is shifted by π / 2, respectively. The b-phase, a'phase, and b'phase signals can be obtained. Furthermore, it is possible to obtain the A-phase output which is differentially amplified by the a-phase-a 'phase and the B-phase output which is also differentially-amplified by the b-phase-b' phase. Then, discrimination of the relative movement direction of the scale from the shift direction of the π / 2 phase of the A-phase output and the B-phase output and the like, and the division of the detection signal electrically are performed to detect the displacement amount with high resolution. By the way, a light-emitting diode or the like having a long life and good stability is generally used for the light-emitting element, and the light-emitting diode or the like deteriorates with lighting time, and the light emission efficiency gradually decreases. Further, the luminous efficiency also varies depending on the temperature change.

As a result, the detection signal s may be affected and the displacement detection accuracy may be reduced. Therefore, in the prior art, the sum of the currents output in response to the light received by the light receiving elements 20a ... 20d is proportional to the amount of light emitted from the light emitting element 18, and therefore, based on the output of the light receiving elements 20a. The sum of the currents is measured, and the light emission current of the light emitting element 18 is feedback-controlled so that the currents are constant, so that the light emission amount is kept constant (JP-A-61-137012, JP-A-2-1372). 98630).

[0008]

[Problems to be solved by the device]

 By the way, in addition to the deterioration with time of the light emitting element and the temperature change, a change in the moving speed of the main scale, or a change in the mounting position of the index scale or the detecting part with respect to the main scale also affects the detection signal s, which lowers the displacement detection accuracy. Let's do it. That is, for example, when the relative moving speed of the main scale and the index scale becomes high, the waveform becomes blunt due to the response characteristics of the light receiving element and the like, which affects the displacement detection accuracy.

However, when the change in the moving speed and the change in the mounting posture occur as described above, only the differential output (sine wave output) from the light receiving element, that is, the amplitude of the sine wave changes. Therefore, the sum of the output signals from the light receiving elements hardly changes. Therefore, the conventional encoder described above has a problem that it cannot cope with the change in the moving speed and the change in the mounting attitude, and the displacement detection accuracy may be deteriorated. The present invention has been made in view of the above-mentioned problems of the prior art, and its purpose is to obtain a differential signal from the light receiving element even when the moving speed of the main scale changes or the mounting posture of the detector with respect to the main scale changes. An object of the present invention is to provide an optical displacement detection device capable of constantly maintaining the output (sine wave output) constant.

[0010]

[Means for Solving the Problems]

 In order to achieve the above object, an optical displacement detection device according to the present invention comprises a main scale, an index scale, a light emitting element, a light receiving element, and an emission current control means. Here, a predetermined reference grid is formed on the main scale. A detection grating is formed on the index scale and is arranged in parallel so as to be movable relative to the main scale. The light emitting element irradiates the reference grating with light. The light receiving element receives the light limited by the reference grating and the detection grating and outputs the relative movement amount of the main scale and the index scale as a substantially sine wave. The light emission current control means controls the light emission current of the light emitting elements so that the amplitude of the sine wave voltage output from each of the light receiving elements is always constant.

It should be noted that the index scale includes four detection gratings each shifted by π / 2 phase and four light receiving elements installed for each detection grating. A differential amplitude amplification output with a π / 2 phase shift is obtained, and the light emission current control means obtains an A ′ phase with a π phase shift with the A phase and a B ′ phase with a π phase shift with the B phase. The circuit and the total output obtained by half-wave rectifying the differential amplitude amplified outputs of the A phase, A'phase, B phase, and B'phase are compared with a reference voltage to obtain a difference current, which is inversely proportional to the difference voltage. And a light emission current control circuit for increasing / decreasing the light emission current of the light emitting element.

Further, in the present invention, it is also preferable to include an output compensating means for compensating a variation of the output level of the substantially sine wave signal output from the light receiving element and outputting the substantially sine wave signal of a constant level. is there. In this case, the index scale includes four detection elements that are each shifted by π / 2 phase and four light receiving elements that are installed for each detection grating. The A-phase and B-phase differential amplitude amplified outputs having a π / 2 phase shift are obtained, and the output compensating means obtains an addition signal of the output signals of the four light-receiving elements; and an addition output of the addition circuit. A differential amplifier circuit that obtains a differential signal from the reference signal, and amplifies the A-phase or B-phase signal that changes according to the output of the differential amplifier circuit so that the amplification level becomes constant. And an output circuit for outputting the data to the outside.

[0013]

[Action]

 Since the optical displacement detection device according to the present invention has the above-described means, the measurement light emitted from the light emitting element is partially shielded by the reference grid of the main scale and further limited by the detection grid of the index scale. It reaches the light receiving element. The light quantity limited by the reference grating and the detection grating changes according to the relative position change of the main scale and the index scale, and each light receiving element outputs the relative movement amount of both scales as a substantially sine wave. Here, for example, the index scale is formed with two pairs of detection gratings, each of which has four phases shifted by π / 2 phases. Therefore, it is shifted by π / 2 phases from the light receiving element corresponding to each detection grating. A substantially sine wave is output.

Further, in the present invention, the light emitting current control means controls the light emitting current to the light emitting element so that the amplitude of the sine wave voltage output from each of the light receiving elements always matches the reference voltage. Even when only the amplitude of the sine wave voltage that is output changes, such as changes in the moving speed of the main scale and changes in the mounting posture of the detector, changes in the light intensity of the light emitting element can be compensated appropriately. It is possible to maintain the displacement detection with high accuracy. Further, it is possible to obtain substantially the same effect by compensating for the variation of the output level of the substantially sine wave signal output from the light receiving element and outputting the substantially sine wave signal of a constant level.

[0015]

【Example】

 Hereinafter, preferred embodiments of the present invention will be described with reference to the drawings. FIG. 1 shows a circuit configuration of a linear encoder as an optical displacement detection device according to an embodiment of the present invention. A portion corresponding to the above-mentioned prior art is denoted by reference numeral 100 and its description is omitted. .. As shown in the figure, the linear encoder according to the present embodiment employs a three-grating system, and the light-emitting side grating 112, the reference grating 116, and the light-receiving element 120a, 120b, ... , And the detection gratings 114a, 114b, ... 114d corresponding to the respective light receiving elements 120a, 120b ,.

Here, the detection gratings 114a and 114b are formed with gratings shifted from each other by π phase, and the detection gratings 114c and 114d are also formed with gratings shifted from each other by π phase, so that the detection gratings 114a and 114c are formed. Are configured by shifting the π / 2 phase. Therefore, the detection gratings 114a, 114c, 114b, 114d are out of phase with each other by π / 2, and the relative movement of the reference grating 116 causes the light receiving elements 120a, 120c, 120b, 120d by π / 2 each. A detection signal of a substantially sinusoidal wave with a phase shift can be obtained. Here, the light receiving element 120 is formed of a phototransistor in this embodiment, and a current is conducted from the power supply 100 in proportion to the amount of light applied to each base. Taking the light receiving element 120a as an example, the current I ₁ conducted from the power source 100 to the light receiving element 120a forms an a-phase signal by the light receiving element 120a and is converted into a voltage by the current-voltage conversion circuit 140a. ..

The current-voltage conversion circuit 140a is composed of an operational amplifier 142, a parallel circuit of a noise prevention capacitor 144 and a variable resistor 146 connected in parallel with the operational amplifier 142, and a resistor 148. Then, the light receiving element 120a is connected to the inverting input terminal of the operational amplifier 142, and the reference voltage Vref is applied to the non-inverting input terminal via the resistor 148. Similarly, the respective currents I ₂ , I ₃ , I ₄ conducted from the power source 100 to the light receiving elements 120b, 120c, 120d form an a'phase signal, a b phase signal, and a b'phase signal, respectively. Are converted into voltages by current-voltage conversion circuits 140b, 140c, 140d having the same configuration as the current-voltage conversion circuit 140a. The voltage waveforms at the points P ₁ , P ₂ , P ₃ , P ₄ output from the current-voltage conversion circuits 140a ... 140b by the relative movement of the reference grid 116 are respectively π / shown in FIG. 2 as described above. It becomes a substantially sine wave whose phase is shifted by two.

Next, the a-phase signal and the a′-phase signal are made into an a-phase-a′-phase signal by the differential amplifier circuit 150a, and the DC component (V _SDC shown in FIG. 2) is removed to obtain the differential amplitude. Output the amplified phase A. That is, the differential amplifier circuit 150a is composed of an operational amplifier 152 and resistors 154, 156, 158, 160, and the inverting input terminal of the operational amplifier 152 receives an a-phase signal converted into a voltage by the current-voltage conversion circuit 140a. It is input via the resistor 154. Further, the non-inverting input terminal of the operational amplifier 152 is input with the a′-phase signal converted into the voltage by the current-voltage conversion circuit 140b via the resistor 156. Further, the resistor 158 is connected in parallel with the operational amplifier 152, and the resistor 160 connects the non-inverting input terminal of the operational amplifier 152 and the reference voltage Vref. Then, the a-phase signal is differentially amplitude-amplified by the operational amplifier 152 with reference to the a'-phase signal whose π phase is shifted to form an A-phase output. In the same manner, the b-phase signal and the b′-phase signal are made into the b-phase−b′-phase signal by the differential amplifier circuit 150b, and the DC component is removed to form the B-phase output which is differentially amplified. To do.

FIG. 3 shows voltage waveforms at points P ₅ and P ₆ output as the A phase and the B phase from the differential amplifier circuits 150a and 150b as described above. As described above, the A-phase and B-phase outputs are out of phase by π / 2, and the A-phase and B-phase voltages are directly extracted by the comparator as a two-phase square wave, or the A-phase and B-phase are internally generated. It is used as a position signal by inserting it and improving the resolution, and it is possible to discriminate the reference grating 116 (main scale) in the relative movement direction. By the way, the light emitting element 118 including a light emitting diode or the like has a change in the light emitting efficiency due to a temperature change or a change with time, and a particularly large decrease in the light emitting efficiency can be observed when the light emitting diode is used at a high temperature for a long time. Then, due to the decrease in the light emitting efficiency, the amplitudes of the voltage waveforms of the A phase and the B phase decrease. Further, the light receiving element 120 including a phototransistor has a distribution capacitance by a PN junction, and the capacitance of the anti-noise capacitor 144 shown in FIG. 1 is usually required to be 100 pF or more in consideration of noise resistance. is there. Due to these capacitances, the amplitude (Vpp) of the A-phase and B-phase voltage waveforms decreases as the frequency increases, that is, when the moving speed of the reference grating 116 increases.

Further, in the top view of the linear encoder shown in FIG. 5, the detection unit 134 is attached to the main scale 132 at an angle Δθ, or the main scale 132 is inclined at an angle Δθ during movement. Also in this case, the amplitude of the A-phase and B-phase voltage waveforms (Vpp shown in FIG. 3) decreases. That is, for example, in the linear encoder having the pitch P ₁ = 20 μm of the reference grating 16 shown in FIG. 13, the amplitude is reduced to about 90% at the angle θ≈0.1 °. Therefore, unless the detector 136 is attached to the main scale 132 so that the angle θ ≦ 0.1, the amplitude is sensitive to the magnitude of the angle θ, and the amplitude is greatly reduced. .. Then, the decrease in the amplitudes of the voltage waveforms of the A phase and the B phase described above deteriorates the detection accuracy of the relative movement amount of the main scale 130. The present invention compensates for changes in the light emission characteristics of the light emitting element 118, changes in the moving speed of the reference grating 116 (main scale 132), and changes in the mounting attitude of the detector 136. A differential amplifier circuit 170 and a light emission current control circuit 200 are provided as control means.

The differential amplifier circuit 170 a includes an operational amplifier 172 and resistors 174, 176, 178 and 180, like the differential amplifier circuit 150. The operational amplifier 172 has an inverting input terminal connected to the differential amplifier circuit 170 a. The a'phase signal converted into a voltage by the current-voltage conversion circuit 140b is input via the resistor 174. Further, the non-inverting input terminal of the operational amplifier 172 is input with the a-phase signal converted into the voltage by the current-voltage conversion circuit 140a via the resistor 176. Further, the resistor 178 is connected in parallel with the operational amplifier 172, and the resistor 180 connects the non-inverting input terminal of the operational amplifier 172 and the reference voltage Vref. Then, the differential amplification circuit 170a removes the DC component and differentially amplifies the amplitude, and a'is shifted from the a phase-a 'phase signal (A phase output) from the differential amplification circuit 150a by π phase. A phase-a phase signal (A 'phase output) is created.

Similarly, based on the b phase and the b ′ phase output from the current-voltage conversion circuits 140c and 140d by the differential amplifier circuit 170b, the b phase−b ′ phase signal from the differential amplifier circuit 150b. A b'phase-b phase signal (B 'phase output) that is out of phase with (B phase output) is created. FIG. 6 shows A-phase output and B-phase output at points P ₅ and P ₆ output from the differential amplifier circuits 150a and 150b, and P ₇ and P output from the differential amplifier circuits 170a and 170b. The voltage waveforms of A'phase output and B'phase output at ₈ points are shown. Then, in the present embodiment, the light emission current control circuit 200 half-wave rectifies the outputs of the A phase, B phase, A ′ phase, and B ′ phase, and based on the total output, the amplitude value (Vpp The light emitting current of the light emitting element 118 is controlled so that the value of () is constant.

Here, the light emission current control circuit 200 includes detection diodes 210 a, 210 b, 210 c, 210 d, a voltage adjustment circuit 212, a differential amplification circuit 214, and a reference voltage comparison and amplification circuit 216. Contains. The detection diode 210a is connected to the output terminal of the differential amplifier circuit 150a and the reference power source 218 via a parallel circuit of a resistor 220 and a capacitor 222 and a resistor 224. Similarly, the detection diodes 210a, 210b, 210d also connect the output terminals of the differential amplifier circuit 150b and the differential amplifier circuits 170a, 170b to the power supply 218, respectively. FIG. 7 shows the voltage (V ₉ ) at point P ₉ output to the detection diodes 210a ... 210d, and the voltage becomes a rectified voltage. Then, assuming that the resistance values of the resistors 220 and 224 are: resistor 220 >> resistor 224, the voltage V ₉ shown in FIG.

## EQU00001 ## V ₉ .apprxeq.Vref- (Vpp / 2-D _VF ) (D _VF : forward voltage of the detection diode). Therefore, if the light emitting current to the light emitting element 118 is controlled so that the voltage V ₉ becomes a constant voltage, the voltage waveform (V pp) output from the light receiving elements 120a ... 120d can be made constant. Become. Here, the power source 218 is connected to the voltage adjusting circuit 212 via a parallel circuit of a resistor 220 and a capacitor 222, and a resistor 224.

The voltage adjustment circuit 212 includes an operational amplifier 230 and a diode 226 and a parallel circuit of the diode 226 and a diode 228 having an opposite polarity. Then, the diodes 226 and 228 serve to compensate the forward voltage (D _VF ) of the detection diodes 210a ... Further, in the operational amplifier 230, the output terminal of the operational amplifier 230 is connected to the inverting input terminal and the parallel circuit of the diodes 226 and 228 is connected to the non-inverting input terminal, so that the voltages at the points P ₁₀ and P ₁₁ are equal. To work. The output terminal of the operational amplifier 230 is connected to the differential amplifier circuit 214 and amplifies the voltage difference between the point P ₁₁ and a predetermined voltage. That is, the differential amplifier circuit 214 includes resistors 232, 234, 236, 238 and an operational amplifier 240, and a resistor 232 connects the output terminal of the op-amp 230 and the inverting input terminal of the op-amp 240, and The resistor 234 is connected in parallel with the operational amplifier 240. Further, the resistor 236 and the resistor 238 are connected in series as a voltage dividing resistor, the other end of the resistor 236 is connected to the reference voltage power supply (V _ref ), and the other end of the resistor 238 is grounded. The connection point of the resistor is connected to the non-inverting input terminal of the operational amplifier 240.

Then, the operational amplifier 240 amplifies the voltage difference between the voltage at the inverting input terminal P ₁₂ point and the predetermined voltage at the P ₁₃ point adjusted by the voltage dividing resistors 236 and 238. Therefore, the voltage difference between the point P _{12 and the} point P ₁₃ is amplified and output to the point P ₁₄ which is the output terminal of the operational amplifier 240. Then, the voltage at the output terminal P ₁₄ of the operational amplifier 240 is input to the reference voltage comparison / amplification circuit 216, compared with the reference voltage in the reference voltage comparison / amplification circuit 216, and the voltage at the P ₁₄ point and the reference voltage Amplify the difference. That is, the reference voltage comparison / amplification circuit 216 includes resistors 242, 244, 246, 248, 250, capacitors 252, 254, 256, and an operational amplifier 258. The resistor 242 is connected to the inverting input terminal of the operational amplifier 258, and the parallel circuit of the resistor 244 and the capacitors 252 and 254 is connected in parallel with the operational amplifier 258. Further, the resistor 246 connects the voltage dividing resistors 248, 250 and the capacitor 256 to the non-inverting input terminal of the operational amplifier 258, the other end of the resistor 250 is connected to the power supply 260, and the resistor 248 and the capacitor 256 are connected to each other. The end is grounded.

In the reference voltage comparison / amplification circuit 216, the voltage at the point P ₁₅ that is the connection point of the voltage dividing resistors 248 and 250 becomes the reference voltage, and the voltage at the point P ₁₄ is amplified by the operational amplifier 258, and the operational amplifier 258 It is output from the output terminal of 258. Further, the voltage at the output terminal P _{16 of the} operational amplifier 258 is applied to the base of the transistor 264 via the resistor 262. The light emitting element 118 is connected to the collector of the transistor 264 via the resistor 266, and the emitter is grounded via the resistor 268. Here, the connection point between the resistor 262 and the transistor 264 is grounded via the resistor 270, and assuming that the resistance values of the resistor 262 and the resistor 270 are resistor 270 >> resistor 262, h _{fe of} the transistor 264 is usually 100 or more. since the voltage of the P ₁₆ points, the voltage of the P ₁₇ point is a connection point of the resistors 262 and transistor 264 becomes the voltage of the voltage ≒ P ₁₇ points P ₁₆ points.

Therefore, the amplitudes (Vpp) of the voltages output from the differential amplifier circuits 150a and 150b and the differential amplifier circuits 170a and 170b at points P ₅ , P ₆ , P ₇ , and P ₈ are optimal. In this state, if the voltage at the point P ₉ is set to the predetermined voltage (P ₁₅ voltage) via the circuits 212 and 214, the voltage at the point P ₁₆ also becomes the reference voltage, and the base of the transistor 264 has the voltage. A reference voltage is applied. Further, if the amplitude (Vpp) of the voltage at the points P ₅ , P ₆ , P ₇ , and P ₈ increases and the voltage at the point P ₉ decreases, the voltage at the point P ₁₆ increases, and therefore the voltage of transistor 264 increases. The current conducted between the collector and emitter increases. Therefore, the amount of light emitted from the light emitting element 118 increases and the amount of light received from the light receiving element 120 also increases, so that the voltage amplitude (Vpp) at points P ₅ , P ₆ , P ₇ , and P ₈ increases. , P ₉ voltage is kept constant.

As described above, according to the linear encoder according to the present embodiment, the amplitudes of the four-phase output voltages shifted from each other by π / 2 phases from the four light-receiving elements can be made constant. The displacement detection accuracy can be maintained at a high level with respect to changes in the luminous efficiency of the light emitting element due to temperature changes and deterioration over time, changes in the moving speed of the main scale, and changes in the mounting orientation of the detector. That is, according to the experiment, the linear encoder according to the present embodiment allows the moving speed of the main scale to double and the error in the mounting posture of the detector to triple, compared with the conventional encoder, and it is possible to cope with it. Became.

In the present embodiment, the diode 210 in the light emission current control circuit 200 is installed so that it is conducted from the light emission current control circuit 200 side so that the negative voltage is constant. It is also possible to install the diode 310 so that the direction of the diode 310 is conductive from the output side of the light receiving element, as shown in FIG. 7, so that the positive voltage at the point P ₉ 'is constant as shown in FIG. Further, although the light emitting element 118 and the light emitting side grating 112 are separately configured in the present embodiment, the present invention is not limited to this, and the light emitting element 118 and the light emitting side grating 112 are integrated into an array light source. It is also preferable that It is also preferable that the detection grating 114 and the light receiving element 120 are also integrated to form an arrayed light receiving element.

FIG. 9 shows another embodiment of the optical displacement detection device according to the present invention, and this embodiment will be described below with reference to the same drawing. The components corresponding to those of the embodiment shown in FIG. 1 are designated by the same reference numerals, the description thereof will be omitted, and different points will be mainly described below. In contrast to the embodiment shown in FIG. 1 in which the light emitting current of the light emitting element 18 is controlled based on the so-called feedback loop control, this embodiment is based on the so-called open loop control. It differs from the embodiment shown in FIG. 1 in that the outputs of the A phase and the B phase are made constant.

That is, in the embodiment shown in FIG. 9, the adder 400 that adds the output signals of the current-voltage converters 140a to 140d, and the difference between the output voltage of the adder 400 and the reference voltage is amplified. The differential amplifier circuit 402 and the output adjusting circuit 404 are included. In the embodiment shown in FIG. 9, the voltage adjusting circuit 212, the differential amplifying circuit 214, the reference voltage comparing and amplifying circuit 216 and the transistor 264 in the embodiment shown in FIG. Although no output circuit is provided, the differential amplifier circuit 402 in this embodiment is functionally substantially equivalent to the reference voltage comparison amplifier circuit 216 in FIG.

The adder 400 includes an operational amplifier 406, resistors 407 to 410 connecting the inverting input of the operational amplifier 406 and the output points P1 to P4 of the current-voltage converters 140a to 140d, and the inverting input and output of the operational amplifier 406. And a resistor 411 for connecting to the first reference voltage V_{ref (1)}It has a resistor 412 as an input resistor input to the inverting input. Then, at the output of the adder 400, a voltage as a sum of the output voltages of the current-voltage converters 140a to 140d is obtained. Here, the sum of the output voltages of the current-voltage converters 140a to 140d becomes a substantially DC voltage because the AC component is substantially canceled. FIG. 10 shows the voltage levels P1 to P4 and the first reference voltage V._ref( ₁₎ A waveform diagram showing the relative relationship with is shown. In order for the output of the adder 400 to be a DC signal, at least two phases of the a phase and the a'phase or the b phase and the b'phase are sufficient, and the four phases are not necessarily required.

In the adder 400, assuming that the resistance values of the resistors 407 to 410 are all equal, the values are R _c , the four-phase DC voltage level is V _SDC, and the non-inverting input of the operational amplifier 406 is input via the resistor 412. The voltage V _{5 at} the output point P ₅ of the adder 400 can be expressed by the following equation 2 when the reference voltage applied to V _{ref (1)} is determined.

[Number 2] _{V 5 = V ref (1)} + 4R FB · V SDC / R C Note that, R _FB is the resistance value of the resistor 411. Next, the differential amplifier circuit 402 includes an operational amplifier 413, a resistor 414 as a so-called feedback resistor connected between the inverting input and the output of the operational amplifier 413, and a second reference voltage V _{ref (2 )} , A resistor 416 connected between the output of the adder 400 and the non-inverting input of the operational amplifier 413, and the first reference voltage V _{ref (1 ) For} applying a voltage ₎ and capacitors 418 and 419 connected in parallel with the above-mentioned resistor 414. Then, the differential voltage between the output voltage V ₅ of the adder 400 and the second reference voltage V _{ref (2)} is amplified by a predetermined amplification degree and output to the output of the differential amplifier 402. Is becoming

The output adjustment circuit 404 includes first and second field effect transistors (hereinafter, referred to as “first FET” and “second FET”, respectively) 420 and 421, and a gate of the second FET 421. P ₆ side Ano between the output point P ₆ of the differential amplifier circuit 170a - and de 422, this diode - - diodes connected as a de between the scan - cathode of de 422 - de and a The connected resistor 423, the third and fourth field effect transistors (hereinafter referred to as "third FET" and "fourth FET", respectively) 424, 425, the gate of the fourth FET 425, and differential amplification. Between the output point P ₇ of the circuit 170b, a diode 426 connected so that the P ₇ side becomes an anode, and between the cathode and the ground of this diode 426. The resistor 427 and the anode are described above. The output point P ₁₀ of the differential amplifier 402, cathode - a de 428, this diode - - De is diode connected to the first and third FET 420, 424 cathode of de 428 - De and A - connected between the scan And a resistor 429 that has been set. Here, the first reference voltage V _{ref (1)} is applied to the sources of the first and third FETs 420 and 424. The drains of the first and second FETs 420 and 421 and the drains of the third and fourth FETs 424 and 425 are connected to each other. The connection point P ₁₁ between the drains of the first and second FETs 420 and 421 is connected to the non-inverting input of the non-inverting amplifier circuit 430a via the resistor 431, and the connection point between the drains of the third and fourth FETs 424 and 425 is It is connected to the non-inverting input of the non-inverting widening circuit 430b through the resistor 432.

The output adjusting circuit 404 constitutes an output circuit together with the non-inverting amplifier circuits 430a and 430b. In the above structure, the gate voltage of the second FET 421 is always on the negative side by the forward voltage (about 0.6 V) of the diode 422 than the source voltage by the action of the diode 422 and the resistor 423. Will be biased to. This circuit operation is basically the same for the gate voltage of the fourth FET 425, and the description thereof will be omitted.

The gate voltage of the first FET 420 is minus the forward voltage (about 0.6V) of the diode 428 from the voltage V _{10 of} P ₁₀ by the action of the diode 428 and the resistor 429. It is biased to the side. Thus, source of the FET 41 420 - scan voltage so held by the first reference voltage V _{ref (1),} holding the voltage V ₁₀ of P ₁₀ to the first reference voltage V _{ref (1)} As a result, the gate voltage of the first FET 420 is biased to the minus side by the forward voltage of the diode 428 from the source voltage, and the gate-source voltage relationship of the second FET 421 is affected. It turns out that it will be the same as the staff. The same applies to the third FET 424, and a description thereof will be omitted.

From this point of view, first, in order to simplify the circuit analysis, first, the characteristics of the first and second FETs 420 and 421 are the same without variations. Further, the first FET 420 is considered to be equivalently a resistor, and its resistance value is defined as R _F1 , and the second F ET421 is also considered to be equivalently a resistance, so its resistance value is defined as R _F2. Then, R _F1 = R _F2 from the above precondition for the characteristic matching. Then, these two resistance values are defined again as R _F = R _F1 = R _F2 . Then, when the voltage V _{11 of} P ₁₁ , which is the connection point between the drain of the first FET 420 and the drain of the second FET 421, is obtained, it is obtained by the following equation 3.

Equation 3] _{V 11 = R F1 · V 6} / (R F1 + R F2) = 0.5V 6 wherein _{-R F · V 6 / 2R F} , V 6 is the output voltage of the differential amplifier 170a. Further, since the first FET 420 can be considered as a variable resistance and the second FET 421 can be considered as a fixed resistance, R _F2 becomes a fixed value and R _F1 and V ₆ can be considered as variables in _Formula 3. Therefore, for example, when V ₆ decreases, V ₁₁ is kept constant by increasing R _F1 to compensate for the decrease, and as a result, as described above, the gate voltage of the first FET 4 20 always remains at the source voltage. The forward voltage of the diode 428 is biased to the negative side from the negative voltage. The signals at P ₆ and P ₇ are substantially sine waves having a phase difference of π / 2 as shown in FIG. 11, and are basically the same as those in the embodiment shown in FIG. 1 ( (See FIG. 3).

In order to realize such an action, in the present embodiment, V₆Decreases, P₁~ P  Paying attention to the fact that the DC level at each point of 4 also decreases, the decrease in the DC level is detected by the adder 400, and the output voltage of the adder 400 as the detection result is detected by the differential amplifier circuit 402. 2 reference voltage V_{ref (2)}And the difference is P_TenBy changing the gate voltage of the first FET 420 (ie, V₆R can only compensate for the decrease in_F1Is equivalent to changing₁₁Is kept almost constant. In this embodiment, for example, the voltage V₆Is reduced by 20%, R_F1The resistances 413 to 417 of the differential amplifier circuit 402 are set so that the resistance value is about 1.67 times. Therefore, V₁₁= 1.67R_F1× 0.8V₆/(1.67R_F1+ R_F2), This value is about 0.5V₆Therefore, V₁₁Will be kept substantially constant. Note that the third and fourth FETs 424 and 425 have exactly the same operations as those of the first and second FETs 420 and 421, so description thereof will be omitted here.

As described above, in the embodiment shown in FIG. 9, the outputs of the light receiving elements 120a to 120d are caused by the change of the moving speed of the main scale and the change of the mounting posture of the detector. Even if the voltage at P _{1 to} P ₄ changes as a result of the change, by changing the equivalent resistance of the first FET 420 and the third FET 424 by the amount corresponding to this change, the inputs of the non-inverting amplifier circuits 430a and 430b are changed. Since the voltage is kept constant, that is, the output voltage is kept constant, an optical transformer tester that can always obtain a constant level output signal with a simple configuration without the need for a so-called feedback circuit is provided. The output device is to be provided.

In the embodiment shown in FIG. 9, the second FET 421 and the fourth FET 425 can be equivalently treated as fixed resistances, and therefore may be actually replaced by resistances instead of FETs. FIG. 12 shows still another embodiment of the embodiment shown in FIG. 9, and hereinafter, this embodiment will be focused on the points different from the embodiment shown in FIG. 9 with reference to FIG. I will explain about this.

In this embodiment, FETs connected in series (portions of the first and second FETs 420 and 421 and the third and fourth FETs 424 and 425 in FIG. 9) are provided in the inverting amplifier circuits 530a and 530b. The amplification degree of the circuits 530a and 530b is changed. That is, in FIG. 12, the inverting amplifier circuits 530a and 530b are provided with FETs for changing the amplification degree of the circuits. The specific configuration will be described with reference to the inverting amplifier circuit 530a. In the same circuit 530a, the first FET 520 and the second FET 521 are connected between the drains, while the source of the first FET 520 is the first reference. The voltage V _{ref (1)} is applied. The source of the second FET 521 is connected to the output of the operational amplifier 552.

Further, a resistor 558 as a feedback resistor is connected between the connection point between the drains of the first FET 520 and the second FET 521 and the inverting input of the operational amplifier 552. A diode 560 is connected between the source and the gate of the second FET 521 so that the source side becomes an anode, and the gate is connected to the ground potential via the resistor 562. Is held. The gate of the first FET 520 is connected to the output of the differential amplifier circuit 502 via the diode 528. The operation of the inverting amplifier circuit 530a in the above configuration will be considered as follows.

First, the first FET 520 and the second FET 521 may be considered as variable resistors and fixed resistors, respectively, which are the same as those of the embodiment shown in FIG. Now, the equivalent resistance value of the first FET 520 is defined as R _fa , the equivalent resistance value of the second FET 521 is defined as R _fb , the resistance value of the resistor 558 is defined as Ra, and the resistance value of the variable resistor 564 is defined as Rv. , R _v > R _fa , R _fb ) is satisfied, the output voltage V _out of the inverting amplifier circuit 530a is calculated by the following formula 4.

V _out = R _a (1 + R _fb / R _fa ) · V _in / R _v Here, V _in is the voltage applied to the inverting input of the operational amplifier 552 through the variable resistor 564. Further, R _fa is changed so as to compensate for the decrease in the voltage V ₁₆ of P ₁₆ and Vout becomes constant, which is the same as the embodiment shown in FIG. The output voltage of the inverting amplifier circuit 530a is also kept constant in the same manner as described above.

Note that, in the embodiment shown in FIG. 12, the second FET 521 and the fourth FET 525 can be equivalently treated as fixed resistances, so they may be actually replaced by resistors instead of FETs. In the present embodiment, the light emission current control is performed in an analog manner, but it is also possible to digitally measure the sine wave voltage amplitude of the light receiving element and control the light emission current. Further, although the linear encoder of the three-grating system has been described in the present embodiment, it is needless to say that the encoder of the normal two-grating system or a displacement detector such as a rotary encoder can be applied in the same manner.

[0045]

[Effect of the device]

 As described above, according to the optical displacement detection device according to the first and second aspects of the invention, the amount of the light emitting current supplied to the light emitting element is controlled so that the amplitude of the voltage waveform from the light receiving element is constant. Since the encoder is controlled, the size and cost of the encoder can be reduced, and the circuit configuration is simple and the displacement detection accuracy can be improved. Further, according to the optical displacement detection device according to the third and fourth aspects of the present invention, the amplification degree of the output stage is changed so that the output fluctuation of the light receiving element is compensated and the final output level becomes constant. With this configuration, an output signal with a substantially constant level can be obtained without using a feedback circuit, unlike the prior art, so it is possible to provide an optical displacement detection device with a simple configuration and high reliability. it can.

[Brief description of drawings]

FIG. 1 is an explanatory diagram of a circuit configuration of a linear encoder according to an embodiment of the present invention.

2 is an explanatory diagram of voltage waveforms at points P ₁ , P ₂ , P ₃ , and P ₄ shown in FIG.

FIG. 3 is an explanatory diagram of voltage waveforms at points P ₅ and P ₆ shown in FIG.

FIG. 4 is an explanatory diagram of a relationship between a moving speed of a reference grid and an amplitude of a voltage waveform.

FIG. 5 is an explanatory diagram of a mounting posture change of the detector.

6 is an explanatory diagram of voltage waveforms at points P ₅ , P ₆ , P ₇ , and P ₈ shown in FIG. 1.

7 is an explanatory diagram of rectified voltages at points P ₅ , P ₆ , P ₇ , and P ₈ shown in FIG. 1. FIG.

FIG. 8 is an explanatory diagram of a circuit configuration of a linear encoder according to another embodiment of the present invention.

FIG. 9 is an explanatory diagram of a circuit configuration of a linear encoder according to another embodiment of the present invention.

10 is a voltage waveform diagram of points P ₁ , P ₂ , P ₃ , and P ₄ shown in FIG. 9.

11 is a voltage waveform diagram of points P ₆ and P ₇ shown in FIG. 9.

FIG. 12 is an explanatory diagram of a circuit configuration of a linear encoder according to another embodiment of the present invention.

FIG. 13 is a conceptual explanatory diagram of a dencoder of a general 3-grid system.

14 is an explanatory diagram of a specific configuration of the encoder shown in FIG.

FIG. 15 is an explanatory diagram of a cross section taken along line I-I of FIG.

16 is an explanatory diagram of a main scale of the encoder shown in FIG.

FIG. 17 is an explanatory diagram of an index scale of the encoder shown in FIG.

[Explanation of symbols]

 12, 112 ... Light-emission-side grating 14, 114 ... Detection grating 16, 116 ... Reference grating 18, 118 ... Light-emitting element 20, 120 ... Light-receiving element 30, 130 ... Index scale 32, 132 ... Main scale 140 ... Current-voltage conversion circuit 150 Differential amplifier circuit 170 Differential amplifier circuit 200 Light emission current control circuit 400 Adder 402 Differential amplifier circuit 404 Output adjustment circuit

Claims (4)

[Scope of utility model registration request] 1. A main scale on which a predetermined reference grating is formed, an index scale on which a detection grating is formed and which is arranged in parallel so as to be movable relative to the main scale, and light emission for irradiating the reference grating with light. An optical displacement detecting device including: an element; and a light receiving element that receives light limited by the reference grating and the detection grating and outputs a relative movement amount of the main scale and the index scale as a substantially sine wave. An optical displacement detection device comprising light emission current control means for controlling the light emission current of the light emitting element so that the amplitude of the sine wave voltage output from the element is always constant. 2. The apparatus according to claim 1, wherein the index scale includes four detection gratings each shifted by π / 2 phase, and four light receiving elements installed for each detection grating, and the light reception is shifted by π phase. Π / each from the output of two sets of elements
Two phase-shifted A-phase and B-phase differential amplitude amplified outputs are obtained, and the light-emission current control means outputs an A ′ phase that is π-phase-shifted from the A-phase and a B′-phase that is π-phase-shifted from the B-phase. The obtained differential amplifier circuit and the total output obtained by half-wave rectifying the differential amplitude amplified outputs of the A phase, A'phase, B phase, and B'phase are compared with a reference voltage to obtain a difference current, and the difference voltage And a light emission current control circuit that controls the light emission current of the light emitting element in inverse proportion to the optical displacement detection device. 3. A main scale on which a predetermined reference grating is formed, an index scale on which a detection grating is formed and which is arranged in parallel so as to be movable relative to the main scale, and an optical scale on the reference grating. An optical displacement including a light-emitting element for irradiating the light and a light-receiving element for receiving the light limited by the reference grating and the detection grating and outputting the relative movement amount of the main scale and the index scale as a substantially sine wave. In the detection device, the optical displacement detection is provided with an output compensating means for compensating for a variation of the output level of the substantially sine wave signal output from the light receiving element and outputting the substantially sine wave signal of a constant level. apparatus. 4. The apparatus according to claim 3, wherein the index scale includes four detection gratings each shifted by π / 2 phase and four light receiving elements installed for each detection grating, and the index scale is shifted. From the output of two pairs of light receiving elements
The A-phase and B-phase differential amplitude amplified outputs with two phases shifted are obtained, and the output compensating means obtains an addition signal of the output signals of the four light receiving elements, an addition output of the addition circuit, and a reference signal A differential amplifier circuit that obtains a differential signal between the differential amplifier circuit and the differential amplifier circuit, and amplifies the A-phase or B-phase signal that changes so that the amplification level becomes constant according to the output of the differential amplifier circuit and outputs the amplified signal to the outside. An optical displacement detecting device, comprising: