JPH0578273B2 - - Google Patents

Description

[Detailed Description of the Invention] [Industrial Application Field] The present invention applies to transistors, FETs, GTOs, and
This invention relates to a resonant converter using a self-extinguishing switch element such as an IGBT.

[Problems to be solved by conventional technology and invention]

Traditionally, transistors, FETs, GTOs and IGBTs
Resonant converters using self-extinguishing switch elements, such as the It is used in various fields.

However, in such a resonant converter, in order to always maintain the operating mode in resonant mode even when there are power supply fluctuations or load fluctuations, the operating frequency must be controlled by detecting the circuit current, voltage, and its phase. It is. Therefore, the control circuit becomes complicated,
When controlling the output voltage to the lower limit, the operating frequency may extend into the audible range.

[Means for solving problems]

In order to solve the above-mentioned problems, the present invention configures a bridge circuit using self-extinguishing switch elements having anti-parallel diodes, and connects a series circuit of a resonance capacitor and a resonance inductance between the AC terminals of the bridge circuit. In a resonant converter in which a DC output is taken out from both ends of the resonant capacitor via a rectifying means, the resonant capacitor and resonant inductance are as follows during rated operation of the converter:
When the switch element is closed, the current rises from approximately 0, and when the switch element is opened, the current is approximately 0, and the terminal voltage of the resonant capacitor is in the range of approximately 1.5 to 2.1 times the converter power supply voltage. This paper proposes a resonant converter characterized by the following selection.

[Effect]

Since the present invention has the features described above,
Without adhering to the control concept of conventional resonant converters,
The operating frequency is kept almost constant, and complete resonance mode is only used during rated operation.Output control in response to power supply fluctuations and load fluctuations is achieved by controlling the on-time of the switch element.
Or, it is performed by vector control, ie phase shift control, etc. of two bridge arms.
Under frequently rated operating conditions, when the switch element closes, the current rises from almost 0, and when the switch element opens, the current drops to almost 0, demonstrating the advantages of a resonant converter.

〔Example〕

FIG. 1 is a diagram showing the principle of a resonant converter according to the present invention. In the figure, E is a DC power supply, S1 to S4 are switch elements bridge-connected across the power supply E, C1 is a resonant capacitor, L1 is a resonant inductance, and LS is the excitation inductance of the output transformer (not shown). , are connected in parallel to the resonance capacitor C1. B is a bridge rectifier circuit, C2 is a filter capacitor, and RL is a load.

In such a circuit, switch elements S1 to S
4 is turned on alternately with a pulse width of T/2 with a 180 degree shift with the period being approximately constant T as shown in Fig. 2 1 and 2, and the AC output points X and Y of the bridge are connected as shown in Fig. 2 3. Add a rectangular AC voltage Vxy. In such a state, if the values of resonance C1 and L1 and the load condition are changed, the operation mode of the inverter changes significantly, and the forward current and reverse current flowing through S1 to S4 change. In the present invention, as described later, L1,
By selecting the value of C1, the current flowing through S1 to S4 can be adjusted to the value shown in Figs. 2, 4 to 5 in the rated operating state.
As shown in the figure, the current rises from almost 0 and has a peak value Ip in phase with the voltage Vxy (when the switch elements S1 to S4 are closed, the current rises from almost 0 to the current of the switch elements S1 to S4).
The forward current is set to a characteristic in which the current is almost 0 when the cap is open), and the voltage Vc1 of C1, that is, the equivalent output voltage V0, is increased to a value higher than E, about 1.5 to 2.1E, as shown in Fig. 2. let In addition, filter C
If 2 is large enough, the bridge B and C2, RL seen from both ends of C1 can be considered as a bipolar battery with a voltage Vo, and the voltage of C1 is clamped at Vo. By selecting L1 and C1 in this way, in a heavy load state, Is1 and Is3, for example, will lag behind Vxy as shown in Figure 2 7, and in the case of an output short circuit, the current will lag 90 degrees with Vxy as shown in 8. It becomes a positive and negative triangular wave. In Figure 2 7 and 8, the negative current indicates the reverse current of each switch element, that is, the current that returns to the power supply.The heavier the load, the more the feedback current increases, and in a short circuit state, the forward current and feedback current peak.
Isp becomes almost equal, and all the power supplied from the power supply to the resonant circuits of L1 and C1 is fed back to the power supply.

On the other hand, when the load is light, the current flowing through the switch elements S1 to S4 becomes a leading current with respect to Vxy as shown in FIG. The peak values of the forward current and reverse current become approximately equal, and the power supplied from the power source is returned to the power source as is.

However, in general, due to the inherent characteristics of a resonant converter, the no-load voltage or light-load voltage increases with this resonant converter. need to be suppressed.

In this resonant converter, the following formula is expressed as follows, with the current in the switch element maintained at a chevron characteristic (when the switch element closes, the current rises from almost 0, and when the switch element opens, the current becomes almost 0): Let the rate of increase of the output voltage be k (k=Vo/E), and the excitation inductance of the output transformer Ls
This is the circuit analysis equation when is set to infinity.

PoT−kIpE(T/2−t1)=0 (1) Ip+E/L(1−k)(T/2−t1)=0 (2) (1−k)(1+k)cosωt1=0 (3) Ip −ωCE(1+k)sinωt1=0 (4) However, ω=1/√, Po=output power, t1: first half of the conduction period of the switch element These equations are relatively simple when Ls=infinity. However, it becomes very complicated when considering the excitation inductance Ls. Therefore, the formula is not shown here, but it is a calculation formula with excitation inductance Ls, and as an example, Po = 1000W, E =
The computer calculation results at 240V and T=50μs are illustrated in FIG. 3 with k as a variable and Ls as a parameter. Note that this calculation result is in good agreement with the experiment, which proves the validity of this analytical calculation.

From Figure 3, as k increases, the resonant inductance L1 increases linearly without much relation to the excitation inductance Ls, and the resonant capacitor C1 increases rapidly at k=2, and then gradually decreases. . This resonance capacitor C1 varies greatly depending on the excitation inductance Ls.

Also, Ip decreases gently as k increases,
It is not affected by Ls very much. By the way, since the peak Isp of the current flowing through the switch element when the load is short-circuited is expressed as Isp=E/L·T/4 as shown in FIG. 2, this will be graphed.

That is, when k is small, Isp increases rapidly.
In an actual inverter, the current utilization rate of the switch element is highest when Isp = Ip, so it is reasonable for k to be in the range of k = 1.5 to 2.1, especially k = 1.8, as Isp = Ip.

Furthermore, the volume of the iron core used for the inductance L1 is proportional to the larger of LI ² p or LI ² sp if the materials are the same. In the case of air-core inductance, this is proportional to the volume occupied by the inductance. LI ² shown in FIG. 3 is a graph of the larger of LI ² p or LI ² sp, and is minimum around k=1.8. this
LI ² sp or LI ² p are not significantly related to Ls. Also, when k = 1.5 to 2.1, the short circuit current/rated current ratio Isp/Ip
1.4 to 1.7, which is suitable for applications where a large short-circuit current is preferred, such as capacitor chargers. Also, since L1 has little relation to Ls, the above equation (1)
You may apply the solution of ~(4) as is. Especially k=
L1 in 1.8 can be roughly estimated as follows.

A simple method for determining L1 (K = 1.8) The average output current Io is Io = Po / kE = Po / 1.8E The average output current Iso during short circuit is from the graph Iso = IoX1.5 Iso = 1 / 2. Since Isp, Isp=2×1.5×Io =2×1.5×Po/1.8E =1.7Po/E Isp=E/L・T/4, so ET/4L=1.7Po/E L1≒0.15E ² T/Po C1 is determined by adjustment corresponding to this L1.

FIG. 4 shows an embodiment in which the resonant converter of the present invention is applied to a high voltage charger for a capacitor. E is a DC power supply, Q1 to Q4 are FETs bridge-connected across the DC power supply E, D1 to D4 are each FET,
Diode C1 connected in antiparallel to Q1 to Q4
is a resonant capacitor, L1 is a resonant inductance, T1 is a step-up transformer, CW is a high voltage rectifier circuit such as a Kotscroft-Walton circuit connected to the secondary side of T2, and Co is a capacitor to be charged.

The charging voltage of the capacitor Co is detected by the detection resistor R1, and the reference voltage is detected by the error amplifier A1.
An error signal is generated by comparing it with Vref, and in response to this error signal, gate signals G1 to G4 for each FET are generated in the gate signal generation circuit A2.

Here, so that the current of FETQ1 to Q4 becomes a mountain shape, and when the power supply voltage is E, 1 of transformer T1
L1 and C1 are adjusted so that the next input voltage is 1.8E.

In such a configuration, when a charging start command is received, until the charging voltage Vref is reached, the error amplifier uses FETs, Q1 to Q4 have the maximum pulse width, approximately T/2.
Gate signals G1 to G4 are applied to turn on.
The capacitor Co is then equivalent to a load short circuit initially, so the converter supplies an output current of about 1.7 Io, as described above. As shown in Figure 5, this current decreases as the capacitor Co is charged, and the charging voltage increases to
When this happens, it becomes the rated operating state with the rated current Io. When the charging voltage reaches Vo, the error amplifier stops the gate signal and stops charging. This charging time tc can be made shorter than the charging time tc' by using a normal constant current Io. In reality, the voltage of the capacitor Co decreases due to parasitic current in the circuit, so it must be supplemented to compensate for this amount.

In this case, since the amount of supplied energy is large when the T/2 pulse width signal is intermittent, the gate signals G1 to G4 are
pulse width control, or as shown in Figure 6, gate signals G1 to G4 are sent at T/2, 180 degrees, and gate signals G2 and G3 are similarly sent at T/2, 180 degrees.
Vector control that shifts the two sets of phases Φ is suitable.

Such a control method is effective when the resonant converter of the present invention is used in other than a capacitor charger. In particular, vector control has the advantage that the change in output with respect to phase Φ is relatively linear and control characteristics are good.

Above, the bridge circuit has been explained in the embodiments, but for example, the switch element S in FIG.
By replacing 1 and S4 with capacitors, it can be implemented in a half-bridge type.

〔Effect of the invention〕

Since the present invention is configured as described above, it has the following effects and advantages.

(1) Since the transformer input voltage Vc1 can be made 1.58.1 times the power supply voltage, the transformer input current is reduced and efficiency is high. Especially in the case of high-voltage power supplies, the number of transformer turns can be reduced and the coupling ratio can be improved.

(2) The size of the resonance inductance L1 including the iron core can be minimized.

(3) Since the drive frequency of the converter is almost constant, the drive control circuit is simplified. Since the driving frequency is almost constant, there is little interference with other circuits. Even when the output voltage is controlled to the lower limit, the operating frequency does not reach the audible range.

(4) The rated current Ip and the short-circuit current Isp are almost equal, and the current utilization rate of the switch element is good. No special overcurrent protection circuit is required.

(5) The output current increases to the required value when the load is short-circuited, and the no-load voltage is high, making it suitable for, for example, capacitor chargers.

(6) The distributed capacitance of the transformer's secondary winding can be incorporated as part or all of the resonant capacitor, and the distributed capacitance of the transformer's secondary winding, which was a disadvantage in conventional high-voltage output converters, can be actively used. It can be used effectively.

[Brief explanation of the drawing]

FIG. 1 shows a principle diagram of the resonant converter circuit according to the present invention, FIG. 2 shows a waveform diagram of each part to explain the operation of the circuit shown in FIG. 1, and FIG. Figure 4 shows an example in which the resonant converter of the present invention is applied to a high voltage charger for a capacitor.
5 shows the current waveform flowing through the capacitor Co of the circuit shown in FIG. 4, and FIG. 6 shows the gate signal G of each switching element Q1 to Q4 of the circuit shown in FIG.
1 to G4 vector control mutual waveform diagrams are shown. E...DC power supply, B...Bridge rectifier circuit, S
1 to S4...Switch element, C1...Resonance capacitor, L1...Resonance inductance, RL...
...Load, C2...Filter capacitor, Q1~Q
4...FET, D1-D4...Diode, T1
...Step-up transformer, CW...High voltage rectifier circuit, Co
... Capacitor, R1 ... Detection resistor, Vref ...
Reference voltage, A1...Error amplifier, A2...Gate signal generation circuit, G1-G4...Gate signal.

Claims (1)

[Scope of Claims] 1. A bridge circuit is constituted by a self-extinguishing switch element having an anti-parallel diode, and a series circuit of a resonance capacitor and a resonance inductance is connected between the AC terminals of the bridge circuit, and the resonance In a resonant type converter in which DC output is taken out from both ends of a resonant capacitor via a rectifying means, the value of the resonant capacitor and the value of the resonant inductance are set such that the switch element closes when the converter is in rated operation. When the current rises from approximately 0, when the switch element opens, the current becomes approximately 0, and the terminal voltage of the resonant capacitor is approximately 1.5 to 2.1 of the converter power supply voltage.
A resonant converter characterized by being selected to have a double range.