JPH0557826B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION <Industrial Application Field> The present invention relates to improvement of control characteristics of a current resonant converter.

<Conventional technology> Conventional current resonant converters utilize LC resonance to convert changes in the current of switching elements such as MOSFETs into smoothly changing sinusoidal waveforms for switching. The current waveform is a sine wave (resonant waveform), and switching can be performed when the resonant current becomes zero, so it is characterized by low noise and switching loss during switching. Therefore, this current resonant converter is considered to be an effective method for reducing the noise and increasing the frequency of the switching power supply (related to miniaturization of the device).

The output voltage in this current resonant converter is controlled by the ratio of the switching frequency fs to the resonant frequency fn (f
s/fn), and the relationship is expressed by the following equation.

V ₀ /Vi=fs/fn... However, Vi: Input voltage (DC) V ₀ : Output voltage (DC).

<Problem to be solved by the invention> However, in the current resonant converter shown in the above-mentioned prior art, since a fixed inductor or a fixed capacitor is used as a resonant element, the resonant frequency fn shown in the above equation is constant. , in order to control the output voltage V ₀ , the switching frequency fs
It was necessary to change. Therefore, since the switching frequency fluctuates due to load fluctuations and input fluctuations, the output filter must be designed at the lowest frequency, making it impossible to make the device compact. Furthermore, when converters are operated in parallel, beats and malfunctions are likely to occur. Furthermore, since the switching frequency varies over a wide range, noise is distributed over a wide range, making it difficult to reduce the noise.

The present invention has been made based on the problems of the prior art described above, and in a current resonant converter,
The objective is to provide a current resonant converter that can control the output voltage while keeping the switching frequency fixed, reduce the dynamic range of output voltage control, and improve output voltage accuracy and transient response characteristics. be.

<Means for Solving the Invention> The configuration of the present invention for solving the above problems is that in a current resonant converter that controls the output voltage to a constant value in response to input fluctuations and load fluctuations, two variable inductors and a capacitor are used. an LC resonant circuit consisting of;
An output voltage control circuit that increases or decreases the control current of one of the variable inductors in accordance with the difference between the output voltage and a reference voltage, and an input voltage control circuit that increases or decreases the control current of the other variable inductor in accordance with the input voltage. The present invention is characterized in that the output voltage is controlled to a constant value by changing the inductance values of the two variable inductors and changing the resonant frequency using a control current.

<Function> According to the present invention, two variable inductors for resonance are used.
By dividing the switching frequency into two, one corresponds to the input voltage, and the other corresponds to the output voltage, and by changing the inductance value and changing the resonant frequency, the output voltage is controlled to a constant value while the switching frequency remains constant. In addition, by changing the inductance in a feedforward manner from the input voltage, the controllability of the feedback loop can be improved.

<Example> Hereinafter, the present invention will be explained based on the drawings.

FIG. 1 is a block diagram showing one embodiment of a current resonant converter according to the present invention, and shows all waveforms of a half-bridge circuit type converter.

In Figure 1, Vi is the input voltage, 1 is an input smoothing capacitor, 2 and 3 are dividing capacitors that divide the input voltage Vi into two, and 4 and 5 are switching transistors ( (hereinafter simply referred to as a transistor), 6 is a resonant variable inductor with one end of the inductor winding connected to the connection point of the split capacitors 2 and 3, and 7 is an inductor connected to the other end of the inductor winding of the resonant variable inductor 6. A variable inductor for resonance to which one end of the primary winding is connected, 8 a transformer to which both ends of the primary winding are connected to the connection point of the transistors 4 and 5 and the other end of the inductor winding of the variable inductor 7, and 9 a transistor. 8 is a resonance capacitor connected to both ends of the primary winding, 10 and 11 are rectifier diodes whose anode sides are connected to both ends of the secondary winding of the transformer 8, and 12 is a rectifier diode 10 and 11. A chiyoke coil whose one end is connected to the cathode side, 13 and 14 are an output smoothing capacitor and a load resistor connected between the other end of the chiyoke coil 12 and the midpoint of the transformer 8, respectively.
3 constitutes an output filter. V ₀ is load resistance 1
is the output voltage applied across 4. Further, 15 is an input voltage control circuit that applies a control current corresponding to a change in the input voltage Vi to the control winding of the variable inductor 6;
16 is an output voltage control circuit that applies a control current corresponding to the difference between the output voltage V ₀ and the reference voltage to the control winding of the variable inductor 7. Further, V _G1 and V _G2 are the gate drive voltages of the transistors 4 and 5, V _DS is the drain-source voltage of the transistor 4, V _c is the voltage across the resonance capacitor 9, I _l is the resonance variable inductor 6, This is the current flowing through 7.

In addition, Figure 2 is a configuration diagram (Figure A) and an equivalent circuit diagram (Figure B) of the variable inductors 6 and 7 used in Figure 1.
and a characteristic diagram (Figure C).

In Fig. 2 A, b1-b2 is wrapped around the central leg.
A control current _c1 ( _c2 ) is passed through the winding, and a1-a2
The windings are wound around the two legs sandwiching the center leg so that the magnetic fluxes generated cancel each other out at the center leg.
The equivalent circuit is shown in figure (b), and b1-b
The voltages applied to the two windings cancel each other out and become zero in the a1-a2 windings. Its characteristics are as shown in (c) the control current flowing through the b1-b2 windings.
By _c1 ( _c2 ), the core is saturated and b1-b2
The inductance L of the winding changes. In FIG. 1, the configuration shown in FIG. 2 is used for input and output, respectively. Note that the variable inductor is not limited to the one shown in FIG. 2, and may be any other type as long as the inductance can be made variable by controlling the control current.

3 and 4 are configuration diagrams showing specific examples of the input voltage control circuit 15 and the output voltage control circuit 16 used in FIG. 1.

In the input voltage control circuit 15 shown in FIG. 3, the input voltage Vi is divided by voltage dividing resistors R ₁ and R ₂ and a functional operation is performed on the divided output voltage E _i . The output E ₀ is current-amplified by a circuit consisting of an operational amplifier OP ₂ , a transistor Q ₁ , and a resistor R _fi , and is used as a control current _c1 that drives the control winding of the variable inductor 6. On the other hand, the output voltage control circuit 1 shown in FIG.
6, resistors R ₁ to R ₄ and operational amplifier OP ₁
The difference between the output voltage V ₀ and the reference voltage V _r is taken and amplified by a subtraction circuit consisting of the following. The output of the op amp
A circuit consisting of OP ₂ , transistor Q ₁ , and resistor R _f0 performs current amplification to provide a control current _c2 that drives the control winding of variable inductor 7.

FIG. 5 is an operation waveform diagram for explaining the operation of FIG. 1. When the transistor 4 is turned on, that is, the gate drive voltage V _G1 of the transistor 4 becomes high level, a resonant current _l flows through the resonant variable inductors 6 and 7 and the resonant capacitor 9 in the direction of the arrow shown in FIG. Since this resonant current I _l is in a resonant state, it becomes sinusoidal and returns to zero again, but it continues to flow in the opposite direction due to the parasitic diode (not shown) of the transistor 4. The negative part of the resonant current _l is a state in which energy is regenerated to the input. In this state, the drain-source voltage V _DS of transistor 4 is very small due to the parasitic diode, and no current flows through transistor 4, so if transistor 4 is turned off in this state, switching loss will be reduced. becomes very small. Transistors 4 and 5 operate symmetrically, and currents and voltages in opposite directions are generated in the transformer winding when transistor 4 is on and when transistor 5 is on. This is rectified on the secondary side to provide an output voltage _V0 .

Here, FIG. 6 is a diagram showing input/output characteristics,
The horizontal axis shows the switching frequency fs and the resonance frequency fn.
The ratio (fs/fn) is taken, and (2N·V ₀ /Vi) is taken and expressed on the vertical axis. Note that N is 1 of transformer T.
It is the ratio (n1/n2) of the number of turns _n1 on the secondary side and the number n2 of turns on the secondary side. Further, r=R ₀ /Z _o is a normalized resistance, which is assumed to be the load resistance 14, and Z _o is a characteristic impedance, and Z _o =√r _r L _{r =} L _r1 +L _r2 .

As shown in Figure 6, the load is increased by 10 times (r=1→r=
10), the input/output characteristics hardly change, indicating that it is resistant to load fluctuations. But the output voltage
V ₀ is proportional to the input voltage Vi, and the effects of input fluctuations appear directly on the output.

The relationship in Figure 5 is 2N・V ₀ /Vi=k・fs/fn... However, k can be expressed as a constant of proportionality, but this equation can be transformed as shown in the following equation.

2N・V ₀ /Vi =k・fs・2π√ _{r r} … However, fn=1/2π√ _{r r} .

From this above equation, if √ _r can be made proportional to 1/Vi, the output voltage V ₀ will not depend on the input voltage Vi and will be constant. In other words, it is possible to realize a device that is not affected by input fluctuations and load fluctuations. In conventional devices, input fluctuations are also stabilized by feedback, so a large dynamic range of feedback is required. Therefore, the loop gain is sacrificed, resulting in problems such as a drop in output voltage accuracy or poor transient response characteristics. Therefore, in the present invention, as shown in FIG.
The variable inductor is divided into two parts, and one variable inductor 6 is changed in accordance with the input voltage Vi.

Here, in the input voltage control circuit 15 shown in FIG. 3, a functional calculation is performed so that the relationship between the input voltage Vi and the control current _c1 becomes _c1 = f(Vi) . Also, in the characteristic diagram of the variable inductor shown in Figure 2 C, if the relationship between the inductance L and the control current _c1 is L = g (I _c1 )..., then from the formula and formula, L = g {f (Vi)} By selecting g and f such that = k/(Vi) ² , the output voltage V ₀ becomes independent of the input voltage Vi from the above equation. That is, by the input voltage control circuit 15,
By applying such feedback, the burden on the feedback loop is greatly reduced.

In addition, since the full waveform current resonant converter is almost unaffected by load fluctuations, the control range of the feedback loop can be made considerably smaller by eliminating the influence of input fluctuations using feed forward. Become. (Control range of variable inductor 7 << Control range of variable inductor 6) Furthermore, since controlling the variable inductor becomes an inductance load, reducing the control current to reduce loss in the drive circuit increases the number of turns of the control winding.
Although the inductance increases and the transient response characteristics worsen (although the magnitude of the current change decreases),
Since the control range is small, the number of turns of the control winding can be reduced, and the transient response characteristics can be accelerated.

<Effects of the Invention> As described above in detail with the embodiments, according to the present invention, the output voltage can be controlled to a constant value while the switching frequency remains constant. It is possible to eliminate the occurrence of beats and malfunctions that occur when operating in parallel.

(2) Since the output filter can be designed to match the switching frequency, noise countermeasures can be easily taken.

(3) By increasing the switching frequency, the output filter can be made smaller.

Also, by setting up a feed forward loop,
Since the configuration is designed to absorb input voltage fluctuations, (4) the variable inductance of the feedback loop can be reduced and the transient response characteristics can be improved.

(5) Input changes are generally gradual and transient response characteristics are not much of a problem, so the number of turns of the variable inductance on the input side can be increased and the control power can be reduced.

(6) Since the control range of the feedback loop is small, there is no need to increase the open loop gain of the feedback loop to a large value, realizing a current resonant converter that can improve loop stability and output voltage accuracy. I can do it.

[Brief explanation of the drawing]

Fig. 1 is a block diagram showing one embodiment of the current resonant converter according to the present invention, Fig. 2 is a block diagram, equivalent circuit diagram, and characteristic diagram of the variable inductor used in Fig. 4 is a configuration diagram of the output voltage control circuit used in FIG. 1, FIG. 5 is an operation waveform diagram for explaining the operation of FIG. 1, and FIG. The figure is an input/output characteristic diagram of FIG. 1. 6, 7...Variable inductor for resonance, 9...Capacitor for resonance, 15...Input voltage control circuit, 16...Output voltage control circuit, Vi...Input voltage, _V0 ...Output voltage.

Claims (1)

[Claims] 1. In a current resonant converter that controls the output voltage to a constant value in response to input fluctuations and load fluctuations, 2.
an LC resonant circuit consisting of two variable inductors and a capacitor, an output voltage control circuit that increases or decreases the control current of one of the variable inductors with respect to the difference between the output voltage and a reference voltage, and an output voltage control circuit that increases or decreases the control current of the other variable inductor to the input voltage. an input voltage control circuit that increases or decreases the input voltage according to the control current, and changes the inductance value of the two variable inductors according to the control current,
A current resonant converter characterized by being configured to control the output voltage to a constant value by changing the resonant frequency.