JPH0542163B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION Field of Industrial Application The present invention relates to an oscillation device that can be used in transmitters and receivers of televisions, radios, stereo tuners, personal radios, and other communication devices in general.

Conventional configurations and their problems In recent years, the number of broadcast waves from televisions and radios and communication waves from communication devices has increased, and the performance of the oscillator that generates the signal at the desired frequency in the local oscillator requires high tuning accuracy, Stability and reliability requirements are increasing. On the other hand, reducing the manufacturing costs of these receivers, transmitters, and communication devices is also a major issue, and drastic technological development is required especially for frequency selection circuits in the high frequency section, which is difficult to rationalize.

A conventional oscillation device will be described below with reference to the drawings. Figure 1 is a circuit diagram of a conventional oscillation device, where 1 is a tuning coil, 2 is a trimmer capacitor,
Reference numeral 3 denotes voltage variable capacitance diodes, each of which constitutes a tuning circuit 4. The voltage variable capacitance diode 3 was supplied with a voltage obtained by variably dividing the voltage of a DC power supply 6 by a potentiometer 7 via a resistor 5 for blocking an AC signal. A portion of the tuning coil 1 of the tuning circuit 4 is connected to a feedback amplifier 8 to oscillate a signal, and the oscillation frequency is controlled by controlling the potentiometer 7.

Furthermore, FIG. 2 is a conventional component configuration diagram for the tuning circuit 4 in FIG. 1, where 9 is a tuning coil, 1
0 is a trimmer capacitor, 11 is a voltage variable capacitance diode, and each is a circuit conductor 12.
and 13 were connected.

However, in the above configuration, the inductor parts and capacitor parts are larger in size than other high-frequency parts, and the height dimension in particular hinders miniaturization and thinning of the device.

The inductance of an inductor component tends to shift due to mechanical vibration, and since the ferrite core has a large temperature dependence, the inductance is unstable and the tuning frequency fluctuates greatly.

The inductor component and capacitor component exist as separate components and are connected by a conductor routing circuit, which causes a large amount of lead inductance and stray capacitance, making circuit operation unstable.

Since it is a collective circuit of individual parts with independent minimum unit functions, there are limits to reducing the number of parts and rationalizing manufacturing.

Furthermore, the control voltage for the voltage variable capacitance diode is unstable, thus significantly degrading the tuning accuracy.

It is not possible to respond to the industrial trend of digitization and LSI as configuration technology for control systems, and it is not possible to realize advanced multi-functional control of oscillators and equipment that use them.

It had the following problems.

Purpose of the Invention The purpose of the present invention is to realize a tuned circuit block that integrates an inductor part and a capacitor part, and to make it possible to control the oscillation frequency of an oscillator including the integrated tuned circuit block using a digital signal. This allows the form of the tuned circuit block to be ultra-thin and compact, and is also stable against mechanical vibrations, improves the tuning accuracy of the oscillation frequency, has small temperature dependence of the oscillation frequency, and reduces the need for connecting leads. The goal is to eliminate negative effects and be stable at high frequencies, and to reduce the number of parts and streamline the manufacturing process.

Structure of the Invention The oscillation device of the present invention is configured such that the ground terminals of the electrodes that are placed opposite to each other via a dielectric or on the surface of the dielectric are set to face oppositely to each other, and any one of the electrodes is set to face oppositely. A voltage variable reactance element is connected to the open terminal of the electrode, and the tuned control output voltage of the control section consisting of a PLL circuit including a variable frequency divider is supplied to the voltage variable reactance element, and the required parts of each of the electrodes are connected. It is connected to the oscillation element of the control section and configured to control the division ratio of the variable frequency divider using a frequency setting code, so that one of the opposing electrodes acts as a distributed inductor. In addition, by facing this electrode and the other electrode, a distributed constant circuit with an open tip is formed, thereby realizing distributed capacitance due to the generated negative reactance, and acting in parallel with the above-mentioned distributed inductor. This is a basic tuning circuit, and by applying the tuning control output voltage of the PLL circuit as the control voltage of the voltage variable reactance element connected to this tuning circuit, the digital code that is the oscillation control signal is set and the oscillation frequency is set. This is used to perform variable control.

DESCRIPTION OF EMBODIMENTS Hereinafter, embodiments of the present invention will be described with reference to the drawings.

FIG. 3 shows a configuration circuit diagram of an oscillation device in an embodiment of the present invention. 14 is a transmission line electrode that acts as a distributed inductor, and 15 is a transmission line electrode that is disposed opposite to the transmission line electrode 14 via a dielectric (not shown) and generates a distributed capacitor. The grounding of the respective transmission line electrodes 14 and 15 is set in opposite directions (the positions of the respective grounding terminals or the common terminal are not symmetrical with respect to the center plane between the respective electrodes). A parallel circuit 16 of an inductor and a capacitor is formed. A parallel circuit 16 of an inductor and a capacitor is thus formed. A voltage variable capacitance diode 18 is connected to the open terminal 17 of the transmission line electrode 15 to constitute a tuner 19. The voltage variable capacitance diode 18 is connected to the PLL circuit 2 via a resistor 20 for blocking AC signals.
The output voltage of one low-pass filter 22 is supplied. On the other hand, the open terminal 17 of the transmission line electrode 15 of the parallel circuit 16 of an inductor and capacitor in the tuner 19 is connected to the feedback amplifier 23 to form an oscillation circuit, and a part of the oscillation signal output of the feedback amplifier 23 is
It is supplied to the fixed frequency divider 24 of the PLL circuit 21 and divided, and the divided output is further supplied to the variable frequency divider 25 and input to the digital tuning control signal input terminal 26.
The frequency is variably divided according to the digital code input to the . The phase of the frequency-divided output is compared with the oscillation output of the crystal oscillator 27 by a phase comparator 28, and the phase comparison detection signal is supplied to the low-pass filter 22.

FIG. 4 shows a configuration circuit diagram of an oscillation device in another embodiment of the present invention. 19 is the third
It is the tuner explained in the figure, and 21 is the PLL as well.
23 is also a feedback amplifier. On the other hand, the digital tuning control signal input terminal 26 of the variable frequency divider 25 has a latch, RAM or
The output of a digital signal processor 29 consisting of a ROM is supplied. This digital signal processor 29 functions to store a digital signal code for tuning control inputted to an input terminal 30 or to convert it into another digital signal code.

FIG. 5 shows a configuration circuit diagram of an oscillation device in another embodiment of the present invention. tuner 19,
The PLL circuit 21, feedback amplifier 23 and digital signal processor 29 are the same as those explained in FIG. 4 above. On the other hand, digital signal processor 2
The output of the code converter 31 is supplied to the input terminal 30 of 9. This code converter 31 has an input terminal 3
It acts to convert the serial format digital signal code for tuning control inputted into the parallel format digital signal code.

Figures 6 to 13 are the same as Figures 3 to 5 above.
This figure shows an embodiment of the parallel circuit 15 of an inductor and a capacitor explained in the figure. In FIG. 6, a shows a front view, b shows a side view, and c shows a back view.
(The same applies to Figures 7 to 13 below) 6th
In the figure, 100 is a dielectric substrate, and 101 and 102 are electrodes forming a distributed constant circuit to realize a distributed inductor and a distributed capacitor. Electrode 1
The ground terminals 01 and 102 are set so that the opposing electrodes are on arbitrary opposite sides as shown in FIG.

(Similarly in Figures 7 to 13 below) A side, B shown in Figure 6a and A side, B shown in Figure 3c
correspond to each other. (Figures 7 to 13 below)
(Same in the figure) In FIG. 7, 1
Electrodes 104 and 105 having bent portions are placed opposite each other.

In FIG. 8, electrodes 107 and 108 each having a plurality of bent portions are placed facing each other with a dielectric substrate 106 in between.

In FIG. 9, meander-shaped electrodes 110 and 111 are placed facing each other with a dielectric substrate 109 in between.

In FIG. 10, spiral-shaped electrodes 113 and 114 are placed facing each other with a dielectric substrate 112 in between.

In FIG. 11, electrodes 116 and 117 are installed on the surface of a dielectric substrate 115, facing each other laterally.

In FIG. 12, electrodes 119 and 120 are provided inside a dielectric substrate 118, facing each other.

In FIG. 13, an electrode 122 is installed inside a dielectric substrate 121, an electrode 123 is installed on the surface of the dielectric substrate 121, and the electrodes 122 and 123 are opposed to each other.

In the embodiments shown in FIGS. 6 to 13 above, the electrodes installed opposite each other had the same shape and completely faced each other on the entire surface, but even if one arbitrary electrode has a different equivalent length compared to the other electrode, It can also be realized by partially opposing the partner electrodes. Further, the shapes of the electrodes used in the embodiments shown in FIGS. 11 to 13 can also be realized using the shapes shown in the embodiments shown in FIGS. 7 to 10.

Next, the operating principle of this tuner will be explained.

FIGS. 14a to 14e are equivalent circuits for explaining the operation of the tuner used in the oscillation device of the present invention. In FIG. 14a, a signal source that generates a voltage e is applied to a transmission path formed by transmission path electrodes 270 and 271 having an electrical length l and having their ground terminals set in opposite directions. 272 is connected to the transmission line electrode 270 to supply a signal. Accordingly, the transmission line electrode 270
It is assumed that a traveling wave voltage e _A is excited at the open terminal at the tip of . On the other hand, the transmission line electrode 271
are disposed close to and opposite to the transmission line electrode 270 or in parallel, so that a voltage is induced by mutual induction. Let e _B be the traveling wave voltage induced in the open terminal at the tip of the transmission line electrode 271.

Since the respective ground terminals of the transmission line electrodes 270 and 271 are set in opposite directions, the induced traveling wave voltage e _B has an opposite phase to the excited traveling wave voltage e _A. Since the tips of the transmission paths of the traveling wave voltages e _A and e _B are open, voltage standing waves are formed in the transmission path formed by the transmission path electrodes 270 and 271. Here, the voltage distribution coefficient indicating the distribution mode of the voltage standing wave in the transmission line electrode 270 is K.
The voltage distribution coefficient at the transmission line electrode 271 can be expressed as (1-K).

Next, the potential difference V generated at any opposing portion of the transmission line electrodes 270 and 271 can be expressed as V=Ke _A -(1-K)e _B (1). Here, if each transmission line electrode 270 and 271 has the same electrical length l, then e _B = -e _A ...(2), so the potential difference V in the first equation is V = Ke _A + (1 -K) e _A = e _A ...(3). That is, a potential difference V can be generated in all parts where the transmission line electrodes 270 and 271 face each other.

Here, it is assumed that the transmission line electrodes 270 and 271 have an electrode width W (the thickness of the electrodes is thin), and are opposed to each other at a distance d via a dielectric material having a dielectric constant ε _s . . In this case, the capacitance C _O formed per unit length of the transmission path is C _O =Q/V=Q/e _A ...(4) Q=ε _{p ε s} W・V/d=ε _p ε _s W・e _A /d ...(5), and therefore C _O =ε _p ε _s W/d ...(6).

Therefore, the transmission line shown in FIG. 14a becomes a transmission line including a distributed capacitor 273 of C _O determined by the formula 6 per unit length as shown in FIG. 14b. Furthermore, as shown in FIG.
78 and a distributed capacitor 273.

Next, the relationship between the formation of the distributed capacitor 273 and the electrical length l of the transmission path will be explained.
The characteristic impedance Z _O per unit length in a balanced mode transmission line as shown in Figure 15a is
It can be expressed by the equivalent circuit shown in Figure b. Its characteristic impedance Z _O is generally becomes. If the transmission path is lossless, then becomes. This assumption can be applied to many of the embodiments of the tuner of the present invention, and to simplify the explanation, the characteristic impedance Z _O
Use. The capacitance C _O in the eighth equation is the same as the capacitance C _O per unit in the path determined in the sixth equation. In other words, the characteristic impedance per unit length of the transmission line Z _O
is a function of the capacitance C _O , which is also a function of the permittivity ε _s of the dielectric material involved in the capacitor C _O , the width W of the transmission line electrodes, and the installation spacing d of the respective transmission line electrodes.

As described above, the characteristic impedance per unit length in the transmission line is Z _O , the electrical length is l, and the equivalent reactance X generated at the terminal of the transmission line with the end open is X = -Z _O cotθ can be expressed as (9). Here, θ=2πl/λ...(10), and especially in the case of θ=O~π/2 θ=π~3/2π...(11), the equivalent reactance X is X≦O...(12) becomes. That is, the equivalent reactance at the terminal of the transmission line can be the capacitance reactance. Therefore, depending on the electrical length l of the transmission path, θ becomes the
When formula 11 is satisfied, for example, a capacitor can be formed by setting the electrical length l to λ/4 or less. And the capacitance C of the capacitor that can be formed is As expressed by , any capacitance C can be realized by changing θ, that is, by setting the electrical length l of the transmission path.

FIG. 16 is a diagram illustrating the operation mode of the transmission line explained in Equations 9 to 13 above. FIG. 16 shows how the equivalent reactance X generated at the terminal changes in accordance with the change in the electrical length l of the transmission line with its tip in an open state. As is clear from Fig. 16, the electrical length l of the transmission line is λ/4 or less or λ/2 ~
In cases such as at 3λ/4, it is possible to form a negative terminal reactance, ie equivalently to form a capacitor.
Further, by arbitrarily setting the electrical length l of the transmission path under conditions that generate negative terminal reactance, it is possible to realize the capacitance C to an arbitrary value.

The capacitor C thus formed is a lumped constant capacitor 279 shown in FIG. 14d.
can be equivalently replaced as . and,
The inductor formed by the sum of the distributed inductor component existing in the transmission line and the lumped inductor component generated by bending the transmission line can be equivalently replaced as the lumped constant inductor 280. If the ground terminal is shown in common in FIG. 14d, it is clear that the final result is a lumped constant capacitor 279 and a lumped constant inductor 280, as shown in FIG. 14e.
It is equivalent to a parallel resonant circuit consisting of the following, and can realize a tuner.

Although the configuration and operation described above realize the tuner of the present invention, the configuration of the tuner of the present invention and its operating principle are completely different from those of conventional tuners.
Therefore, in order to prove that the tuner according to the present invention is completely different from conventional tuners or those having other configurations even if they use the same transmission line as the tuner of the present invention. , the construction and operation of a conventional tuner or tuner with other transmission line configurations will now be described and contrasted. This clarifies the difference from the tuner according to the present invention and also clarifies the novelty of the tuner according to the present invention.

FIG. 17 shows the operation when the transmission line electrodes are made of the same material as used in the tuner of the present invention, but the difference is that the ground terminals are set in the same direction. be. In FIG. 17a, transmission line electrodes 281 and 282
It is assumed that the transmission path of the tip oven consisting of the following is driven by a signal source 283 that generates voltage e. As a result, a standing wave voltage e _A is excited at the open terminal at the tip of the transmission line electrode 281, and a standing wave voltage e _B is excited at the open terminal at the tip of the transmission line electrode 282, which is installed opposite or in parallel. shall be induced. Here, since the ground terminals of the transmission line electrodes 281 and 282 are set in the same direction, the standing wave voltages e _A and e _B are in phase with each other. Therefore, the respective voltage distribution coefficients at transmission line electrodes 281 and 282 have the same K. As a result, the potential difference V at any part where the transmission line electrodes face each other becomes V=Ke _A -Ke _B (14). Here, if the electrical lengths of the respective transmission line electrodes 281 and 282 are the same length, e _A = e _B ... (15), so the potential difference V in Equation 14 is V = Ke _A - Ke _A = O...(16). In other words, no potential difference occurs in any part of the transmission path. FIG. 17b shows a configuration in which the signal source 283 in FIG. 17a is replaced at the end of the transmission line, and is equivalent to installing an unbalanced signal source 284 that generates voltage e'. In this equivalent circuit, there are only parallel transmission lines with no potential difference between them. In other words, it is clear that this is equivalent to the case where only one transmission line electrode 285 exists, as shown in FIG. 17c. Then, by equivalently replacing the signal source 283 and the ground terminal with the original circuit as shown in FIG. 17a, the result is as shown in FIG. 17d. In other words, only an equivalent lumped constant inductor 286 is formed, which is composed of a distributed inductor component of the transmission line and a lumped inductor component generated by the curved shape of the transmission line. As is clear from the above, since a capacitor cannot be formed in parallel with an inductor, the intended parallel resonant circuit tuner cannot be realized.

Figure 18 shows a general microstrip line formed with the same transmission line electrode as in the tuner of the present invention as one side of the transmission line electrode, but the electrode facing the transmission line electrode has a sufficiently wide ground. This shows the operation when the curved points are different. In FIG. 18a, a transmission line electrode 287 is placed opposite a sufficiently wide ground electrode 288, and is driven by a signal source 289 that generates a voltage e, and a standing wave voltage e _A is excited at the oven terminal at the tip of the transmission line. and its voltage distribution coefficient is K. On the other hand, assuming that a standing wave voltage e _B having a voltage distribution coefficient K is virtually generated at the ground electrode 288, the transmission line electrode 287 and the ground electrode 28
The potential difference V at any part where 8 faces each other is expressed as: V=Ke _A −Ke _B (17). However, the standing wave voltage e _B at the ground electrode 288 is uniformly at the ground potential (zero potential), and e _B =O (18). Therefore, there is no voltage distribution coefficient at ground electrode 288 either. As a result, the potential difference V becomes V=Ke _A (19). This makes it possible to form a distributed capacitor between the transmission line electrode 287 and the ground electrode 288. However, the transmission line electrode 28
7 closely faces the ground electrode 288, the two ends of the transmission line electrode 287 are almost in a shot state due to mutual induction. Therefore, the Q performance of the inductor component in the transmission line electrode 287 is significantly degraded. That is, this microstrip line has an equivalent loss resistance 29 as shown in FIG. 18b.
A parallel resonant circuit is formed of a lumped constant inductor 291 and a lumped constant capacitor 292, each including zero. Here, the equivalent loss resistance 290 actually has a considerably large resistance value, so
Losses in the resonant circuit become very large. Therefore, it is obvious that only a tuner with very low Q performance can be realized, and is not suitable for practical use.

FIG. 19 shows the circuit configuration of a λ/4 resonator, which is most commonly used in the past. This shows that the tuner is completely different from the tuner of Second
In FIG. 0, the balanced mode transmission line electrodes 293 and 294 have an electrical length l set equal to λ/4 at the resonance frequency, and have shortened tips. and a balanced signal source 2 that generates a voltage e
95, each transmission line electrode is driven in a balanced mode. The ground terminal is set at the neutral point of the balanced signal source 295, and does not specifically ground any terminal in the transmission line electrode. In this case, the equivalent terminal reactance X generated at the terminal of the transmission line is as follows, where Z _O is the characteristic impedance of the transmission line: X=Z _O tanθ (20). Here, the characteristic impedance Z _O is the same as that shown in the 8th equation, and θ is also the same as that shown in the 10th equation. In this resonator, the electrical length l of the transmission path is l=λ/4...(21), so θ=π/2...(22). Therefore, _the terminal reactance X in Equation 20 is as follows: However, when comparing the configuration of this λ/4 resonator with the configuration of the tuner of the present invention, first of all, regarding the terminal conditions of the transmission line, the tuner of the present invention is in an open state, whereas the conventional λ/4 resonator has an open state. It is clear that the /4 resonator is in a short state and therefore has a completely different configuration in terms of terminal conditions. Furthermore, regarding the setting of the electrical length l of the transmission line, in the tuner of the present invention, it is set to λ/4 or less of the tuning frequency, and in practice it is set to a very short value of about λ/16. However, the conventional λ/4
In the resonator, the resonant frequency is strictly set to λ/4, and it is therefore clear that the configuration is fundamentally different in setting the electrical length l of the transmission path. Also, the electrical length l of the transmission path in the configuration
Due to the difference in λ/
In the case of four resonators, it is necessary to provide a very long transmission path, which has the disadvantage of increasing the size. Conventional λ/
4.In order to downsize the resonator, there are some cases in which the length of the transmission path is shortened by interposing a dielectric material with a very high permittivity, but the dielectric material with a high permittivity used for this purpose generally has a dielectric loss tanδ. It is very large, and therefore has the disadvantage that the Q performance as a resonator is significantly reduced. Furthermore, the temperature dependence of the dielectric constant of a dielectric material having a high dielectric constant is generally large, and therefore there is also the disadvantage that it is difficult to ensure the stability of the resonant frequency.

Next, in order to clarify the superiority of the performance of the tuner of the present invention, experimental results will be shown and explained in comparison with the performance of a conventional tuner. FIG. 20 is a graph showing the experimental results of measuring the temperature dependence of the tuning frequency. FIG. 21 is a graph showing the experimental results of measuring the temperature dependence characteristics of resonance Q. In FIGS. 20 and 21, characteristic (A) is the temperature dependence of the tuner according to the present invention, and is an experimental result when an alumina ceramic material or a resin printed circuit board is used as the dielectric. On the other hand, characteristic (B) is as shown in Figure 2.
This is the temperature dependence characteristic of the tuner that has been most commonly used in the past. From these experimental results, it is clear that in the tuner of the present invention, the tuned frequency is extremely stable even if it is constructed using a general dielectric material, and furthermore, the resonance Q is high and the frequency is high. . On the other hand, in conventional tuners, the tuning frequency and Resonance Q
It was difficult to ensure stability. Thereby, other temperature compensation components or other self-stabilizing compensation circuits were added to compensate for the instability.

The operation of the tuner used in the oscillation device of this embodiment configured as described above will be described below.
FIG. 14 shows an operational equivalent circuit of the tuner. 22nd
In FIG. a, the grounding of electrodes 124 and 125, which are placed opposite to each other with a dielectric (not shown) in between, is set in opposite directions, and the open terminal 126 of the electrode 124 is connected to the voltage variable capacitance element 12.
7 are connected to form a basic circuit.
If an AC signal is applied to the open terminal 126 at this point, the ground terminals of the electrodes 124 and 125 are set in opposite directions, so each electrode 1
The alternating currents driven into electrodes 124 and 125 are in antiphase with each other, thereby causing electrodes 124 and 125 to
A distributed capacitance can be generated between the two. This situation is shown in FIG. 22b, where the distributed capacitor 128 is formed and the second
The inductive component of the electrode 125 shown in FIG. 2a is canceled and becomes equivalent to the ground plane 129. Electrode 1
A distributed inductance is present at 24, forming a distributed inductor 130 and a distributed capacitor 128 as shown in FIG. 22c, forming a distributed constant circuit. If this is equivalently converted into a lumped constant circuit, a parallel resonant circuit of the inductor 131, capacitor 132, and voltage variable capacitance element 127 will be constructed. By changing the tuning control voltage applied to the control terminal 133 of the voltage variable capacitance element 127, the tuning frequency of this tuner can be variably controlled.

The operation of the oscillation device according to the embodiment of the present invention including the tuner configured as described above will be described below using the oscillation device shown in FIG. 5 as a representative example. Input terminal 3
The tuning control signal in the form of a serial digital signal code supplied to the digital signal processor 2 is converted into a parallel digital signal code by the code converter 31 and input to the digital signal processor 29 . When a latch is used as this digital signal processor 29, the digital signal code input to it is temporarily stored as is and stored in RAM or ROM.
When using the digital signal code, the digital signal code is arbitrarily converted according to the stored contents written in advance. The digital signal code processed by one of them is supplied to the variable frequency divider 25, which sets a frequency division ratio according to the digital signal code.
As a result, the tuner 19, the feedback amplifier 23 and
The oscillation signal is set to a frequency at which the PLL circuit 21 is locked. By arbitrarily setting the serial digital signal code input to the input terminal 32 in this manner, the tuning frequency, that is, the oscillation frequency of the tuner 19 can be variably set in accordance with the code. The oscillation device in the embodiment shown in FIG. 3 supplies a parallel format digital signal code directly to the input terminal 26 of the variable frequency divider 25, and the oscillation device in the embodiment shown in FIG. 29 input terminal 3
0 directly to the parallel format digital signal code. In this way, Figures 3 to 5
The oscillation devices of the embodiments shown in the figures all operate so as to variably set the oscillation frequency using a digital signal code. Here, the tuner 19 in the oscillation device of the embodiment shown in FIGS.
By cutting a required portion of the transmission line electrode 14 (not shown), it is possible to variably set the capacitance of the distributed capacitor generated in the parallel circuit 16 of the inductor and capacitor, and the tuning frequency band can be arbitrarily set. can do.

In the configuration of the tuner of the oscillation device in each of the above embodiments, the voltage variable capacitance element is connected to the open terminal at the tip of the electrode, but the desired purpose can be achieved by connecting it to the terminal at any part of the electrode. Note that metal conductors, printed conductors, or thin film conductors can be used as the electrodes of the tuner in each of the above embodiments, and alumina ceramic, plastic, Teflon, glass, mica, etc. can be used as the dielectric substrate. I can do it.

Effects of the Invention As is clear from the above description, the present invention forms a tuning section using opposing electrodes with a thin dielectric layer interposed therebetween, and also varies the control voltage of the voltage variable reactance element using a digital signal code. Because of this structure, the inductor and capacitor parts of the tuner can be integrated with a simple structure.

An ultra-thin and compact tuner can be realized.

Since the inductor and capacitor of the tuner are connected by a lead race, there is no influence of the lead inductor or the stray capacitor, and therefore the operation of the oscillator becomes extremely stable and the tuning accuracy is improved.

It is possible to reduce the number of parts in the tuner, which enables rationalization of manufacturing and cost reduction. Furthermore, by using a tuning control method using a digital signal code and a frequency locking function of a stable PLL circuit, it is possible to obtain a very stable tuning control voltage, and the tuning accuracy of the oscillator can be significantly improved.

It is possible to connect to a computer-applied multifunctional digital control system, and it is possible to realize advanced multifunctional control of the oscillation device and equipment using it.

This excellent effect can be obtained.

[Brief explanation of the drawing]

Figure 1 is a circuit diagram of a conventional oscillation device, Figure 2 is a perspective view of the configuration of a tuner used in the conventional oscillation device, and Figure 3 is a circuit diagram of a conventional oscillation device.
5 to 5 are configuration circuit diagrams of an oscillation device in an embodiment of the present invention, and FIGS. 6 to 13 are configuration diagrams of a tuner used in an oscillation device in an embodiment of the present invention, and in each, a is a surface view. , b is a side view, c is a back view, Figures 14 a to e, Figure 15 a,
b, FIG. 16 is an explanatory diagram showing the operating principle of a tuner, FIGS. 17 a to d, FIGS. 18 a to b, and FIG. 19 are explanatory diagrams showing the operating principle of a conventional tuner,
20 and 21 are characteristic diagrams of the tuning frequency and resonance Q with respect to temperature changes of the present invention and the conventional tuner, and FIG. 22 is a diagram illustrating the operating principle of the tuner used in the oscillation device in the embodiment of the present invention. be. 14, 15, 101, 102, 104, 10
5,107,108,110,111,113,
114, 116, 117, 119, 120, 12
2,123,124,125...electrode, 18,1
27... Voltage variable capacitance element, 21...
PLL circuit, 23... Feedback amplifier, 22... Low pass filter, 24... Fixed frequency divider, 25... Variable frequency divider, 28... Phase comparator, 27... Crystal oscillator, 29... Digital Signal processor, 31...
code converter.

Claims (1)

[Claims] 1. Terminals connected to the ground of electrodes that are arranged opposite to each other via a dielectric or arranged side by side on the surface of a dielectric are set so that they are not opposed to each other but have different opposing positions. , an oscillator including a tuner with a variable voltage reactance element connected to the open terminal of any one of the above electrodes is installed, and a PLL circuit including a variable frequency divider is installed, and the tuning control output of the PLL circuit is A voltage is supplied to the voltage variable reactance element, an oscillation output of the oscillator is supplied to the variable frequency divider, and the division ratio of the variable frequency divider is arbitrarily controlled by a frequency setting code. An oscillation device that 2. The oscillation device according to claim 1, wherein a latch is installed at the code setting terminal of the variable frequency divider as a control section. 3 Connect to the code setting terminal of the variable frequency divider as a control section.
An oscillation device according to claim 1, which is equipped with RAM or ROM. 4. The oscillation device according to any one of claims 1 to 3, further comprising a code converter for converting a serial input code into a parallel output code as a control section. 5. Claim 1 in which an electrode having at least one bending part exhibiting an arbitrary bending angle or bending rate and an arbitrary bending direction is used.
The oscillation device according to any one of Items 1 to 4. 6. The oscillation device according to any one of claims 1 to 4, using an electrode having a spiral shape. 7. According to any one of claims 1 to 6, the length of one electrode is arbitrarily set shorter than the length of the other electrode, and the electrodes are arranged facing each other or in parallel at an arbitrary part. oscillator. 8. The oscillation device according to any one of claims 1 to 7, wherein each electrode or a portion or all of one of the electrodes is installed inside the dielectric. 9. The oscillation device according to any one of claims 1 to 8, wherein the terminals connected to the ground in each of the electrodes are not connected to the ground but are used as a common terminal.