JPH0537243A - Orthogonal modulator, orthogonal demodulator and surface elastic acoustic wave device - Google Patents

Abstract

PURPOSE:To offer the orthogonal modulator and the orthogonal demodulator of small size with high accuracy. CONSTITUTION:A distribution means distributing an original local signal from a local oscillation circuit 3 into two and a 90 deg. phase shift circuit giving a phase difference of 90 deg. to the original local signal distributed into two to obtain 1st and 2nd local signals 5,6 are formed by using two surface elastic wave (SAW) elements 4 each comprising input output electrodes 11-13 and a piezoelectric substrate 14 whose phase shift through the SAW propagation path differs from 90 deg. in the vicinity of the local signal frequency.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a quadrature modulator / quadrature demodulator and a surface acoustic wave device used therein.

[0002]

2. Description of the Related Art With the recent rapid increase in demand for mobile radio communication, in place of the conventional mainstream analog modulation system,
Digital modulation schemes that enable effective use of radio waves are being actively studied. Typical digital modulation methods include multi-level phase modulation (BPSK, QP
SK, etc.), frequency shift keying (FSK), amplitude phase modulation method (QAM, etc.), and the like.

A quadrature modulator is one that specifically realizes these modulations. FIG. 15 shows a block diagram of a conventional quadrature modulator. This quadrature modulator multiplies two series of baseband signals 101 and 102 which are modulation inputs by a multiplication circuit 10
7 and 108, the two local signals 105 and 106 in the orthogonal relationship generated from the output of the local oscillation circuit 103 by the two-way distribution means and the 90 ° phase shift circuit 104 are amplitude-modulated, and then added by the adder 109. Modulation output 1
10 is generated.

The quadrature modulator has an input baseband signal 1
It is also possible to realize it by performing digital signal processing on 01 and 102, but for that purpose, high-speed A / D
Since a converter is required, power consumption increases, the circuit becomes large, and frequency conversion up to the transmission frequency becomes difficult. For these reasons, it is desired to realize the quadrature modulator with an analog circuit in a communication device such as a mobile radio communication terminal that requires low power consumption and miniaturization. In that case, the problem is the accuracy of the analog circuit, and especially the accuracy of the multiplication circuit and the 90 ° phase shift circuit.

As the multiplication circuit, a circuit form of a type in which a local signal component does not leak to the output is required, and a balanced double modulation circuit using a Gilbert cell or an analog switch is usually used. These circuits require a differential signal as a local signal. However, when a single-phase local signal is converted into a differential signal inside the integrated circuit, in-phase components cannot be removed in the conversion process, The phase difference may vary between the input and output. Since these errors directly appear as errors in the output of the quadrature modulator, there is a very difficult problem in terms of circuit configuration. Further, the relative accuracy of the gain and the phase rotation amount of the two multiplication circuits does not matter so much when both multiplication circuits are formed on the same substrate, but the amplitude accuracy of the modulation input signal and the local signal is less than that of the quadrature modulator. Since it affects the accuracy, the relative accuracy of the amplitude of the local signal becomes important.

Next, for the 90 ° phase shift circuit, a circuit having high orthogonality, that is, high accuracy of amplitude and phase and low spurious is required. There are roughly three types of configuration methods for implementing such a phase shift circuit on an integrated circuit. The first is twice or four of the local signal
This is a method of generating a double harmonic and dividing the frequency to obtain a phase shift amount of 90 °. This method has a drawback that the operating frequency is higher than that of the local signal, which leads to an increase in current consumption and makes it difficult to operate at high frequency. Second,
This is a method for realizing a 90 ° phase shift circuit by combining CR phase shift circuits. In this method, the absolute accuracy of C and R becomes a problem, so there is a large problem in terms of accuracy and cost.
The third is a method of controlling the phase and the amplitude by a feedback loop or an open loop using two oscillation circuits or an oscillation circuit and a variable phase shift circuit. This method has a problem that the circuit is complicated because the control system is required for two types of amplitude and phase, and the operation becomes unstable.

When constructing a quadrature modulator, it is also important to minimize harmonic components in the local signal input. Normally, the frequency of the local signal is several 100M
Since the frequency is on the order of Hz, fundamental wave oscillation is not possible when a crystal oscillator is used for the oscillation circuit, and it is necessary to use an overtone transmission circuit or a multiplication circuit, or to configure a PLL using a VCO. is there. In either case, since the output contains harmonic components, a filter is required to remove spurious. Further, when the local signal is supplied from the outside, the frequency is high, and therefore impedance matching with an external circuit is required.

On the other hand, in the receiving system, it is possible to perform quadrature demodulation by the reverse operation of quadrature modulation. FIG.
FIG. 4 is a block diagram of a conventional quadrature demodulator. The input high frequency signal 111 is supplied to the multiplication circuit 107,
And a 90 ° phase shift circuit 104.
The baseband signals 101 and 102 having orthogonal components are output through the low-pass filters 112 and 113 by being demodulated by being multiplied by the two local signals having a phase difference of 90 ° generated by the above. Even in such a quadrature demodulator, the quadrature of the 90 ° phase shift circuit 104 directly affects the demodulation accuracy, and thus has the same problem as the quadrature modulator.

Considering the use of the quadrature modulation method in a mobile radio terminal, miniaturization is very important. However, the above-mentioned conventional quadrature modulator and quadrature demodulator must satisfy this requirement. Is difficult. For example, as described above, the local oscillator circuit output requires a filter for removing unnecessary components, and the modulator / demodulator input / output section also requires a filter. Of these, the low-frequency filters used for the input part of the modulator and the output part of the demodulator are
Although it can be integrated by the current active filter technology, the high-frequency filter used for removing an unnecessary component of the output part of the modulator or the input part of the demodulator or the output of the local oscillation circuit is difficult to integrate. It is inevitably an external filter, which is a factor that hinders the miniaturization of the device.

[0010]

As described above, in the prior art, when it is attempted to realize the quadrature modulator and the quadrature demodulator by analog circuits, two local signals are 90 ° apart.
As a 90 ° phase shift circuit for providing the phase difference between the two, it is difficult to realize a circuit in which the outputs of two local signals have the same amplitude and high accuracy and are capable of differential output. It has been difficult to implement modulators and quadrature demodulators.

An object of the present invention is to solve the above problems and provide a small-sized and highly accurate quadrature modulator and quadrature demodulator.

Another object of the present invention is to provide a surface acoustic wave device suitable for these quadrature modulator and quadrature demodulator.

[0013]

In order to solve the above-mentioned problems, the quadrature modulator of the present invention has a dividing means for dividing an original local signal from a local oscillator circuit into two parts, and 90 ° for the divided original local signal. A 90 ° phase shift circuit that obtains the first and second local signals by applying the phase difference of 2 is obtained by two surface acoustic wave delay elements in which the amount of phase shift of the surface acoustic wave propagation path differs by 90 ° in the vicinity of the local signal frequency. It is characterized by being configured.

The quadrature demodulator of the present invention is configured to multiply the input signals to be demodulated by the first and second multiplication circuits by the first and second local signals having a phase difference of 90 ° to obtain a demodulated output. , A dividing means for dividing the original local signal from the local oscillator circuit into two and a 90 ° phase shift circuit for giving a phase difference of 90 ° to the divided original local signal to obtain the first and second local signals. The surface acoustic wave propagation path is composed of two surface acoustic wave delay elements which differ in the amount of phase shift by 90 ° in the vicinity of the local signal frequency.

Further, another quadrature demodulator of the present invention uses first and second multiplication circuits for dividing the input signal to be demodulated into two and giving the first and second signals having a phase difference of 90 °. In a structure for obtaining a demodulated output by multiplying each of the local signals, a dividing means for dividing the input signal into two and a first and a second for giving a phase difference of 90 ° to the divided original local signals.
The 90 ° phase shift circuit for obtaining the local signal is constituted by two surface acoustic wave delay elements in which the amount of phase shift of the surface acoustic wave propagation path differs by 90 ° in the vicinity of the frequency of the input signal.

In the surface acoustic wave device of the present invention, in the quadrature modulator or the quadrature demodulator as described above, at least a part of the 90 ° phase shift circuit and the local oscillator circuit are constituted by surface acoustic wave elements on the same substrate. It is characterized by

Further, another surface acoustic wave device of the present invention is
The amount of phase shift in the surface acoustic wave propagation path is 90 ° at the operating frequency.
It is characterized by having a 90 ° phase shift circuit composed of two surface acoustic wave delay elements which are different from each other and whose center frequency determined by the electrode period is different from the used frequency.

[0018]

When the distribution means and the 90 ° phase shift circuit are composed of two surface acoustic wave delay elements whose surface acoustic wave propagation paths are different in phase shift amount by 90 ° at a specific frequency, the phase and amplitude are close to the frequency. It is possible to realize a highly accurate 90 ° phase shift circuit with less error and less harmonic spurious without adjustment.
If this 90 ° phase shift circuit is used in a local signal system of a quadrature modulator or a quadrature demodulator, a differential signal can be obtained without an external circuit. When this 90 ° phase shift circuit is used for the input signal system of the quadrature demodulator, unnecessary waves can be removed.

If the surface acoustic wave device is constructed by forming this 90 ° phase shift circuit together with other surface acoustic wave elements constituting a local oscillation circuit, for example, a device such as a mobile radio communication terminal can be made compact. Can be converted. Further, by mounting the 90 ° phase shift circuit or the surface acoustic wave device including the same on the same package as the integrated circuit, the number of pins of the package is reduced and further miniaturization is achieved.

Further, by making the center frequency determined by the electrode period of the 90 ° phase shift circuit different from the working frequency, that is, the local signal frequency or the input signal frequency, the multiple reflections generated between the input and output electrodes of the surface acoustic wave element can be prevented. The impact is reduced.

Further, the 90 ° phase shift circuit according to the present invention can be used also as a filter in a conventional modulator / demodulator, and the size of the device is hardly changed.

[0022]

Embodiments of the present invention will be described below with reference to the drawings. 1 is a diagram showing a configuration of a quadrature modulator according to a first embodiment of the present invention. This quadrature modulator is a surface acoustic wave (hereinafter referred to as SAW) forming a 90 ° phase shift circuit with a local oscillator circuit 3, local signal distribution means.
Device 4), multiplication circuits 7 and 8 and an adder 9.

The SAW device 4 includes a pair of input electrodes 11
And the output electrodes 12 and 13 facing each other at a predetermined distance from each other are formed on the piezoelectric substrate 14, and the SA is provided between the input electrode 11 and the output electrodes 12 and 13, respectively.
A W delay element is constructed. The input electrode 11 and the output electrodes 12 and 13 are formed of interdigital electrodes in the illustrated example. Here, the phase shift amount of the SAW propagation path between the input electrode 11 and the one output electrode 12 and the phase shift amount of the SAW propagation path from the input electrode 11 to the other output electrode 13 are
90 at the used frequency (eg local signal frequency)
°, more generally (2n + 1/2) π (n is a natural number)
Only different.

The operation of the quadrature modulator shown in FIG. 1 will be described. The original local signal output from the local oscillator circuit 3 is the input electrode 1
1, and the first and second local signals having a phase difference of 90 ° from the output electrodes 12 and 13 are respectively taken out as differential signals. The first and second local signals are multiplication circuits (mixers) used as amplitude modulation circuits
7 and 8 and are multiplied by the first and second baseband signals 1 and 2 which are modulation inputs. That is, the baseband signals 1 and 2 amplitude-modulate the first and second local signals. The outputs of the multiplication circuits 7 and 8 are added by the addition circuit 9 to generate a desired quadrature modulation output 10.

An advantage of using the SAW device in the 90 ° phase shift circuit is that the phase error is very small. Currently, the piezoelectric substrate materials used in SAW devices are quartz, LiNbO ₃ , and Li.
Such as TaO ₃ . Assuming that 128 ° Y-XLiNbO
_{If 3} is used, the sound velocity will be 3,960 m / s. Therefore, if the center frequency is 400 Hz,
The wavelength is about 10 μm. On the other hand, considering that the relative accuracy of the position between the output electrodes 12 and 13 is the accuracy of the position of the center of the output electrodes 12 and 13 and that the electrodes 11 to 13 are formed with the same mask, It is fully possible to obtain an accuracy of about ± 0.01 μm, estimated from the process technology. When this position accuracy is converted into phase accuracy, it becomes ± 0.36 °. Furthermore, the phase error due to temperature is ± 30 ° C in the operating temperature range and -80 in the temperature coefficient.
When it is ppm / ° C, it becomes ± 0.86. When these values are combined, it is possible to realize a 90 ° phase shift circuit with an accuracy of ± 2 ° as a whole.

Further, by taking out the output of the 90 ° phase circuit as a differential signal, there is a feature that the constitution of the multiplication circuit in the subsequent stage becomes easy. It is desirable to use a double-balanced modulation circuit without carrier leakage as a multiplication circuit in the subsequent stage, but a differential input is required to fully operate this double-balanced modulation circuit. For this reason, in the past, processing such as differential conversion was performed on single-phase input, and it was very difficult to ensure accuracy due to problems such as phase error, amplitude error, and distortion that occur in the process. In the invention, a differential signal can be obtained by simply extracting the signal as it is from the output electrode of the SAW device, and such a problem is solved.

Further, as a problem of the high frequency circuit, there is impedance matching between the 90 ° phase shift circuit and the multiplication circuit. However, by appropriately selecting the number of pairs of input and output electrodes of the SAW device and the electrode width, an external circuit is not used. It is possible to achieve impedance matching with both. In addition, since the delay element formed by the SAW device operates as a bandpass filter in principle, it is effective in removing spurious signals from local signals.

The 90 ° phase shift circuit constructed by using the SAW device in this way has (1) a small phase error and (2)
Small amplitude error, (3) Easy to obtain differential signal,
(4) There are many advantages that impedance matching is easy and (5) spurious can be removed.

The construction method of the 90 ° phase shift circuit using the SAW device is not limited to that shown in FIG. 1, and various modifications can be considered. For example, as shown in FIG. 2, the output electrodes 12 and 13 are arranged on both sides of one input electrode 11, and the center distance between the input electrode 11 and the output electrodes 12 and 13 is λ / 4.
By shifting (λ is the wavelength of the SAW), it is possible to take out the outputs having the same amplitude but different phases by 90 ° near the center frequency from the output electrodes 12 and 13.

Further, as shown in FIG. 3, the output electrodes 12 and 13 are arranged so as to face each other on the same side of the input electrode 11, and the center distance between the input electrode 11 and the output electrodes 12 and 13 is λ / 4.
You may shift it.

Further, as shown in FIG. 4, the output electrodes 12 and 13 are sequentially arranged on one side of the input electrode 11, and the distance between the centers of the output electrodes 12 and 13 is (N + 1/4) λ (N is an integer). You may just shift it. In addition, a method to reduce the interference between input and output electrodes by using a multi-strip line,
It is also possible to appropriately use a method conventionally applied to SAW devices, such as a method using a unidirectional conversion circuit for the input / output electrodes.

FIG. 5 is a diagram showing the structure of a quadrature demodulator according to the second embodiment of the present invention. This quadrature demodulator is composed of a local oscillator circuit 3, a SAW device 4 forming a 90 ° phase shift circuit with a local signal distributing means, multiplication circuits 7 and 8 and low pass filters 16 and 17.

The high frequency signal 15 to be demodulated is divided into two and inputted to the multiplying circuits 7 and 8, and is multiplied by the first and second local signals from the SAW device 4 which are equal in amplitude but differ in phase by 90 °. By removing the unnecessary high frequency components from the outputs of the multiplication circuits 7 and 8 through the low pass filters 16 and 17, two series of baseband signals 1 and 2 are obtained as demodulation outputs. Also in this case, the same advantages as those of the first embodiment can be obtained by forming the 90 ° phase shift circuit with the SAW device 4. The structure of the SAW device 4 is in the form shown in FIG. 2 in this example, but may be the one shown in FIGS. 1, 3, 4 and the like.

FIG. 6 is a block diagram of a quadrature demodulator according to the third embodiment of the present invention. Inductors 18 and 19 are connected in parallel to the output electrodes 12 and 13 for outputting the local signal of the SAW device 4, respectively. The connection point is different from FIG. SA
W devices generally have a capacitive component at the center frequency, and it is desirable to add an inductive component to cancel this component. However, in general, the mismatch due to the capacitive component is negligible even if the inductor is not attached practically, so that the inductor is often omitted for downsizing as shown in FIG.

However, in the case of quadrature demodulation, since the demodulation output appears in the base band, low frequency noise in the multiplication circuits 7 and 8 may be a problem. In particular, when the local signal input portions of the multiplication circuits 7 and 8 are open in terms of direct current, low frequency noise of the bias circuit or the like appears directly in the output.
Therefore, it becomes difficult to realize a demodulator with good NF. In this respect, when the inductors 18 and 19 are inserted in parallel to the local signal output section of the SAW device 4, that is, the local signal input sections of the multiplication circuits 7 and 8 as in the present embodiment, the local signal input section is low frequency. Because it can be regarded as a short circuit,
The low frequency noise of the bias circuit can be ignored, and it becomes possible to realize a quadrature demodulator with good NF.

FIG. 7 is a diagram showing the structure of a quadrature demodulator according to the fourth embodiment of the present invention. In this quadrature demodulator, the high-frequency signal 15 to be demodulated is made into first and second signals having the same amplitude and different phases by 90 ° by the SAW device 4 forming a 90 ° phase shift circuit with the distributing means. These first and second signals are input to the multiplying circuits 7 and 8 and are multiplied by the local signals of the same phase from the local oscillating circuit 3 and then passed through the low pass filters 16 and 17 to generate the two series of baseband signals 1. 2 is taken out as a demodulation output. In this case, since the 90 ° phase shift circuit in the SAW device 4 also has a characteristic as a bandpass filter,
There is also an advantage that spurious of the input high frequency signal can be removed.

FIG. 8 is a diagram showing a fifth embodiment of the present invention, in which the local oscillation circuit 3 is composed of a SAW device and is formed on the same substrate as the 90 ° phase shift circuit. That is, in this case, the input electrode 11 in the SAW device 4
Is a component of the local oscillation circuit 3, and is formed on the same piezoelectric substrate 14 together with the output electrodes 12 and 13 which are components of the 90 ° phase shift circuit, and is elastically coupled. According to this embodiment, there is an advantage that the entire size can be further reduced.

Further, the same effect can be obtained by integrally forming the SAW device 4 with a SAW device used in other parts and using a common substrate. The sixth embodiment of the present invention shown in FIG. 9 is such an example.
SAW used elsewhere on substrate 14 of device 4
The electrodes 20 of the device are formed.

FIG. 10 is a diagram showing a seventh embodiment of the present invention, in which the SAW device 4 comprises a local oscillating circuit composed of a vector synthesizing type oscillating circuit using two SAW delay elements, a distributing means and 90. ° The phase shift circuit is integrated to further reduce the size. Also in this case, 9
The phase shift amount of the two SAW propagation paths in the 0 ° phase shift circuit is
They differ by 90 ° near the center frequency. Further, in this embodiment, the variable phase shifter 22 is realized by vector-synthesizing the output of the 90 ° phase shift circuit, and the variable frequency oscillator is configured by combining with the amplifier 21.

FIG. 11 is a diagram showing the configuration of the eighth embodiment of the present invention, in which the SAW device 4 described above and the integrated circuit 23 including the multiplication circuit and other circuits are mounted in the same package 24. It was done. In the present invention, SAW
Since the output is taken out as a differential signal from the 90 ° phase shift circuit in the device 4, five wires including an input terminal are required at least between the SAW device 4 and the integrated circuit 23. If the SAW device 4 and the integrated circuit 23 are composed of different chips, the number of pins of each chip increases, and miniaturization becomes difficult.

On the other hand, when the SAW device 4 and the integrated circuit 23 are integrated as in this embodiment, the wiring between the two can be performed by the wiring inside the package 24, so that the package is different from the case where it is mounted in another package. The total number of pins in is reduced by at least 10. Further, since the SAW device is usually larger than the integrated circuit chip as shown in FIG. 11, it is possible to increase the number of pins without increasing the package size, which is also advantageous for downsizing.

12 to 13 are characteristic diagrams for explaining the ninth embodiment of the present invention. In a low-loss SAW device, ripples occur in the characteristics due to the effect of multiple reflection occurring between the input / output electrodes near the center frequency determined by the period of the electrode. This is generally called TTE (Tripple Tra)
nsite Echo).

In the case of the present invention, as shown in FIG. 12, since the distance between the input and output of the two SAW delay elements is different, the TTE
Also occurs at different frequencies, resulting in amplitude and phase errors. Figure 1 shows the amplitude and phase differences between the outputs.
3, shown in FIG. Therefore, the center frequency of the SAW device determined by this electrode period and the frequency actually used are shifted so that the influence of TTE is almost zero.
By designing for use as a 0 ° phase shift circuit, it is possible to eliminate the error due to TTE. In short, the phase shift amount of the SAW propagation path is 90 at the used frequency.
A 90 ° phase shift circuit may be configured by two SAW delay elements that are different from each other and have a center frequency determined by the electrode period different from the used frequency.

In the above embodiments, the local signal frequency is fixed for the sake of simplicity, but there are some fluctuations in practice. The configuration of the present invention has an intermediate frequency of
There is no problem when applied to a modulator and demodulator in the 00 MHz band and used at a single frequency, but when performing direct modulation or demodulation at the transmission frequency, the error of the 90 ° phase shift circuit caused by fluctuations in the local signal frequency Becomes a problem. If the frequency fluctuates by 1%, the phase error of the 90 ° phase shift circuit due to the fluctuation is about 3.6 °. Therefore, if the ratio between the center frequency and its band is approximately 1% or less, 90 ° ± 3 ° including the error in the first embodiment described above.
It is possible to suppress the error within. If you need a wider band than this, you can switch the delay element depending on the frequency, or apply a chirp filter, etc.
A delay element having a different propagation time depending on the frequency may be used.

[0045]

According to the present invention, in the quadrature modulator and the quadrature demodulator, the local signal or input signal distribution means and the 90 ° phase shift circuit are composed of SAW delay elements, so that the phase and amplitude can be adjusted without adjustment. It is possible to perform highly accurate modulation and demodulation with few errors. Furthermore, due to the characteristics of SAW devices, filters, impedance matching, single-phase-
Since a differential conversion function is obtained, a circuit for these processes becomes unnecessary, and miniaturization of the quadrature modulator and the quadrature demodulator can be achieved.

[Brief description of drawings]

FIG. 1 is a configuration diagram of a quadrature modulator according to a first embodiment of the present invention.

FIG. 2 is a plan view showing another example of the SAW device used in the present invention.

FIG. 3 is a plan view showing another example of the SAW device used in the present invention.

FIG. 4 is a plan view showing another example of the SAW device used in the present invention.

FIG. 5 is a configuration diagram of a quadrature demodulator according to a second embodiment of the present invention.

FIG. 6 is a block diagram of a quadrature demodulator according to a third embodiment of the present invention.

FIG. 7 is a configuration diagram of a quadrature demodulator according to a fourth embodiment of the present invention.

FIG. 8 is a configuration diagram of a main part according to a fifth embodiment of the present invention.

FIG. 9 is a configuration diagram of a main part according to a sixth embodiment of the present invention.

FIG. 10 is a configuration diagram of a main part according to a seventh embodiment of the present invention.

FIG. 11 is a configuration diagram of a main part according to an eighth embodiment of the present invention.

FIG. 12 is a characteristic diagram for explaining a ninth embodiment of the present invention.

FIG. 13 is a characteristic diagram for explaining a ninth embodiment of the present invention.

FIG. 14 is a characteristic diagram for explaining a ninth embodiment of the present invention.

FIG. 15 is a block diagram of a conventional quadrature modulator.

FIG. 16 is a block diagram of a conventional quadrature demodulator.

FIG. 17 is a block diagram of a conventional quadrature demodulator.

[Explanation of symbols]

1 ... 1st baseband signal 2 ... 2nd baseband signal 3 ... Local oscillation circuit 4 ... SAW device 5 ... 1st local signal 6 ... 2nd local signal 7 ... 1st multiplication circuit 8 ... 2nd Multiplying circuit 9 ... Adder circuit 10 ... Modulation output 11 ... Input electrode 12, 13 ... Output electrode 14 ... Piezoelectric substrate 15 ... High frequency signal 16, 17 ... Low pass filter 18, 19 ... Inductor 20 ... Electrode 21 of other SAW device ... Amplifier 22 ... Variable phase shifter 23 ... Integrated circuit 24 ... Package

Claims (5)

[Claims] 1. A local oscillating circuit, a dividing means for dividing an original local signal from the local oscillating circuit into two, and a phase difference of 90.degree. 90 ° phase shift circuit for obtaining a second local signal, first and second multiplication circuits for multiplying the first and second local signals by an input signal, and outputs of the first and second multiplication circuits And a 90 ° phase shift circuit, wherein the dividing means and the 90 ° phase shift circuit are configured by two surface acoustic wave delay elements whose phase shift amounts in the surface acoustic wave propagation path differ by 90 ° in the vicinity of the local signal frequency. A quadrature modulator. 2. A local oscillating circuit, a dividing means for dividing the original local signal from the local oscillating circuit into two, and a phase difference of 90 ° to the original local signal divided into two by the dividing means. A 90 ° phase shift circuit for obtaining a second local signal, and first and second multiplication circuits for multiplying the input signals to be demodulated by the first and second local signals, respectively, the distribution means and A quadrature demodulator characterized in that a 90 ° phase shift circuit is configured by two surface acoustic wave delay elements in which the amount of phase shift in the surface acoustic wave propagation path differs by 90 ° in the vicinity of the local signal frequency. 3. A dividing means for dividing an input signal to be demodulated into two parts, and a dividing means for dividing the input signal into two parts.
90 ° phase difference is applied to obtain the first and second signals 90
A phase shift circuit, a local oscillator circuit, and first and second multiplication circuits for multiplying the first and second signals by the local signal from the local oscillator circuit, and the distribution means and the 90 ° shift circuit. A quadrature demodulator characterized in that the phase circuit is composed of two surface acoustic wave delay elements in which the phase shift amount of the surface acoustic wave propagation path differs by 90 ° in the vicinity of the frequency of the input signal. 4. The quadrature modulator according to claim 1 or the quadrature demodulator according to claim 2, wherein 90 °
A surface acoustic wave device, characterized in that at least a part of a phase shift circuit and a local oscillator circuit are constituted by surface acoustic wave elements on the same substrate. 5. The quadrature modulator according to claim 1 or the quadrature demodulator according to any one of claims 2 and 3, wherein the phase shift amounts of the surface acoustic wave propagation paths are 90 ° with each other at the used frequency.
A surface acoustic wave device having a 90 ° phase shift circuit composed of two surface acoustic wave delay elements which are different from each other and whose center frequency determined by the electrode period is different from the working frequency.