JPH0526436B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a vector control device for an induction motor.

In recent years, as a control method to improve the quick response of induction motors, the primary current of the motor is controlled by dividing it into an exciting current and a secondary current, and the secondary magnetic velocity and secondary current vector are always orthogonal. A vector control method has been proposed that attempts to obtain equivalent responsiveness. However, when a voltage source inverter such as a pulse width modulation (PWM) inverter is used in the power converter that is actually used, the voltage becomes the manipulated variable even though the primary current is controlled. There was a problem in that the primary current did not flow as per the settings, resulting in poor response and making accurate variable speed control difficult.

The present invention provides vector control that can cancel mutual interference between the secondary magnetic velocity component and the secondary current component in the primary voltage control of the electric motor.
It is an object of the present invention to provide a vector control device that solves conventional problems and can simplify an arithmetic circuit including interference cancellation.

EMBODIMENT OF THE INVENTION Hereinafter, following the principle explanation of the present invention, embodiments will be explained in detail.

First, the voltage equation expressed by the α-β axis that rotates the induction motor in synchronization with the primary voltage is the following equation (1), and the generated torque T is the equation (2).

T=3/2(λ′ ₂ 〓i′ ₂ 〓−λ′ ₂ 〓i′ ₂ 〓)…
...(2) Here, each symbol is the quantity shown below.

e ₁ ; Primary voltage (α, β components) e′ ₂ ; Secondary voltage (α, β components) i ₁ ; Primary current (α, β components) λ′ ₂ : Magnetic velocity (α, β components) r ₁ ; Primary resistance r ₂ ; Secondary resistance M; Excitation inductance L ₂ ; Secondary inductance L〓; Equivalent leakage inductance (L〓=
L ₁ L ₂ −M ² /L ₂ ) L ₁ ; Primary inductance P; Differential symbol d/dt ω _p ; Power supply angular frequency ω _r ; Rotor angular frequency i′ ₂ ; Secondary current (α, β components) Equations (1) and (2) are expressed as the first block diagram.
As shown in the figure, for the two-phase voltages e ₁ 〓, e ₁ 〓, the α-axis and β-axis components of the primary current and secondary magnetic velocity i ₁ 〓, i ₁ 〓,
This is an equivalent block diagram of an induction motor that generates λ' ₂ 〓, λ' ₂ 〓 and torque T.

Here, the α and β axes, which rotate in synchronization with the primary voltage, may be set at any position, but if the α axis is set in the direction of the secondary magnetic velocity, the condition that the secondary current matches the β axis, that is, , the condition that the secondary current is orthogonal to the magnetic velocity is λ′ ₂ 〓=constant λ′ ₂ 〓=0 ……(3), as clarified by vector control theory, and the primary frequency ω _p is ω _p = ω _r + Mr ₂ /L ₂ λ′ ₂ 〓・i ₁ 〓 ……(4) In this way, when α and β axes are determined, the primary current i ₁
The α component i ₁ 〓 (=constant) is a primary current corresponding to the magnetic flux λ′ ₂ , and the β-axis component i ₁ 〓 is a primary current corresponding to the secondary current i′ ₂ .

Next, when the conditions of equations (3) and (4) above are inserted into the block diagram of FIG. 1, the block diagram shown in FIG. 2 is obtained. That is, point a in FIG. 1 is controlled to zero from the relationship of equation (4). Since point c is λ′ ₂ 〓=0, all the quantities connected to this point are zero, and similarly i′ ₂ 〓
The quantity connected from =0 to point d is also zero, and since λ' ₂ = constant, its differential at point b is also zero. _Then , when calculating the part of the broken line block A in FIG. 1 under the same conditions, L ₂ /r ₂ =τ ₂ , L ₌
M ² )/L ₂ ], (L〓+M/L ₂ r ₂・L ₂ /L ₂ P+r ₂・M/L ₂ )ω _p i ₁ 〓=(L
〓＋M ₂ ／L ₂ ／τ ₂ P＋１）ω _p i ₁ 〓 ＝L〓τ ₂ P＋L ₁ L ₂ −M ² ／L ₂ ＋M ₂ ／L ₂ ／τ ₂ P＋１ω _p i ₁ 〓
=L〓/L ₁ τ ₂ P+1/τ ₂ P+1L ₁ ω _p i ₁ 〓 Here, since i ₁ 〓=constant, let P=0 =L ₁ ω ₀ i ₁ 〓 In this way, Fig. The block diagram of
This becomes the block diagram shown in the figure. As is clear from Fig. 2, the secondary magnetic flux λ′ ₂ 〓 cannot be uniquely set by the α-phase primary voltage e ₁ 〓, but is set by +L 〓 by the β-phase primary current i ₁ 〓.
There is interference of ω _p i ₁ 〓, and the secondary current i ₂ 〓 cannot be uniquely set by the β-phase primary voltage e ₁ 〓, but −L ₁ ω ₀ i due to the α-phase primary current i ₁ 〓. There is an interference of ₁ 〓. Therefore,
By correcting the primary voltages e ₁ 〓 and e ₁ 〓 so as to have control amounts that compensate in advance for the interference caused by the primary currents i ₁ 〓 and i ₁ 〓, it is possible to control the magnetic flux and the secondary current without interfering with each other. This correction can be realized by preparing an arithmetic circuit represented by the following equation (5).

e ₁ 〓 e ₁ 〓=r ₁ i ^* ₁ 〓 i ^* ₁ 〓＋ω _p −L〓i ^* ₁ 〓 L ₁ i ^* ₁ 〓 ……(5) In other words, the voltage e ₁ 〓 is set by From the value obtained by multiplying the α-phase primary current setting value i ^* ₁ 〓 by the coefficient of the primary resistance r ₁ to control the secondary magnetic flux λ′ ₂ at a constant level, the interference amount due to the β-phase primary current i ₁ 〓 is calculated as L〓ω _p i ₁ 〓 to control the secondary current i′ ₂ by preparing an arithmetic circuit that subtracts the value obtained by multiplying the β-phase primary current setting value i ^* ₁ 〓 by the power supply angular frequency ω _p and the coefficient L〓, and similarly To set the voltage e ₁ 〓, set value i ^* ₁ 〓 multiplied by coefficient r ₁ to set value i ^* ₁ 〓.
By providing an arithmetic circuit that adds a value obtained by multiplying ω _p and L ₁ by ω p and L 1 , it becomes possible to perform control in which interference is canceled.

The present invention aims to simplify the arithmetic circuit while providing the above-mentioned interference cancellation function. The principle of this simplification of the arithmetic circuit will be explained below.

First, the primary voltages e ₁ 〓, e ₁ 〓 obtained from the calculation circuit based on the above-mentioned equation (5) are taken into the phase voltage calculation circuit together with the trigonometric functions COSω _p t and SINω _p t obtained from the angular frequency ω _p , By performing calculations according to the following equations (6) and (7), three-phase voltage setting values e ^* _a , e ^* b, and e ^* _c of the a, _b , and c phases of the voltage source inverter can be obtained.

e _1d e _1q = COSω _p t−SINω _p t SINω _p t＋COSω _p te ₁ 〓 e ₁ 〓 ……(6) From the above, the primary current of the motor i ₁ 〓,
For control that cancels the interference caused by i ₁ Setting voltages e ^* _a , e ^* _b , and e ^* _c of the voltage source inverter can be obtained by calculating _p and preparing a trigonometric function generation circuit from _ωp .

Here, when formula (5) is introduced into formula (6), for e _1d , e _1d = (r ₁ i ^* ₁ 〓−ω _p L〓i ^* ₁ 〓)COSω _p t−(r ₁ i ^* ₁ 〓−
ω _p L,
i ^* ₁ 〓）SINω _p t ...(8) From the following relationship using the Laplace operator S in the above equation, −ω _p SINω _p t=d/dt(COSω _p t) =S・COSω _p t ω _p COSω _p t=d/dt (SINω _p t) = S・SINω _p t e _1d = (r ₁ +L ₁ S) COSω _p t・i ^* ₁ 〓−(r ₁ +L〓S) SIN
ω _p t・i ^* ₁ 〓……(9). Similarly, e _1q is converted to the following equation (10).

e _1d = (r ₁ + L ₁ S) Sinω _p t・i ^* ₁ 〓 + (r ₁ + L〓 S) COS
ω _p t・i ^* ₁ 〓...(10) Focusing on such conversion, the present invention provides synchronous rotation based on the above-mentioned equation (6) in the calculation circuit for interference cancellation and the phase voltage calculation circuit. Instead of preparing a calculation circuit for converting coordinates to fixed coordinates, (9),
By preparing an arithmetic circuit based on equation (10),
Aim to simplify the arithmetic circuit.

FIG. 3 is a block diagram showing one embodiment of the present invention. In controlling the motor 1 by supplying a voltage-controlled primary voltage from the PWM inverter 2 so that the magnetic flux and the secondary current are orthogonal to each other, the arithmetic circuit 3 calculates the equations (9) and (10) described above. α, β phase primary current setting value i ^* ₁ 〓,
Obtain fixed coordinate voltage setting values e ^* _1d and e ^* _1q for i ^* ₁ 〓.
The two-phase three-phase conversion circuit 4 has an arithmetic function based on the above-mentioned equation (7), and calculates three-phase coordinate voltage set values e * _a , e ^* for the outputs e ^* _1d , e ^* _1q of the arithmetic circuit ^3. _b , e ^* _c are obtained, and this set value is used as the voltage command for the inverter 2.

The β-phase primary current set value i ^* ₁ 〓 is a speed adjustment that performs proportional integral calculation (PI) of the deviation between the speed set value V ^* _S and the detected value (rotor angular frequency ω _r ) of the speed detector 5 coupled to the motor 1. It is obtained as the output of the device 6. The power supply angular frequency ω _p is obtained by the angular frequency calculation circuit 7. This arithmetic circuit 7 includes a divider ₇₁ that divides the set values i ^* ₁ 〓, i ^* ₁ 〓, and a coefficient that multiplies the division result i ^* ₁ 〓/i ^* ₁ 〓 by a coefficient 1/τ _2. The _rotor angular frequency ω _r is added to the slip angular frequency ω _S to obtain the _power supply angular frequency ω _p . Calculation of the slip angular frequency ω _S by the calculator 7 ₁ and the coefficient unit 7 ₂ is based on the following conditions mentioned above in the second term on the right side of equation (4) and from _{FIG .} Substitute M and get i ^* ₁ 〓/(i ^* ₁
〓・
τ ₂ ).

L〓=L ₁ L ₂ −M ² /L ₂ τ ₂ =L ₂ /r ₂ The trigonometric function generating circuit 8 obtains a sine wave SINω _p t and a cosine wave COSω _p t from the power supply angular frequency ω _p , and the arithmetic circuit 3 gives the trigonometric functions for that operation. The triangular wave generation circuit 9 uses the angular frequency ω _p to extract a triangular wave synchronized with the angular frequency ω _p , and supplies this triangular wave to the inverter 2, which generates the triangular wave and set voltages e ^* _a , e ^* _b , by level comparison with e ^* _c
Obtain PWM waveform.

FIG. 4 shows a circuit diagram of the main part of FIG. 3. The calculation circuit 3 generates a sine wave SINω _p t and a cosine wave COSω _p t passed through proportional differential (PD) calculators 3 ₁ to 3 ₄ and a set value i ^* ₁ 〓,
i ^* ₁ 〓 is input to multipliers ₃₅ to ₃₈ , multiplication is performed in these multipliers ₃₅ to ₃₈ based on equations (9) and (10), and these multiplication results are added and subtracted to set Values e _1d , e _1q
get. The arithmetic units 3 ₁ to 3 ₄ are realized, for example _, by the arithmetic circuit shown in FIG. It is set to ₁ . Here, the computing units 3 ₁ to 3 ₄ have a sine wave SINω _p t and a cosine wave COSω _p t
Instead of directly inputting , the integrator 3 ₉ , 3 ₁₀
intervene. The integrators 3 ₉ and 3 ₁₀ function as first-order delay elements for suppressing the amplification of high frequency components such as noise superimposed on the trigonometric function waves by the arithmetic units 3 ₁ to _{3 4} including differential elements. Figure 5b
This is realized by the circuit shown in . As the time constant T (=CR) of this first-order lag integrator circuit, the trigonometric function wave SINω _p
t, COSω _p is set to be sufficiently small compared to the period ω _p of t, and is set so as to remove high frequency components such as noise while making the influence of first-order delay elements in the control frequency region of the motor negligible. The integrators 3 ₉ and 3 ₁₀ are provided when the trigonometric function generating circuit 8 is a digital system in which positive and cosine waves are created in a digital circuit and D/A conversion is performed.
Even if the cosine wave is shaped in stages with many noise components, it becomes more effective in removing the noise components.

Next, the two-phase three-phase conversion circuit 4 has a circuit configuration based on the above-mentioned equation (7). In other words, in addition to using the value e _1d obtained by the arithmetic circuit 3 as the set value e ^* _a ,
Add _e1q to the value passed through the inverting amplifier ₄₁ with a gain of 1/2 and the value passed through the continuation circuit of the non-inverting amplifier ₄₂ with a gain of √3/2 and the inverting amplifier ₄₃ to obtain the set value e ^* _b . do. Further, the output of the amplifier 4 ₁ and the output of the amplifier 4 ₂ are added to obtain a set value e ^* _c .

The set values e ^* _a , e ^* _b , e ^* _c obtained in this way are compared in level with the output of the triangular wave generating circuit 9 by comparators 2 ₁ , 2 ₂ , and 2 ₃ in the inverter 2, respectively, and are then connected to the main circuit of the inverter. Switch PWM control signal
Ba, a, Bb, b, Bc, c are created.

As described above, according to the present invention, when performing vector control of an induction motor using a voltage source inverter, the inverter set voltage e is determined from the motor excitation current set value i ^* ₁ 〓 and the secondary current set value i ^* ₁ 〓. ₁ 〓, e ₁ 〓, by canceling the mutual interference due to the motor's primary currents i ₁ 〓, i _{1 〓} , the magnetic flux and secondary Enables control to always make the current vector orthogonal,
Furthermore, the arithmetic circuit for interference cancellation is capable of performing interference cancellation calculation and coordinate conversion calculation in order to perform a batch calculation including conversion from synchronous rotating coordinates to fixed coordinates (Equation 6) for phase voltage conversion. This has the effect of greatly simplifying the arithmetic circuit configuration compared to the case where they are separate.

[Brief explanation of the drawing]

Fig. 1 is an equivalent block diagram of an induction motor with respect to two-phase voltages e ₁ 〓, e ₁ 〓, Fig. 2 is an equivalent block diagram of vector control of an induction motor, and Fig. 3 is a control device showing an embodiment of the present invention. 4 is a circuit diagram of the main part of FIG. 3, and FIG. 5 is a specific circuit diagram a of the proportional differential calculator and a specific circuit diagram b of the integrator in FIG. 4. 1...Induction motor, 2...Voltage type inverter,
3... Arithmetic circuit, 4... 2-phase 3-phase converter, 5...
Speed detector, 6... Speed adjuster, 7... Angular frequency calculation circuit, 8... Trigonometric function generation circuit, 9... Triangular wave generation circuit.

Claims (1)

[Claims] 1. α-phase primary current setting value for driving the induction motor with a voltage-type inverter and setting the magnetic flux of the induction motor.
From i ₁ 〓* and the β-phase primary current setting value i ₁ 〓*, which sets the secondary current component, a, b, c of the above voltage type inverter
In a vector control device for an induction motor that obtains three-phase voltage settings e _a *, e _b *, e _c *, a sine wave having an angular frequency setting value ω ₀ of the voltage source inverter described above is proportional to the primary resistance r ₁ . and equivalent leakage inductance L〓
Calculate the proportional differential using the differential coefficient of , multiply this calculation result by the above setting value i ₁ 〓* to obtain the first calculation value, and calculate the cosine wave with the above setting value ω ₀ proportional to the primary resistance r ₁ and Calculate the proportional differential using the differential coefficient of the primary inductance L ₁ , and use the above set value as the result of this calculation.
Find the second calculated value multiplied by i ₁ 〓*, and make the above sine wave proportional to the primary resistance r ₁ and the primary inductance
Calculate proportional differentiation with the differential coefficient of L ₁ , multiply this calculation result by the above set value i ₁ 〓 to obtain the third calculation value, and calculate the above cosine wave in proportion to the primary resistance r ₁ and the equivalent leakage inductance L 〓 Calculate the proportional differential using the differential coefficient of , multiply this calculation result by the above set value i ₁ 〓 * to find the fourth calculation value, and subtract the first calculation value from the second calculation value to find the value e _1d and a calculation circuit that calculates the value e _1q obtained by adding the fourth calculation value to the third calculation value, and the set values e _a *, e _b by performing two-phase and three-phase conversion from the above values e _1d and e _1q . 2-phase 3 to find *, e _c *
A vector control device for an induction motor, characterized by comprising a phase converter. 2 The value obtained by dividing the above setting value i ₁ 〓* by the setting value i ₁ 〓* is the ratio of the equivalent secondary inductance L ₂ and the secondary resistance r ₂
The present invention is characterized by comprising an angular frequency calculation circuit that calculates the angular frequency set value ω ₀ by adding the detected rotor angular frequency ω _r of the induction motor to the value passed through a coefficient unit set to r ₂ /L ₂ . A vector control device for an induction motor according to claim 1. 3. The sine wave and the cosine wave are each provided to the arithmetic circuit through an integrator having a time constant T that is sufficiently smaller than the period of the angular frequency setting value ω ₀ . 2. A vector control device for an induction motor according to any one of Item 2.