JPH05259747A - Low distortion factor amplifier circuit - Google Patents

Abstract

PURPOSE:To reduce distortion by using a negative feedback type current mirror circuit with high accuracy as a current supply means. CONSTITUTION:Let a current amplification factor of each of transistors(TRs) Q11, Q12, Q13 of a current mirror circuit 10 be hfe, a base current of the TRs Q11, Q12 is equal to each other as iB. Each emitter current of the TRs Q11, Q12 is expressed as (1+ or -hfe).iB. Moreover, a reference side current i2 is nearly (1+hfe).iB. A follower side current i1 is a collector current of the TR Q13 and almost equal to (1+hfe).iB. Thus, the relation of i1=i2 is realized. On the other hand, the reference side of the current mirror circuit 20 is connected to the collector of the TR Q1. The follower side is connected to an output terminal OUT and an output voltage V0 is extracted. Through the constitution above, the relation of i3=i1=i2 is obtained and an amplified output without nonlinear distortion is obtained at the output voltage V0.

Description

Detailed Description of the Invention

[0001]

[0001]

[0002]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a low distortion rate amplifier circuit, and more particularly to a low distortion rate amplifier circuit in which signal distortion is removed without applying negative feedback.

[0003]

[0002]

[0004]

2. Description of the Related Art Conventionally, in order to remove the non-linear distortion of a transistor which is an amplifying element, a low distortion factor amplifying circuit of the type which does not perform negative feedback is known as shown in FIG.

In the figure, Q1 is a PNP type transistor, whose input terminal IN is connected to its base
i is input, the collector is connected to a negative power supply (not shown), the emitter is connected to the base of an NPN transistor Q2, and the PNP is connected between the collector and emitter of a cascode-connected PNP transistor Q3.
Type transistor Q52 is connected to the collector.

[0006]

The emitter of the transistor Q2 is grounded through a resistor RE, and the collector is a PNP type transistor Q.
Connected to 51 collectors. This transistor Q51
The base and the emitter of the PNP type transistor Q53 and the transistor Q52 are commonly connected to each other, and the base and collector of the transistor Q51 are connected to each other, so that the transistor Q51 is the reference side and the transistors Q52 and Q53 are the dependent sides. To constitute the current mirror circuit 50.

The positive power supply + VCC is supplied to the common emitter. The collector of the transistor Q53 is connected to the output terminal OUT via the emitter and collector of the cascode-connected PNP transistor Q4, and is also grounded via the load resistor RL. Load resistance R
The output voltage V0 generated at L is taken out from the output terminal OUT.

[0008]

Now, in this configuration, the transistor Q1
, Q2, Q3, and Q4 are considered to be sufficiently small and ignored, and the emitter current of the transistor Q1 is i1, the collector current of the transistor Q2 is i2, and the collector current of the transistor Q4 is i0. By the action of, the relationship of i0 = i1 = i2 (1) is established.

Assuming that the base-emitter voltages of the transistors Q1 and Q2 are VBE1 and VBE2, respectively, i2 = (Vi + VBE1-VBE2) / RE (2). In general, the relation between the base-emitter voltage VBE and the emitter current IE in a transistor is expressed by VBE = (kT / q) ln (IE / IS) (3), which is non-relative to the emitter current IE. It is a straight line. Here, k is the Boltzmann constant, T is the absolute temperature of the junction, q is the electron charge, and IS is the reverse saturation current.

[0010]

Therefore, VBE1-V in equation (2)
BE2 becomes VBE1-VBE2 = (kT / q) ln (i1 / i2) (4). The temperature conditions of both transistors Q1 and Q2 are assumed to be the same. Here, since i1 = i2 is set, the value of the equation (4) becomes zero, and a current i2 that does not include the non-linear distortion due to the base-emitter voltage VBE is obtained in the transistor Q2. This current i2 becomes i2 = Vi / RE (5).

[0011]

Further, since a current i0 equal to this value flows through the load resistance RL, the output voltage V0 becomes V0 = i0.RL = i2.RL = (RL / RE) .Vi (6).

Therefore, in the circuit of FIG. 4, the voltage amplification factor is RL.
/ RE operates as a voltage amplifier. The current ratio between the currents i0, i1 and i2 in the current mirror circuit 50 may not be set to 1, but may be set to a predetermined ratio by inserting resistors between the emitters of the transistors Q51, Q52 and Q53 and the positive power supply + VCC. However, in this case, the value of the equation (4) does not become zero but becomes a constant value, so that an output that does not include nonlinear distortion can be obtained in the same manner.

[0013]

Next, the functions of the transistors Q3 and Q4 will be described.

As described above, the transistors Q3 and Q4 act as current transmitters in which the collector current and the emitter current are equal, ignoring the base current, and the potentials of the respective emitters are such that the bias voltage VBB is equal to the base-emitter potential. Since it is applied with a voltage, it works to keep it at a nearly constant value.

If the transistors Q3 and Q4 are not provided,
Since the collector-emitter voltages of the transistors Q52 and Q53 change according to the input voltage Vi and the output voltage V0, respectively, the collector current fluctuates due to the Early effect and non-linear distortion occurs.
By 4, the collector-emitter voltage of the transistors Q52 and Q53 is maintained at a substantially constant value, and the occurrence of nonlinear distortion due to the Early effect is prevented.

[0016]

[0008]

[0017]

As described above, the conventional low distortion amplifier circuit requires a current mirror circuit of a system for extracting a plurality of currents on the subordinate side. Therefore, the current mirror circuit shown in FIG. 50, the feedback-type high-precision current mirror circuit that can be used in the present invention described later cannot be used. Therefore, the occurrence of distortion due to the low accuracy of the current value has been a problem.

[0018]

To explain this in detail, referring to FIG.
A transistor Q51 forming the current mirror circuit 50,
The characteristics of Q52 and Q53 are uniform, and the current amplification factor is hfe. Also, since the bases and the emitters are commonly connected and the base-emitter voltages are equal, the base currents are all equal, and this is iB. Then, all the emitter currents are (1 + hfe) · iB and are equal, but regarding the output current on the collector side, the output current of Q51, that is, i2 = (3 + hfe) · iB Q52, that is, i1 = hfe · iB Q53 output current, that is, i0 = hfe · iB, and they are not equal.

[0019]

Therefore, by making the currents i1 and i2 equal, the effect of canceling the non-linear distortion by eliminating the difference in the base-emitter voltage between the transistors Q1 and Q2 has a drawback. .

Further, in order to prevent nonlinear distortion due to the Early effect of the subordinate transistor of the current mirror circuit, a cascode-connected transistor is required.

According to the present invention, a current distortion circuit having a high current accuracy and a small effect of the Early effect can be used as a current supplying means, and a low distortion amplification circuit which eliminates the drawbacks of the conventional one is provided. To aim.

[0022]

[0011]

[0023]

In order to achieve the above object, in a low distortion amplification circuit of the present invention, a first transistor to which an input is applied to the base and an emitter of the first transistor are connected to the base. A second transistor having a conductivity type opposite to that of the first transistor, current supply means for supplying a current having a predetermined ratio to the emitter side of the first transistor and the collector of the second transistor, and the collector of the first transistor. And an output means for taking out an output current or an output voltage according to the collector current of the first transistor.

[0024]

[0012]

[0025]

In the low distortion amplification circuit of the present invention, the input current is amplified by the first transistor and the second transistor, and the current value of the second transistor is determined by the emitter resistance of the second transistor and the potential of the emitter. Since the emitter current of the first transistor is made equal to the current of the second transistor by the current mirror circuit,
The potential of the emitter of the second transistor becomes exactly equal to the input voltage because the base-emitter voltage of both transistors is accurately canceled. Therefore, the emitter current of the second transistor is a current that is not affected by the base-emitter voltage that causes the non-linear distortion and is not the non-linear distortion. Since the collector current of the first transistor is also equal to this, the voltage or current generated thereby from this collector is taken out as output.

[0026]

[0013]

[0027]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 2 is a diagram for explaining the principle of the low distortion amplification circuit of the present invention. In the figure, the first transistor Q
1 is a PNP type transistor, and the input terminal IN is at the base
Are connected and the input voltage Vi is input. The collector is grounded via the load resistance RL, and the voltage generated at this is taken out as the output voltage V0.

The emitter of the transistor Q1 is connected to the base of the second NPN transistor Q2.
The emitter of the transistor Q2 is grounded through the resistor RE, the collector of the transistor Q2 is connected to the current source 31, and the emitter of the transistor Q1 is connected to the current source 32. Here, the current sources 31 and 32 are the current sources 31
Is a reference side, and the current source 32 is a subordinate side whose current value is equal to that of the reference side.

[0029]

In this configuration, the transistors Q1 and Q
Ignore the base current of 2 as sufficiently small, and
If the base-emitter voltages are VBE1 and VBE2, the potential VE2 of the emitter of the transistor Q2 is VE2 = Vi + VBE1-VBE2 (7). The emitter current i2 of the transistor Q2 is i2
= VE2 / RE. Then, the current i1 = i2 is supplied to the emitter of the transistor Q1. Therefore, VBE1-VBE2 in the equation (7) becomes zero as in the above-mentioned equation (4), so that i1 = i2 = VE2 / RE = Vi / RE (8). Since the current flowing through the load resistor RL is equal to i1, the output voltage V0 is V0 = i1.RL = (RL / RE) .Vi (9), and an output voltage free of non-linear distortion can be obtained.

[0030]

FIG. 1 is a circuit diagram showing an embodiment of the present invention based on the above principle. It is to be noted that the components denoted by the same reference numerals as those in FIG. 2 have the same configurations and functions, and the description thereof will be omitted.

In FIG. 1, 10 is a current mirror circuit, which is a specific circuit of the current mirror circuit 30 in FIG. Transistors Q11, Q12, Q13
Constitutes a well-known feedback type current mirror circuit, in which the transistor Q11 side is the reference side and the transistors Q12, Q13 side is the subordinate side.

This feedback type current mirror circuit is characterized in that the accuracy of the current value is higher than that of the basic current mirror circuit shown in FIG. 4 of the conventional example.
The part will be described in detail as an example.

[0033]

Since the transistors Q11, Q12, Q13 have the same characteristics and the current amplification factor is hfe, the bases and the emitters of the transistors Q11, Q12 are connected in common and the base-emitter voltages are equal. , The base current is also the same, and this is iB. Then, the emitter currents of the transistors Q11 and Q12 are equal and (1
+ Hfe) ・ iB.

The emitter current of the transistor Q13 is equal to the collector current of the transistor Q12 and the transistors Q11 and Q12.
Of the base current iB of (2 + hf
e) · iB. Base current iB of transistor Q13
13 is one part of this hfe, so iB13 = (2 + hf
e) · iB / hfe, but if hfe is sufficiently larger than 2, it can be approximated as iB13 堯 iB.

[0035]

Since the reference current i2 is the collector current of the transistor Q11 plus iB13,
It is almost (1 + hfe) · iB.

Since the current i1 on the dependent side is the collector current of the transistor Q13, its emitter current becomes iB13.
Is subtracted, and is approximately (1 + hfe) · iB. Therefore, the relationship of i1 = i2 can be realized with high accuracy.

Further, in this feedback type current mirror circuit, the collector-emitter voltage of the reference side transistor (Q11) is about 1.2 V, and the dependent side transistor (Q12).
Since the collector-emitter voltage of is fixed at about 0.6 V, there is an advantage that the influence of the Early effect itself is small.

[0038]

On the other hand, the collector side of the transistor Q1 is connected to the reference side of another current mirror circuit 20 connected to the negative power source -VEE instead of the load resistance RL in FIG. 2, and the subordinate side of this current mirror circuit 20 is connected. , Is connected to the output terminal OUT while being grounded via the load resistance RL, and the output voltage V0 generated at the load resistance RL is taken out.

The current mirror circuit 20, like the current mirror circuit 10, is a feedback type current mirror circuit, and the transistors Q21, Q22, Q23 are transistors Q11, Q12, Q12 of the current mirror circuit 10, respectively.
Each corresponds to Q13. However, since the polarities of the power supplies are opposite, the conductivity types of the transistors are complementary.

[0040]

With the above configuration, the transistors Q1 and Q
Ignoring the base current of 2, i3 = i1 = i2, and the output voltage V0 is V0 = -i3.RL =-(RL / RE) .Vi (10), and amplification without non-linear distortion Output is obtained.

As is clear from the figures, the circuits shown in FIGS. 1 and 2 have an amplifying function when the range of the input voltage is positive, more precisely, at a voltage higher than the base-emitter voltage. .. Therefore, when the input voltage is AC, a predetermined bias voltage may be added to the input voltage so that the cutoff is not performed in the negative half cycle.

[0042]

FIG. 3 shows another embodiment of the low distortion rate amplifier circuit of the present invention. A bias voltage for this purpose is given as VB, and a positive / negative side is configured to be complementary and push-pull. It is designed to get the output. In addition, in the figure, those having the same reference numerals and having a suffix "" correspond to each other.

Here, i1 = i2 = i3 = (Vi + VB) / RE (11) i1 '= i2' = i3 '= (-Vi + VB') / RE '... (12). The current i0 flowing through the load resistance RL is
If VB = VB 'and RE = RE', then i0 = i3'-i3 = -2.Vi / RE (13). Therefore, the output voltage V0 is V0 = i0.RL = -2. (RL / RE) .Vi (14), and the voltage amplification factor is twice as large as that in FIG. It becomes an amplifier circuit.

[0044]

In the above embodiments, as the method for extracting the amplified output from the collector of the first transistor Q1, the method for directly extracting the voltage from the load resistor and the method for temporarily extracting the current as a current by the current mirror circuit have been described. However, it goes without saying that the output can be taken out by another circuit such as a buffer circuit.

[0045]

[0022]

[0046]

Since the low distortion amplification circuit of the present invention is constructed as described above, a highly accurate feedback type current mirror circuit can be used as a current supply means, and a non-current caused by a base-emitter voltage of a transistor can be eliminated. Cancellation of linear distortion is sufficient, and distortion can be reduced. In addition, the influence of the Early effect, which is another cause of distortion, is reduced.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a circuit diagram illustrating the operating principle of the present invention.

FIG. 3 is a circuit diagram showing another embodiment of the present invention.

FIG. 4 is a circuit diagram showing a conventional example.

[Explanation of symbols]

 10, 20, 30, 30 'Current mirror circuit 40, 40', 50 Current mirror circuit 31, 31 ', 32, 32' Current source 41, 41 ', 42, 42' Current source IN input terminal OUT output terminal RE, RE 'resistance RL load resistance VB, VB', VBB bias voltage Vi input voltage V0 output voltage

Claims (1)

[Claims] 1. A first transistor to which an input is applied to a base, a second transistor having a conductivity type opposite to that of the first transistor, an emitter of the first transistor being connected to the base, and an emitter side of the first transistor. The second
Current supply means for respectively supplying a current of a predetermined ratio to the collector of the transistor, and output means connected to the collector of the first transistor for extracting an output current or output voltage according to the collector current of the first transistor are provided. A low distortion amplification circuit characterized by.