JPH05157837A - Signal processing method for encoding cw rader and apparatus thereof - Google Patents

Abstract

PURPOSE:To prevent increase of erroneous alarm probability due to addition of noise and central spectrum. CONSTITUTION:A power distributer 50 for distributing a demodulated signal outputted from each range bin 38 in 0 and pi phase and a power synthesizer 52 for synthesizing demodulated signals destributed from adjoining range bins 38 in mutually opposite phases are provided. In the case when mutually adjoining range bins 38 are different in code, Doppler signs are canceled by opposite-phase addition in a power distributer 50 and a power synthesizer 52, so that the Doppler signal becomes zero in principle and there is no longer possibility of an erroneous signal even in use of a threshold circuit. In the case when either of the adjoining range bins 38 is different in code, the Doppler signal is intensified by opposite-phase addition in the power distributer 50 and the power synthesizer 52. Then due to installation of a limiter IF amplifier 54 in the subsequent stage of the power synthecizer 52, a sufficient S/N ratio is obtained irrespective of level of the Doppler signal, so that higher harmonics occurring due to the limiter IF amplifier 54 is removed by a low pass filter 56.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a coded CW radar for detecting a target by transmitting and receiving a continuous wave (CW) coded by a pseudo-random sequence, and more particularly to a signal processing system and apparatus thereof.

[0002]

2. Description of the Related Art Conventionally, a coded CW radar using a continuous wave (CW) phase-modulated (binary phase modulation) with a binary value of 0 or π by an M sequence is known. FIG. 2 shows a conventional example of DF Albanese, AMKlein, IEEE TRANSACTION AE.
ROSPACE AND ELECTRONICSYSTEMS vol.AES-15 No.1 Jan.
A conventional coded CW radar disclosed in 1979 is shown. This radar has a carrier frequency of X band and a code bit number of 63 bits.

The device shown in this figure first comprises an oscillator 10 for oscillating a carrier wave. A two-phase modulator 14 and a mixer 16 are connected to the latter stage of the oscillator 10 via a power distributor 12. That is, the power distributor 12 is
A carrier oscillated by the oscillator 10 is supplied to the two-phase modulator 14
And to the mixer 16. The two-phase modulator 14 modulates a carrier wave using the M-sequence code as a modulation signal (pseudo-random coding) and generates a transmission signal. The transmission signal is subjected to wide band power amplification by the power amplifier 18 connected to the latter stage of the two-phase modulator 14 and transmitted from the transmission antenna 20 as a continuous wave.

On the other hand, the M-sequence code supplied to the two-phase modulator 14 is generated by the code generator 22. The code generator 22 generates an M-sequence code in synchronization with the clock supplied from the clock generator 24, and supplies the M-sequence code to the two-phase modulator 14 and a code delay circuit 26 described later.

The mixer which receives the carrier wave distribution by the power distributor 12 is connected to the IF oscillator 28. That is, the IF oscillator 28 generates a reference IF signal and supplies it to the mixer 16. The mixer 16 performs frequency conversion to generate and output a reference signal for reception.

On the other hand, in this conventional example, a receiving antenna 30 is first provided as a receiving structure. The receiving antenna 30 receives a reflected signal (target signal) obtained as a result of the continuous wave wirelessly transmitted by the transmitting antenna 20 being reflected by the target 100, and also receives clutter, spillover, and the like. The reception signal obtained from the reception antenna 30 is input to the power distributor 36 via the mixer 32 and the wideband IF amplifier 34.

The mixer 32 mixes the received signal with the reference signal generated by the above-mentioned mixer 16 and converts it into an IF signal. The received signal as the IF signal obtained by the mixer 32 is wide-band amplified by the wide-band IF amplifier 34, and further L power range bins 38 by the power distributor 36.
Will be distributed to. Each range bin is composed of a two-phase demodulator 40, a notch filter 42, a narrow band IF amplifier 44, a mixer 46 and a Doppler filter bank 48, respectively.

Two-phase demodulator 40 constituting the range bin 38
Outputs the demodulated signal by demodulating the received signal with the demodulation code output from the code delay circuit 26 described above. That is, the code delay circuit 26 delays the M-sequence code generated by the code generator 22 and used for the pseudo-random coding by the two-phase modulator 14 by 1, 2, ... L bits, and delays the M-sequence code. The first, second, and
It is supplied to the two-phase demodulator 40 that constitutes the Lth range bin. The two-phase demodulator 40 applies the received signal (IF signal) distributed by the power distributor 36 to the code delay circuit 26.
A phase shift of 0 or π is performed by the demodulation code supplied from the device, and thereby the code is released and a demodulation signal is output.

The notch filter connected after the two-phase demodulator 40 is a filter for removing components such as clutter and spillover. That is, as described above, the signal received by the receiving antenna 30 includes the target signal as well as clutter, spillover, and the like. The notch filter 42 removes the clutter and spillover from the I signal.
It is tuned to the F center frequency.

Further, a narrow band IF amplifier 44 is connected to a stage subsequent to the notch filter 42, and this narrow band IF amplifier 44 amplifies the IF signal output from the notch filter 42 in a band equal to the Doppler band. The Doppler band is a Doppler frequency band included in the reflected signal that is generated as the target 100 moves.

The mixer connected to the subsequent stage of the narrow band IF amplifier 44 receives the IF signal output from the amplifier 44 and the I signal.
It mixes with the reference IF signal generated by the F oscillator 28, thereby converting it to a Doppler signal having a Doppler frequency. When the Doppler signal is obtained in this manner, this signal is supplied to the Doppler filter bank 48, and the velocity information of the target 100 is obtained.

FIG. 3 shows the autocorrelation function of the M-sequence code used in this conventional example, and FIG. 4 shows its power spectrum. As shown in FIG. 3, the autocorrelation functions of these M-sequence codes have peaks at τ _b / τ = 0, L + 1, ... When the 1-bit length of the code is τ _b, and at other τ _b / τ = 1, 2, ...
It has a value of −1 / L in L. τ is the time difference between the transmission and reception sequences.

The power spectrum of the M-sequence code is
As shown in FIG. 4, 1 / L ^{2 at} frequency f = 0
Has a value of.

Therefore, when the M-sequence code delayed by the code delay circuit 26 with a different number of bits is supplied to the two-phase demodulator 40 of each range bin 38 as a demodulation code and two-phase demodulation is performed, the M-sequence code matches the demodulation code. In the two-phase demodulator 40 of the range bin 38 to which the received signal encoded by the code is input, the code is released, so that the power spectrum of the demodulated signal becomes an unmodulated CW. Further, in the range bin 38 that does not match the demodulation code, the code cannot be released, so the power spectrum becomes the power spectrum as shown in FIG.

On the other hand, a narrow band IF amplifier 44 is used in each range bin 38. As described above, the narrow band IF amplifier 44 is an amplifier in the same band as the Doppler frequency band related to the movement of the target 100. By providing this amplifier 44, the power spectrum of the received signal is the center spectrum (f = 0). Spectrum). Therefore, a relatively high power Doppler signal is output from the range bin 38 in which the demodulation code and the code of the received signal match. On the contrary, from the range bin 38 in which the demodulation code and the code of the received signal do not match, only the Doppler signal having the power of 1 / L ² can be obtained. Therefore, in the Doppler filter bank 48, it is possible to identify whether or not the signal output from the mixer 46 is a Doppler signal related to the target signal, based on whether or not the power of the signal is high.

Since the demodulation code is the M-sequence code for transmission shifted by 1, 2, ..., L bits by the code delay circuit 26 as described above, it is the same as the M-sequence code for transmission. The time difference from each demodulation code is known. Therefore, depending on which range bin 38 the Doppler signal is obtained, the distance of the target 100 is
It can be known with high resolution.

Further, similar to the FM-CW radar, the pulse Doppler radar, etc., since the Doppler filter bank 48 is used to obtain the velocity information, it is possible to obtain good resolution with respect to the velocity information.

[0018]

However, in the conventional coded CW radar having such a configuration,
It was necessary to determine whether or not the power of the signal output from the mixer was high. As a circuit used for this determination, for example, a threshold circuit is generally used.
However, when the threshold circuit is used, there is a limit in reducing the false alarm probability.

That is, for the range bins in which the code included in the received signal and the demodulation code do not match, the target of comparison judgment with the threshold level in the threshold circuit is the center spectrum shown in FIG. That is, it is the sum of 1 / L ² Doppler signal and noise. The probability that this sum exceeds the threshold level (false alarm probability) is higher than the false alarm probability when there is no target in the beam of the receiving antenna, that is, the probability that noise exceeds the threshold level. As a result, it is not possible to reduce the rate of false alarms. To avoid this, set the minimum detection S / N ratio to an extremely large value,
Alternatively, it is necessary to make the code bit number L extremely large.

The present invention has been made to solve the above problems, and in principle, in a range bin in which the received signal code and the demodulation code do not match,
By suppressing the Doppler signal to 0, the minimum detection S /
The object is to reduce the false alarm probability while avoiding the extreme setting of the N ratio and the code bit number L.

[0021]

In order to achieve such an object, the signal processing system for coded CW radar according to the present invention emphasizes a target signal and suppresses components appearing in phase in the demodulated signal of each range bin. It is characterized in that demodulated signals obtained in two adjacent range bins are combined in antiphase so as to suppress the removal.

The coded CW radar signal processing device according to a second aspect of the present invention is provided with first and second signals which are provided for each range bin and have opposite phases with respect to signals obtained by two-phase demodulation.
And a combiner that inputs and combines the first signal from one of the range bins adjacent to each other and the second signal from the other, and outputs the combined signal. Characterize.

Further, in the coded CW radar signal processing device according to the third aspect, a limiter amplifier for amplifying the output of the combiner and limiting the amplitude is provided.

[0024]

In the signal processing system of the present invention, the demodulated signals obtained in two adjacent range bins are combined in antiphase. As a result, the reflected signal from the target is emphasized, and the signals (noise, interference wave, etc.) that appear in phase with the demodulated signal of each range bin are suppressed. As shown in FIG. 3, the pseudo-random sequence autocorrelation function has a large value when the signs match, and has a small value in other cases. When two range bins that are adjacent to each other are range bins related to the sign mismatch, the Doppler signals included in the demodulated signal have the same phase and the same amplitude, and the Doppler signal becomes 0 in principle due to the antiphase combination. In addition, when one of the two range bins adjacent to each other is the range bin related to the sign matching and the other is the range bin related to the sign mismatch, the Doppler signals included in the demodulated signal have opposite phases, as is clear from FIG. As a result, the Doppler signal related to the reflection from the target is emphasized.

Therefore, in the present invention, even when the threshold circuit is used, it is possible to prevent a false alarm due to the addition of noise to the signal of the central spectrum.

Further, in the signal processing device according to the second aspect of the present invention, the device on the receiving side which realizes the above-mentioned action of the first aspect is composed of a distributor and a combiner.

Further, in the apparatus of the present invention, when the limiter amplifier is provided, the amplitude limitation works for a high level Doppler signal, and the amplification effect works for a low level Doppler signal. A sufficient S / N ratio can be obtained regardless of the level.

[0028]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT A preferred embodiment of the present invention will be described below with reference to the drawings. Note that the same components as those of the conventional example shown in FIG.

FIG. 1 shows the configuration of a signal processing device for a coded CW radar according to an embodiment of the present invention. The apparatus shown in this figure is obtained by partially improving the configuration of the receiving side in the apparatus of the conventional example shown in FIG. 2, and includes the configuration of the transmitting side including the code delay circuit 26 and the like, and the configuration of the receiving side. The configurations from the receiving antenna 30 to the wide band IF amplifier 34 are omitted for simplification of the drawing.

This embodiment is largely different from the above-mentioned conventional example in that a power distributor 50, a power combiner 52, a limiter IF amplifier 54, and a low-pass filter 56 are provided between the narrow band IF amplifier 44 and the mixer 46. Is the intervening point. The power distributor 50 is connected to the subsequent stage of the narrow band IF amplifier 44, and distributes the demodulated signal obtained from the narrow band IF amplifier 44 into 0 phase and π phase. That is, signals having the same power but opposite phases are distributed. Power distributor 50
The signal distributed by is input to the power combiner 52. The power combiner 52 includes the range bins 38 adjacent to each other.
A signal having a 0-phase signal and a π-phase signal is input from the power amplifier 50 of FIG. Limiter IF amplifier 54
Performs band-limited amplification of the signal combined by the power combiner 52, and the low-pass filter 56 low-pass filters it and supplies it to the mixer 46.

With such a configuration, particularly the power divider 50 and the power combiner 52, the Doppler signal included in the received signal is emphasized, and clutter, spillover, interfering waves, etc. are removed. That is, when the range bins 38 adjacent to each other are the range bins 38 in which the code of the received signal and the demodulation code do not match, the signal supplied from the narrowband IF amplifier 44 to the power distributor 50 is as shown in FIG. In-phase and amplitude. Therefore, when this is output as a π phase by the power distributor 50 related to one range bin 38 and output as a 0 phase from the power distributor 50 related to the other range bin 38, and both are combined by the power combiner 52, The resulting signals are such that the Doppler signals have been cancelled. That is, the Doppler signal becomes 0 due to the effect of subtraction.

When the code of the received signal and the demodulated code match in one of the range bins 38 adjacent to each other, the Doppler signals included in the signal output from the narrow band IF amplifier 44 are opposite to each other. Be in phase. Therefore, the Doppler signal is emphasized as a result of the anti-phase addition by the power distributor 50 and the power combiner 52.

Therefore, even if the output of the mixer 46 is judged by the threshold circuit, the target 100
The false alarm probability is almost the same as when the antenna is not in the beam of the antennas 20 and 30, and the false alarm can be reduced as a whole.

Further, even when the radio wave received by the receiving antenna 30 contains an interfering wave, this can be similarly removed. That is, since the interference wave appears in the same phase for each range bin 38, it can be removed by the anti-phase addition by the power distributor 50 and the power combiner 52, and a coding CW radar device that is strong against interference can be obtained. ..

Further, by the limiter IF amplifier 54,
A sufficient S / N ratio can be obtained for both high level Doppler signals and low level Doppler signals. For example, for a high level Doppler signal, the limiter I
The amplitude is limited by the F amplifier 54, and a sufficient amplitude can be obtained for the Doppler signal having a low level by the amplifying action of the limiter IF amplifier 54. Therefore, a sufficient S / N ratio can be obtained regardless of the level.

Further, by providing a low-pass filter 56 after the limiter IF amplifier 54, the limiter I
The high frequency generated in the F amplifier 54 can be removed.

[0037]

As described above, according to the present invention,
Since the reflected signals from the target are emphasized and the signals appearing in the same phase in the demodulated signals of each range bin are removed and suppressed, the demodulated signals obtained in the two range bins adjacent to each other are combined in the opposite phase. It is possible to realize a coded CW radar that is resistant to spillover, interference waves, etc. and has a low false alarm probability.

Further, when the limiter amplifier is provided, the S / N ratio of the Doppler signal is thereby increased regardless of the level.
It is possible to secure the ratio, and it is possible to detect multiple targets in the same beam.

[Brief description of drawings]

FIG. 1 is a block diagram showing the configuration of a coded CW radar signal processing device according to an embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration of a coded CW radar signal processing device according to a conventional example.

FIG. 3 is a diagram showing an autocorrelation function of an L-bit pseudo-random sequence.

FIG. 4 is a diagram showing a power spectrum of an L-bit pseudo random sequence.

[Explanation of symbols]

 14 two-phase modulator 20 transmitting antenna 22 code generator 26 code delay circuit 30 receiving antenna 36 power distributor 38 range bin 40 two-phase demodulator 42 notch filter 44 narrow band IF amplifier 46 mixer 48 Doppler filter bank 50 power distributor 52 power Combiner 54 Limiter IF amplifier 56 Low-pass filter 100 Target L Pseudo-random sequence bit number

Claims (3)

[Claims] 1. An L-bit pseudo-random sequence is two-phase modulated and wirelessly transmitted as a transmission signal, while the transmission signal is shifted by n bits (1 ≦ n ≦ L) to generate a demodulation sequence,
The target signal obtained by reflecting the target on the target signal is received, the received signal is distributed to L range bins, and the received signal is two-phase demodulated using a demodulation sequence related to n-bit shift in the nth range bin. In a coded CW radar signal processing method, a target signal is emphasized, and demodulated signals obtained in two adjacent range bins are combined in reverse phase so as to suppress and suppress components appearing in the same phase in the demodulated signal of each range bin. A signal processing method for a coded CW radar, comprising: 2. A means for outputting as a reception signal a target signal obtained by reflecting a transmission signal obtained by two-phase modulating an L-bit pseudo-random sequence by a target, and shifting the pseudo-random sequence for transmission by different numbers of bits. L range bins for two-phase demodulating the received signal by the demodulation sequence
In a coded CW radar signal processing device including: a distributor for distributing signals obtained by two-phase demodulation provided for each range bin as first and second signals having mutually opposite phases; and a range bin adjacent to each other. First from one of the distributors
The signal processing device for a coded CW radar, comprising: a combiner that inputs the second signal from the other, and combines and outputs the second signal. 3. The coded CW radar signal processing device according to claim 2, further comprising a limiter amplifier for amplifying the output of the combiner and limiting the amplitude.