CA1221755A - Radio signalling equipment - Google Patents

Abstract

ABSTRACT

Radio signaling equipment, including receiver apparatus, for receiving signals phase-modulated by a predetermined code sequence of code signals having a predetermined regular digit-signal rate. The apparatus further includes a decoding means for applying a sequence of phase-shifts to the received signal; first and second signal paths; a multi-tap delay means connected in at least one of the signal paths for progressively integrating signals in one of the signal paths; and a signal mixer means to receive at one input the integrated signals and at a second input signals from the second signal path. A time delay between adjacent taps on the multi-tap delay means is an integral multiple of the bit duration. The signal mixer derives an auto-correlation signal whose magnitude is dependent on the relative phases of the signals so received. In one form, the delay means is an acoustic delay line having a plurality of transducers equispaced along its length.

Description

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The present invention relates to radio signaling equipment and in particular to apparatus for enhancing the probability of detection of a signal at a receiver.
In known radar apparatus, it is common to use a buffs modulated carrier wave as a transmitted signal and to provide an identically coded sequence of signals which can be delayed and correlated with signals no-Elected from a target and received at a receiver to give the required target range information. The modulation of the carrier wave is achieved either by phase shift keying techniques in which the keying operations are controlled by the output ox a digital code generator, or injection phase locking the carrier wave oscillator to a stored replica of the previous transmitted pulse in an interrupted carrier wave coherent radar system.
In known telecommunications apparatus, it is common to use a binary signal representation of data for transmission and to code each l-signal and each 0-signal of the representation with a separate code sequence.
In this way each l-signal and each 0-signal of a given duration is sub-divided in a predetermined way into a number of much shorter duration l-signals and 0-signals. The effect of this is to widen or spread -the frequency spectrum of the transmitted signals so that reception and de-coding by an unwanted recipient is difficult. In the telecommunications apparatus it is also common to use a buffs modulated carrier wave as the transmitted signal where the buffs modulations are controlled by the coded data signal.
It is an object of this invention to provide receiver apparatus for a radar or telecommunications system in which the buffs coded signal is received and detected in-a manner which is relatively insensitive to the frequency of the return signal (to Doppler frequency shifts have no-natively little effect on the detection probability) and in which the signal-to-noise ratio of the received signal is enhanced significantly above that achieved in alternative Doppler independent receiver circuits.

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According to the present invention, a radio signaling equipment includes receiver apparatus, for receiving signals phase-modulated by a predetermined code sequence of code signals having a predetermined regular digit-signal rate, the said apparatus including:-a decoding means for applying a sequence of phase-shifts, complementary to those given by -the code sequence, to the received signals;
a first signal path and a second signal path having a common input connected to an output of the decoding means;
a multi-tap delay means connected in at east one of the signal paths for progressively integrating signals in one of these signal paths, the time delay between adjacent taps on the multi-tap delay means being an integral multiple of the said bit duration;
` and signal mixer means connected to receive at one input the integrated signals from the multi-tap delay means and at a second input signals from the second signal path, for deriving an auto-correlation signal the magnitude of which is dependent on the relative phases of the signals so received.
me multi-tap delay means may be an acoustic delay line having a plurality of transducers equispaced along its length, the distance between any two adjacent transducers being an integral multiple of -the said bit duration at the appropriate acoustic wavelength.
The transmitter apparatus associated with the receiver apparatus described above may include a pulse generator circuit for generating pulses at the desired pulse repetition frequency (PRY) of the radio signaling equipment, a surface wave acoustic delay line, two input transducers, one at each end of the surface wave acoustic delay line an integral number of acoustic wavelengths at an intermediate frequency tiff apart and each connected to an output of the PRY generator, and an output transducer - positioned on the delay line between the two input transducers and so-pirated from each by an integral number of acoustic wavelengths at the said IF.

~2~55 The PRY generator may be an IF surface wave acoustic oscillator, such as is described in Ultrasonics - May 1974 at page 115 in the paper by MY Lewis "Surface Acoustic Wave Devices and Applications", driving a step recovery diode impulse generator. The output of the delay line may be connected to a random or pseudo-random buffs modulating circuit synchronizing clock pulses for which may be derived from the output of the step recovery diode impulse generator.
An embodiment of the invention as applied to an interrupted carrier wave phase coherent proximity fuzz radar will now be described by way of example only and with reference to the accompanying drawing which shows the transmitter and receiver circuits of a proximity fuzz radar in scheme r circuit diagram form.
In the drawing the IF stages of a proximity fuzz radar carried for example in the warhead of a Surface-to-Air missile (SAY), comprise a surface acoustic wave oscillator 1 providing signals at 50 MHz which are frequency divided by a frequency divide by 10 circuit 2 to give a 5 MHz reference signal. A step recovery diode impulse generator circuit 3 is connected to the output of the frequency divide by 10 circuit 2 so as to produce sharp rising pulses corresponding to each positive peak of the 5 MHz signal hence determining the pulse repetition frequency (PRY) of the radar system. The output of the step recovery diode impulse generator circuit 3 is connected to two input transducers 4 and 5 at the extremities of a lithium niobate surface acoustic wave delay line 6 and also to the clock input of a pseudo-random code generator circuit 7. The input trays-dupers 4 and 5 on the lithium niobate surface acoustic wave delay line 6 are separated by an integral number of acoustic wavelengths at a frequency of 100 MHz. Each pulse from the step recovery diode impulse generator circuit 3 causes the transducer 4 and the transducer 5 to resonate at 100 MHz for 60 no, to 30% of the interval between successive pulses from the step recovery diode impulse generator circuit 3. Between the input trays-dupers 4 and 5, also an integral number of acoustic wavelengths from each, is an output transducer 8. The output of the transducer 8 is connected to one input of a buffs modulator circuit 9 another input of which is connected to the output of the pseudo-random code venerator circuit 7.
Ire pseudo-random code generator circuit 7 produces a 1023 bit pseudo-random sequence of binary signals each bit of which is produced a-t its output at a time corresponding to the receipt of a clock pulse from the step recovery diode impulse generator circuit 3. The phase of each 60 no 100 MHz signal at the output of the transducer 8 is determined in the buffs modulator circuit by the polarity of the binary signal received coincidentally from the pseudo-random code generator circuit 7. For example, the buffs modulator circuit 9 may give a 0 phase shift to one of the 60 no 100 Miss signals when a 0-signal is received from the pseudo-random code generator circuit 7, and a phase shift to another 60 no 100 MHz signal on receipt of a l-signal.
The output of the buffs modulator circuit 9 is connected to one input of an up-mixer circuit 10 another input of which is driven from a local oscillator circuit 11 which may be a surface acoustic wave device.
The up-mixer circuit lo maintains the phase of the IF buffs modulated pulses but increases their frequency to a suitable microwave frequency for transmission. The output of the up-mixer circuit 10 is amplified in a radio frequency power amplifier 12 and fed via a transmit-receive (TRY) switch 13 to an aerial 14 for transmission. The transmitted signal thus consists of a coded sequence of buffs modulated microwave pulses which will be reflected by a target 15 and received by the aerial 14 at a time dependent on the Range of that target 15. The TRY switch 13 is operated by pulses derived from the step recovery diode impulse generator circuit 3 such that for the first 60 no following a pulse from the generator circuit 3 the transmitter is connected to the aerial 14 and for the remaining 140 no between successive pulses from the step recovery diode impulse generator 3 the aerial is connected to the receiver. This is described as 30% duty cycle operation.

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The receiver circuit is shown in the drawing. In the receiver a preamplifier circuit 16 has an input connected via the TRY switch 13 to the aerial 14 and an output connected to a down-mixer circuit 17 which has another input connected to the output of the local oscillator 11.
The down-mixer circuit 17 converts the received microwave frequency signals to the 100 M~lz IF maintaining the phase relationship between successive pulses. The IF signal is amplified in conventional IF amplifier stages 18 and fed to one input of a demodulator circuit 19. Another input of the demodulator circuit 19 is connected via a range delay circuit 20 to the output of -the pseudo-random code generator circuit 7. The demodulator circuit 19 operates in a similar manner to the buffs modulator circuit 9 to the phase of each 60 no 100 MHz signal received from -the IF stages 18 is determined in the demodulator circuit 19 by the polarity of the delayed binary signal received coincidentally from the pseudo-random code generator circuit 7. Louvre, the demodulator circuit 19 applies phase shifts to the receive signals which are complementary -to those applied by the buffs modulator circuit 9 in the transmitter. Thus, if the example given above of the operation of the buffs modulator circuit 9 was the case, the demodulator circuit 9 would give a or phase shift of one received signal when a 0-signal is received from the pseudo random code generator circuit 7 and a 0-phase shift to another receive signal on receipt of a l-signal. When the target is at a range corresponding to the time delay given to the binary code sequence by the delay circuit 20 all the received signal applied to the demodulator 19 will be converted in the demodulator circuit 19 to a common phase sign].
The output of the demodulator circuit 19 is split into two channels.
One channel is fed directly to one input of a phase sensitive detector 21 and the other channel is fed to an input transducer on a 80 tap quart delay line 22. Each of the 80 taps on the quartz delay line 22 are communed together to form an output of the delay line 22 which is connected to a further input of the phase sensitive detector 21. The output of the phase sensitive detector 21 is connected via a low-pass filter 23, an integrator so circuit 24 and a threshold circuit 25 to an output of the system which in this case is connected to a detonation circuit (not shown).
The 80 tap delay line 22 and the phase sensitive detector I form an auto-correlator which together with the circuits 23 to 25 form an arrangement in which each received pulse is first auto-correlated with a one-bit delayed version of the preceding pulse in order to overcome carrier frequency variations due, for example, to the Doppler effect.
In this case however the one-bit delay is replaced in the auto-correlator by the 80 tap delay line 22 which in total provides a 80 bit delay so that the phase sensitive detector 21 compares the phase of the sty pulse with the integrated phase of the previous 80 pulses. This is a pre-detection integration technique which provides significant signal-to-noise enhance-mint in the receiver.
It will be appreciated that in interrupted carrier wave systems the problem was to provide a transmitter which could be completely switched off during receive periods, so that -the signal-to-noise ratio of the received signal was not impaired, and yet when switched on for the next transmit signal to have its phase coherently preserved in relation to the previous transmission.
In the apparatus described above however a carrier-wave oscillator is not used and the requisite quiet receive periods and phase coherence is achieved by the impulse generator circuit 3 and the delay line 6 to simulate a near perfect interrupted carrier-wave coherent system.
The three transducer configuration of the lithium niobate surface acoustic wave delay line 6 is a commonly used technique for reducing the insertion 1QSS which is associated with two transducer type delay lines.
Typically the insertion loss associated with conventional delay lines is reduced by 3 dub by employing this 3 transducer technique. Phase coherence between successive pulses of the transmitted signal is ensured because of the exact reproducibility of the response of the transducers 4 and 5 to the sharp pulses produced by the step recovery diode impulse generator 3.
The transmitter therefore produces the 30~ duty cycle microwave pulse train coded with buffs modulation according to the 1023 bit pseudo-random I

sequence generated by the pseudo~randomcode generator circuit 7. Between each transmitting burst a period 140 no is allowed in which pulses may be received by the receiver after reflection from the target 15. When the target is at the specified range the demodulator circuit 19 reduces all the phase coded signals received to common phase and after the auto-correlation process described for enhancing the slgnal-to-noise ratio of the signal prior to detection by a predetection integration in the multi-tap delay line 22 the output of the low-pass filter 23 and the integrator 24 will gradually build Up to a level which will eventually exceed the reference level applied to the threshold circuit 25. The output of the threshold circuit 25 is then used to detonate the SAM warhead.
It will now be appreciated that the arrangement described if the example embodiment provides not only the post-detection integration afforded, but also gives predetec-tion integration by the use of the rnulti-tap delay line 22~ The combination of post detection and predetection integration gives more processing gain than a system employing only post detection integration and is less sensitive to Doppler frequency shifts than a system employing only predetection integration. In the embodiment a 80 tap delay line was used but the number of taps used in any particular application must be carefully chosen because in general an N-bit IF pulse integrator which in effect the tap delay line is) is quite frequency son-sitive and the range of frequencies that it will successfully operate upon - YET wide to the 3 dub point, where N is the number of taps or bits to be integrated and T is the bit period. Thus in applications where Doppler frequency shifts are expected the number of taps in the predetection integrator whilst being chosen to be as high as possible to increase the predetection integration must not be so high as to render the system sensitive to Doppler frequency shifts.

It will be noted that the system described in the example makes ox-tensile use of surface acoustic wave devices. The oscillator 1, the local oscillator If and the delay lines 6 and 22 aureole surface acoustic wave devices. Because of this extensive use of surface wave acoustic devices L7S~
the size and hence weight of the radar fuzz system can be considerably less than would be possible with conventional devices due to the 10 velocity reduction of acoustic surface waves as compared with electron magnetic waves of the same frequency. Because the surface acoustic wave oscillator 1 which defines the pulse reputation frequency is based on the same substrate as the 80 tap delay line the surface acoustic wave functions are essentially insensitive to temperature changes. This is a very significant consideration and means that the substrate can be, for example, lithium nioba-te with its high coupling constant, to low coupling losses but its high temperature coefficient which normally makes its choice undesirable. Aluminum nitride on sapphire is another possibility enabling radio frequency surface acoustic wave components for the system to be fabricated without the use of an electron beam.
These devices are planar structures making them very easy to integrate into miniaturized electronic packages such as -the system described here.
Known interrupted carrier wave coherent radar systems have sense-tivities limited by inadequate suppression of the carrier during the receiver interval but the system described hereinabove has a -transmitter which is completely quiescent at the receive times and a carrier suppression of greater than 120 dub is possible.
The method of demodulation of the IF pulses in the receiver gives scope for optimum predetection integration using the surface acoustic wave tap delay line 22 as described above. In general the greater the percentage of time spent in predetection integration compared with post-detection integration the more sensitive is the receiver. The predetection integration time to the number of taps on the delay line is limited by Doppler shift in theory as described above, although in practice it is the maximum size of the surface wave substrate which can be obtained which provides the limitation.
Although a specific example has been described with respect -to a radar fuzz the system of combined predetection and post detection integration described above has application to communications. Slow data for trays-mission may be spread over a wide bandwidth which is a well-known anti-jam .. .. .. . . ..

technique in communications. The improved signal-to-noise ratio provided by the receiver in the above system is of course extremely desirable in communications systems where unknown Doppler shifts are a problem.
Many variations and modifications of -the above example will now suggest themselves to one slcilled in the art. The fixed delay line 20 which determines the activation range can be replaced with a selector for selecting different outputs of the psuedo-random code generator circuit 7 and applying them to the demodulator 19. For example if this pseudo-random code generator circuit consists of a shift register with appropriate feedback to produce the psuedo-random sequence then the selector might be arranged to select different stages of the shift register according to the desired range making use of a digital selector to address the appropriate shift register stage. This ability to derive the range function digitally is an important aid in for example altimetry where range tracking is desired. Another use of the digitally selectable range is to combat multi path problems in a communications system where a programmable correlator is needed in the receiver. These features can be incorporated in the apparatus according to the invention without the severe penalty which occurs in conventional receivers, or the loss in processing gain which occurs in the Doppler insensitive post-detection processors.
Although surface acoustic wave delay lines have been used here there is no reason in principle why they may not be replaced by electron magnetic delay lines, for example the 80 tapped delay line 22 could in principle be a tapped delay on a surface electromagnetic wave device.
Again operation can be at IF and up converted to radio frequency as described above but with improvements in surface acoustic wave technology and integrated circuits it will almost certainly be possible to eliminate the need for IF if the required radio frequency is below 2 GHz.

Claims (4)

I claim:

1. A radio signalling equipment including a receiver apparatus, for receiving signals phase-modulated by a predetermined code sequence of code signals having a predetermined regular digit-signal rate, said apparatus including, a decoding means for applying a sequence of phase shifts, com-plementary to those given by the code sequence, to the received signals;
A first signal path and a second signal path having a common input connected to an output of the decoding means;
a multi-tap delay means connected in at least one of said signal paths for progressively integrating signals in one of said paths, a time delay between adjacent taps on the multiple of the bit duration; and signal mixer means connected to receive at one input thereof the integrated signals from the multi-tap delay means, and at a second input thereof signals from the second signal path, for deriving an auto-correlation signal, the magnitude of which is dependent on the relative phases of the signals so received.

2. The radio signalling equipment defined in claim 1, wherein said multi-tap delay means comprises an acoustic delay line having a plurality of transducers equispaced along the acoustic delay line.

3. The radio signalling equipment defined in claim 2, wherein the dis-tance of spacing between any two adjacent transducers is an integral multiple of the bit duration at the appropriate acoustic wavelength.

4. The radio signalling equipment defined in claim 1, 2 or 3, wherein a transmitter apparatus is associated with the receiver apparatus, and in-cludes a pulse generator circuit for generating pulses at a selected pulse repetition frequency (PRF) of the radio signalling equipment, a surface wave acoustic delay line, two input transducers, one transducer at each end of the surface wave acoustic delay line and each connected to an output of the PRF generator, and an output transducer selectively positioned on the delay line between the input transducers.