JPH0515699U - Constant current drive circuit of step motor - Google Patents

Abstract

(57) [Abstract] [Purpose] Torque fluctuation due to voltage fluctuation in constant current drive circuit of step motor which depends on power source such as dry cell where voltage fluctuation is likely to occur. [Structure] A drive current flowing through a load L is controlled according to a target voltage. A relatively low voltage is applied as the target voltage in the first half of the time width of the pulsed command signal that switches the motor excitation phase, and a relatively high voltage is applied as the target voltage in the second half of the time width of the command signal. Since the stepwise command voltage acts as a limiter of the rising waveform of the drive current, torque fluctuation due to power supply fluctuation is suppressed.

Description

[Detailed description of the device]

[0001]

[Industrial applications]

 The present invention relates to an improvement of a constant current drive circuit of a step motor for driving the step motor with a constant current.

[0002]

[Prior Art]

 As a drive circuit for driving a device having an L component load such as a step motor, generally, a constant voltage drive system that controls so that a constant voltage is applied to the load, and a constant current flow to the load is used. The controlled constant current drive method is known, but the constant current drive method is generally superior in terms of stability against temperature changes and stability against changes in various load impedances.

For example, FIG. 8 shows an example of a conventionally known constant current drive circuit of a bipolar step motor. A coil L, which is a load, is connected to the bridge circuit of the transistors T1, T2, T3, and T4. The transistors T1 and T2 have inverters 1 and 2, and the transistors T3 and T4 have H enable control terminals. The operational amplifiers 3 and 4 are connected.

Now, when the control pulse C1 applied to the inverter 1 and the operational amplifier 4 becomes H level, the transistors T1 and T4 are turned on and the power source VB passes through the transistor T1-coil L-transistor T4-resistor 5 When the drive current I flows and conversely the control pulse C2 applied to the inverter 2 and the operational amplifier 3 becomes H level, the transistors T2 and T3 are turned on and the power source VB passes through the transistor T2-coil L-transistor T3-resistor 5 The drive current -I flows.

A constant voltage generated in the resistor 7 when a current is supplied from the constant current circuit 6 to the resistor 7 is applied as a target voltage to the positive phase inputs of the operational amplifiers 3 and 4, and the reverse phase of the operational amplifiers 3 and 4 is also applied. The terminal voltage of the detection resistor 5 connected in series with the coil L is applied to the input. Therefore, the operational amplifiers 3 and 4 control the driving current so that the voltage generated in the detection resistor 5 by the driving current flowing in the coil L becomes equal to the positive phase input of the operational amplifiers 3 and 4, and the driving current becomes a constant current. The coil L is driven with a constant current.

[0006]

[Problems to be solved by the device]

 However, when the constant current drive method as described above is adopted, it is known that the rising waveform of the drive current is delayed due to the L component of the load when the drive current rises. In the case of equipment that is supplied with a pulsed drive current, a delay in the rising waveform of the drive current is unavoidable. If the delay of the rising waveform of this drive current is also generated uniformly at all times, it will not be a factor of fluctuations such as motor torque, but the above current delay depends on the voltage of power supply voltage VB. Have sex. That is, FIG. 9 shows this principle in principle. When the pulsed command signal IN1 is switched, if the power supply voltage VB is sufficient, the drive current rises sharply as shown by the dotted curve A. However, when the power supply voltage VB is insufficient, the drive current rises slowly as shown by the dashed-dotted curve B.

Therefore, when a step motor used as a drive source of a device such as a camera-like shutter whose power source depends on a dry battery is driven with a constant current, a stable motor torque is generated due to the above voltage dependency. There was a problem that I could not get.

[0008]

[Means for Solving the Problems]

 The present invention has been made to solve such a problem, reduces the influence of voltage dependence, and has a simple circuit configuration even in the case of a device in which power supply fluctuations occur, but is sufficiently stable for practical use. It is an object of the present invention to provide a constant current drive circuit for a step motor capable of obtaining the above motor torque.

In summary, the constant current drive circuit of the step motor of the present invention has a servo system in which the drive current is controlled according to the target voltage, and a pulse for instructing the energization switching of each excitation phase of the step motor. Assuming a constant current drive circuit for a step motor that applies a target voltage to the servo system in synchronization with the pulse-shaped command signal, the absolute value is relatively low in the first half of the time width of the pulse-shaped command signal. A first voltage source means for giving a voltage to the servo system as a target voltage, and a first voltage source means for giving a voltage having a relatively high absolute value as a target voltage to the servo system in the latter half region of the time width of the pulsed command signal. The above-mentioned object is achieved by including the second voltage source means. Also, preferably, the first voltage source means is adapted to generate a voltage output containing a component corresponding to the power supply voltage in inverse proportion.

[0010]

[Action]

 According to the constant current drive circuit of the step motor of the present invention, the drive current of the step motor is controlled according to the target voltage. Therefore, if the target voltage is applied in synchronization with the pulsed command signal that commands the switching of the energization of each excitation phase of the step motor, the pulsed drive current is supplied to the step motor. The drive current of the step motor switches in synchronism with the pulsed command signal, but the rising waveform of the drive current at this time has a time delay, and the rising waveform fluctuates depending on the power supply voltage. In the present invention, the time width of the pulse-shaped command signal is divided, and the target voltage is increased stepwise in the first half area and the second half area of the divided time width. Even if there is no drive current, the drive current can follow the target voltage, and if the power supply voltage is sufficiently high, the gradually increasing target voltage acts as a limiter of the drive current. The voltage dependence is relaxed, and stable motor torque can be obtained.

[0011]

【Example】

 FIG. 1 shows an embodiment of a constant current drive circuit for a step motor according to the present invention. Elements common to those shown in FIG. 8 are designated by the same reference numerals as those shown in FIG. Redundant description will be omitted. Further, the present invention assumes a two-phase excitation type step motor, and each element of the drive circuit CH1 on the first phase side has the drive circuit CH2 on the second phase side in common. Furthermore, reference numeral 100 indicates an IC chip, and the circuit elements drawn on the outside of the IC chip 100 are external in this embodiment. Since the resistors R1, R2, R3, R4 and the resistors 17 to 22 in FIG. 6 which will be described later are merely for setting the voltage division ratio, they may be replaced with resistors in the IC for accuracy.

Reference numeral 10 in the figure denotes a voltage source that supplies a target voltage inversely proportional to the power supply voltage VB to the operational amplifiers 3 and 4. For example, as shown in FIG. 2, the power supply voltage VB is divided and input to the negative phase amplifier 10a. For example, a voltage inversely proportional to the power supply voltage VB is obtained from the negative phase amplifier 10a. Specifically, VC1 = 10b (1−Rd / Rc) −KVB · Rd / Rc, and K = Rb / (Ra + Rb), and set so that K = 10b / VBTYP when the reference power supply voltage is VBTYP. Then, the component corresponding to VB inverse proportionality can be set arbitrarily by Rd / RC. Incidentally, 10b is a constant voltage. The output voltage of the voltage source 10 is divided by the resistors R1 and R2 and added to the positive phase input of the operational amplifiers 3 and 4 as the target voltage VC1, and is divided by the resistors R1 and R2 and the same type circuit on the second phase CH2 side. It is also applied to the element as the target voltage VC2.

Reference numeral 11 is a constant voltage source that supplies a constant voltage to the operational amplifiers 3 and 4 as a target voltage. The output voltage of the constant voltage source 11 is divided by resistors R3 and R4 to obtain a positive phase input to the operational amplifiers 3 and 4. In addition to being applied as the target voltage VC1, the voltage is divided by the resistors R3 and R4 and applied as the target voltage VC2 to the same-type circuit element on the second phase CH2 side.

Between the voltage source 10 and the constant voltage source 11 and the operational amplifiers 3 and 4, there is interposed a switching means 13 which performs a switching operation by a flip-flop 12. The changeover switch 13a of the switching means 13 is a flip-flop. When the flip-flop 12 is set, the voltage dividing points of the resistors R1 and R2 are connected to the operational amplifiers 3 and 4, and when the flip-flop 12 is reset, the voltage dividing points of the resistors R3 and R4 are connected to the operational amplifiers 3 and 4. .. In addition, the changeover switch 13b on the second phase side sets the voltage dividing point of the resistors R1 and R42 when the flip-flop 12 is reset and the voltage dividing point of the resistors R3 and R4 when the flip-flop 12 is set. Each is applied as a target voltage VC2 to the same type element on the second phase side.

As shown in the time chart of FIG. 3, since the pulse-shaped command signals IN1 and IN2 are out of phase with each other by 90 °, the flip-flop 12 is set at the up edge and the down edge of the command signal IN1 to set the command. When the flip-flop 12 is reset by the up edge and down edge of the signal IN2 and the resistance value of the resistor in the voltage source 10 is appropriately selected, the voltage gradually increasing within the time width of the command signal IN1 is set as the target voltage. It can be added to the operational amplifiers 3 and 4.

Next, 14 gives control pulses C1 and C2 to the above-mentioned inverters 1 and 2 and operational amplifiers 3 and 4, and also gives control pulses C3 and C4 to the same type circuit elements on the second phase side. In the control unit, the control pulse C1 is in phase with the command signal IN1, the control pulse C2 is in phase with the command signal IN1, the control pulse C3 is in phase with the command signal IN2, and the control pulse C4 is in phase with the command signal IN2. ..

Next, the operation of the above embodiment will be described with reference to the above matters, the time chart of FIG. 3, and the characteristic curves of FIGS. 4 and 5.

First, when the power-on signal PON is turned on and the inhibit signal INH is turned off, the control unit 14 generates the control pulses C1 and C2 in synchronization with the first-phase command signal IN1 and the second-phase signal. The control pulses C3 and C4 are generated in synchronization with the command signal IN2. In the following description, the first phase CH1 side will be mainly described, but the second phase side operates in exactly the same way except that the phases are different.

The control pulse C1 is input to the inverter 1 and the operational amplifier 4, and the control pulse C2 is input to the inverter 2 and the operational amplifier 3. Therefore, in the time region in which the control pulse C1 is on, the transistors T1 and T4 become conductive, and the drive current flows from the power source VB through the transistor T1-coil L-transistor T4-resistor 5. On the contrary, in the time region where the control pulse C2 is on, the transistor T2 and the transistor T3 become conductive, and the drive current flows from the power source VB through the transistor T2-coil L-transistor T3-resistor 5.

The voltage generated in the detection resistor 5 due to the driving current flowing is fed back to the negative phase input of the operational amplifiers 3 and 4, and the operational amplifiers 3 and 4 are configured so that the negative phase input level and the positive phase input level become an imaginary short. Drive current is controlled.

Now, in the time region from the rising edge of the command signal IN1 to the rising edge of the command signal IN2 (that is, the first half region of the time width of the command signal IN1), the flip-flop 12 is set and from the rising edge of the command signal IN2. The flip-flop 12 is reset in the time region until the down edge of the command signal IN2 (that is, the latter half region of the time width of the command signal IN1).

Therefore, the switching means 13 supplies the voltage division levels of the resistors R1 and R2 as the target voltage VC1 to the operational amplifiers 3 and 4 in the first half region of the time width of the command signal IN1, and the time of the command signal IN1 is supplied. In the latter half region of the width, the voltage division level of the resistors R3 and R4 is supplied to the operational amplifiers 3 and 4 as the target voltage VC1.

Therefore, the drive current flowing through the coil L also changes following the target voltage VC1 that rises in two steps. In the time chart of FIG. 3, the drive current I1 shows the current flowing through the coil L. In FIG. 3, the drive current I1 rises at the rising edge of the control pulse C1 and the drive current I1 falls at the rising edge of the control pulse C2. What is shown is that the direction of the drive current I1 flowing through the coil L is switched with the switching of the control pulses C1 and C2.

When the target voltage VC1 is applied to the operational amplifiers 3 and 4, the drive current flowing through the coil L is controlled so that the voltage generated in the resistor 5 follows the target voltage VC1. In the case of this embodiment, Since the target voltage VC1 is gradually increased, the error range of the drive current due to the change of the power supply voltage VB is reduced, and thus the torque error is also reduced.

4 and 5 show the relationship between the target voltage VC1 applied to the positive phase inputs of the operational amplifiers 3 and 4 and the drive current I1 flowing through the coil L. Of these, FIG. 5 shows that the power supply voltage VB is sufficient. In this case, FIG. 4 shows the case where the power supply voltage VB is at the lowest level at which operation guarantee can be obtained.

Since the present invention does not solve the voltage dependence of the rising waveform of the drive current, the rising waveform I1 (H) of the drive current when the power supply voltage is sufficiently high is used when the power supply voltage is low. Although it becomes steeper than the rising waveform I1 (L) of the current, since the target voltage VC1 changes in two steps, the relatively low level of the target voltage VC1 at the rising of the driving current is the driving current I1 (H ), The driving current I1 (H) reaches a peak. Therefore, the deviation of the integrated value of the drive current I 1 due to the difference in the power supply voltage is reduced, and a stable motor torque can be obtained. In particular, when the first half region of the command signal IN1 is made to include a component corresponding to the inverse proportion to the power supply voltage VB as in this embodiment, the higher the power supply voltage VB, the faster the above limiter operates. The deviation of the integrated value of the drive current due to the fluctuation of the power supply voltage VB is reduced, and a more stable motor torque can be obtained.

Next, FIG. 6 is a circuit diagram showing another embodiment of the present invention. Elements common to the embodiment of FIG. 1 are assigned the same reference numerals as those in FIG. 1 and duplicate explanations are omitted. To do. In the embodiment shown in FIG. 6, the target voltage applied to the operational amplifiers 3 and 4 is changed by changing the voltage division ratio of the resistors.

Since the transistor 15 is turned off when the flip-flop 12 is set, the voltage divided by the resistors 17 and 18 of the output of the constant voltage source 11 is used as the target voltage VC1 and is output to the operational amplifiers 3 and 4. However, when the flip-flop 12 is reset, the transistor 15 is turned on and the resistor 19 is incorporated in parallel with the resistor 18, so that the level of the voltage point rises. Therefore, the target voltage VC1 applied to the operational amplifiers 3 and 4 is increased. Is designed to change in stages. The resistors 20, 21, and 22 and the transistor 16 correspond to the second phase side, and are exactly the same as the first phase side except that on / off is reversed.

Further, in the above, the example in which the drive current flowing through the motor is directly converted into the voltage and applied to the negative phase input of the operational amplifiers 3 and 4 has been described. For example, in the modified circuit of FIG. As shown in the figure, a transistor T5 is connected to the transistor T3, a transistor T6 is connected to the transistor T4, and current-proportional currents flowing in the transistor T5 flow in the transistor T3. A current proportional to the current flowing through the transistor T6 may flow through the transistor T4, or the current flowing through the transistors T5 and T6 may be detected by the resistor 5 and fed back to the operational amplifiers 3 and 4.

In the above description, the present invention is applied to a bipolar drive type step motor, but it goes without saying that the present invention can be applied to any constant current drive type step motor. Further, in the above, an example in which the present invention is applied to a two-phase excitation type step motor has been shown, but it goes without saying that the present invention can be applied regardless of the excitation method. Furthermore, in the above, an example was shown in which the drive current of the step motor was detected directly or indirectly to perform closed-loop control, but the drive current is controlled according to the target current. As long as the present invention can be applied to open loop control using a current mirror circuit, for example.

[0031]

[Effect of the device]

 As described above, according to the present invention, it is possible to sufficiently suppress the fluctuation range of the rising waveform of the drive current due to the fluctuation of the power supply voltage in practical use, even though the circuit configuration is extremely simple. The fluctuation range of torque can be suppressed within a practically sufficient range.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a constant current drive circuit for a step motor according to an embodiment of the present invention.

FIG. 2 is a circuit diagram showing a specific circuit of the voltage source shown in FIG.

3 is a time chart chart of the embodiment shown in FIG.

FIG. 4 is a characteristic diagram showing a relationship between a target voltage and a drive current in the embodiment shown in FIG.

5 is a characteristic diagram showing a relationship between a target voltage and a drive current in the embodiment shown in FIG.

FIG. 6 is a circuit diagram showing another embodiment of the present invention.

FIG. 7 is a circuit diagram showing a modified example of the present invention.

FIG. 8 is a circuit diagram showing an example of such a conventional circuit.

FIG. 9 is a characteristic diagram showing voltage dependence of a drive current.

[Explanation of symbols]

 L coil 3 operational amplifier 4 operational amplifier 5 detection resistance 10 voltage source 11 constant voltage source star 12 flip-flop 13 switching means R1 resistance R2 resistance R3 resistance R4 resistance R5 resistance R6 resistance R7 resistance R8 resistance

Claims (2)

[Scope of utility model registration request] 1. A target voltage is applied to the servo system in synchronism with a pulsed command signal for instructing the energization switching of each excitation phase of a step motor, the servo system having a drive current controlled corresponding to the target voltage. In a constant current drive circuit for a step motor, the first voltage source means for giving a voltage having a relatively low absolute value to the servo system as a target voltage in the first half region of the time width of the pulsed command signal. A constant current drive circuit for a step motor, comprising: and a second voltage source means for applying a voltage having a relatively high absolute value as a target voltage to the servo system in the latter half region of the time width of the pulsed command signal. 2. The constant current drive circuit for a step motor according to claim 1, wherein the first voltage source means generates a voltage output including an inversely proportional corresponding component to a power supply voltage. Current source drive circuit.