JPH05126779A - Dielectric constant detector of fuel - Google Patents

Abstract

PURPOSE:To obtain a very accurate dielectric constant detector of fuel which is suitable for mass production and enables precise control thereof even in case of sudden change of dielectric constant of the fuel. CONSTITUTION:Radio frequency rectangular wave is charged to a single-layer winding coil 4 from an amplifier 18 via a resistor 10 and radio frequency sine wave appearing at a connection point of the single-layer winding coil 4 and the resistor 10, is converted into radio frequency rectangular wave by a wave shaper 11. DC current in charged to the single-layer winding coil 4 so that duty of the converted rectangular wave may be 50%, and frequency of the radio frequency wave is controlled so as for phase difference of signals at both ends of the resistor 10 to be 0 degree.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a fuel permittivity detector for detecting the permittivity of fuel supplied to a combustor or the like in a non-contact manner to determine the property of the fuel, and is particularly used for an engine of an automobile or the like. The present invention relates to a method for measuring the alcohol content of a fuel used.

[0002]

2. Description of the Related Art In recent years, in countries such as the United States and Europe, in order to reduce oil consumption and air pollution caused by exhaust gas from automobiles, fuel mixed with alcohol in gasoline has been introduced for automobiles. It's starting. However, if such an alcohol mixed fuel is used as it is in an engine matched to the air-fuel ratio of gasoline fuel, the theoretical air-fuel ratio of alcohol is smaller than that of gasoline, so the air-fuel ratio becomes lean and operation becomes difficult. Therefore, the alcohol content rate in the alcohol-mixed fuel is detected, and the air-fuel ratio, the ignition timing, etc. are adjusted according to the detected values.

Conventionally, for detecting the alcohol content as described above, a method of detecting the dielectric constant of the alcohol-mixed fuel and a method of detecting the refractive index have been mainly proposed, and the applicant of the present invention Among these, a method for detecting the dielectric constant has been applied for in Japanese Patent Application No. 3-22488, and this method will be described below with reference to the drawings.

FIG. 5 shows a structure of a conventional dielectric constant detecting device shown in Japanese Patent Application No. 3-22488. A is a sensor portion, and 1 is formed of an insulator such as ceramic or oil resistant plastic. A cylindrical insulating tube with a bottom, through which fuel is guided,
3 is provided inside the insulating tube 1, and the outer peripheral surface of the insulating tube 1
Columnar conductive electrode 4 which is substantially parallel to the inner peripheral surface of the insulating tube 1 and coaxial with the insulating tube 1, is a single-layer coil 4a wound at a position corresponding to the conductive electrode 3 on the outer circumference of the insulating tube 1, 4a, Reference numeral 4b is a lead of the single-layer winding coil 4, and reference numeral 2 is a fuel passage formed between the inner peripheral surface of the single-layer winding coil 4 and the outer peripheral surface of the conductive electrode 3 with the insulating tube 1 interposed therebetween.

Reference numeral 5 denotes a flange to which the conductive electrode 3 is attached and which is joined to the insulating pipe 1 through the fuel seal 7 to form a fuel container as a whole, and is formed integrally with the conductive electrode 3 here. There is. Reference numeral 6 is a nipple that guides fuel to the fuel passage 2. B indicates a detection circuit unit, and 10
Is a series resistance (resistance value R _S ) connected in series to the lead 4a of the single-layer winding coil 4, and 14 is a 0 ° phase comparator connected in parallel with the series resistance 10.

Reference numeral 15 is a low pass filter connected to the output side of the phase comparator 14, and 16 is a low pass filter 12.
Output side and a predetermined reference voltage V _ref corresponding to a phase difference of 0 °
Is connected to the comparator / integrator, 17 is a voltage controlled oscillator to which the output of the comparator / integrator 16 is connected, and 18 is a voltage controlled oscillator 1
7 output amplifier, the output of which is connected to the series resistor 10. Reference numeral 19 is a frequency divider for the output frequency of the voltage controlled oscillator 17.

Next, the operation of the above conventional apparatus will be described. The sensor unit A is schematically shown in FIG. 4A or an equivalent circuit of FIG. 4B which is a more detailed version thereof, where L is the inductance of the single-layer winding coil 4, and C _f is the dielectric constant of the fuel in the fuel passage 2. Capacitance generated between the single-layer winding coil 4 and the conductive electrode 3 that changes according to ε, C _P is the stray capacitance parasitic on the lead 4a, the input capacitance of the 0 ° phase comparator 11, etc. It is a capacity unrelated to the rate ε.

When the frequency applied to the lead 4a of the sensor section A is changed, parallel LC resonance as shown in FIG. 4C is exhibited. That is, the parallel resonance frequency f _{r at} this time
Is represented by f _r = 1 / (2π√ (L × (C _P + C _S ))) = K / √ (a + b × ε) (1) Here, K, a, and b are constants determined by the shape of the sensor unit A. Since the resonance frequency f _r depends on the dielectric constant of the fuel ε as shown in equation (1), the dielectric constant of the fuel ε is the lower the atmospheric Indeed resonance frequency f _r.

For example, as a result of measurement with a predetermined sensor shape, the resonance frequency f _r of methanol having a dielectric constant ε = 33 of methanol is 7.5 MHz, and that of gasoline having a dielectric constant ε = 2 is about 9.5 MHz. there were. Further, in an arbitrary mixed fuel of methanol and gasoline, the resonance frequency f _r changes as shown in the schematic diagram (d) according to the content ratio of methanol. Therefore, by detecting the signal corresponding to the resonance frequency _fr , it is possible to detect the permittivity ε of the fuel and further the methanol content in the methanol mixed fuel.

The detection circuit section B is configured to detect the resonance frequency _fr, and the operation of the detection circuit section B will be described below. A high frequency signal is applied from the amplifier 18 to the series circuit of the resistor 10 and the single-layer winding coil 4 while the methanol mixed fuel is flowing in the fuel passage 2, and the voltage signal across the resistor 10 is applied.
That is, the high frequency voltage signal applied to this series circuit and the high frequency voltage signal applied to the single-layer winding coil 4 are the phase comparator 1
4 and the phases of both are compared.

[0011] Now, when a high frequency voltage signal of the same frequency as the resonance frequency f _r is applied to the series circuit, FIG. 4
As shown in (c), the current-voltage phase of the sensor unit A is 0 °.
Therefore, the phase difference of the high frequency voltage across the resistor 10 is 0.
Becomes °. On the other hand, when the high frequency voltage signal of a frequency lower than the resonance frequency f _r is applied to the series circuit, FIG. 4
As shown in (c), since the current-voltage phase of the sensor unit A leads from 0 °, the phase difference of the high-frequency voltage across the resistor 10 is larger than 0 ° with reference to the phase of the high-frequency signal applied to the series circuit. Becomes

Therefore, the output of the phase comparator 14 is converted into a DC voltage corresponding to the phase difference through the low pass filter 15, and this DC voltage and a DC voltage V corresponding to the phase difference of 0 °.
_ref is input to the comparison integrator 16 to integrate the difference between the two,
A phase locked loop is formed by inputting the output of the comparator / integrator 16 to the voltage controlled oscillator 17 applying a high frequency signal to the series circuit via the resistor 10.

The phase-locked loop of the voltage controlled oscillator 17 causes the phase difference between the high frequency voltage signals across the resistor 10 to be zero.
° and controlled to be the oscillation frequency of the voltage controlled oscillator 17 will always be oscillated at the resonance frequency f _r. Therefore, the output frequency of the voltage controlled oscillator 17 is divided by the frequency divider 19 into an appropriate frequency to obtain the frequency output f _out .
Further, since the oscillation frequency of the voltage controlled oscillator 17 and the control input voltage correspond integrally, the output of the comparison integrator 16 is taken _out as the voltage output V _out .

Next, a specific example of the conventional device will be described. FIG. 6 shows an exclusive OR circuit 14 as the phase comparator 14.
FIG. 7 shows a specific example of the case where a phase-locked loop is formed by using a circuit including c and a high-frequency voltage signal appearing across the resistor 10 has a phase difference of 0 °.
The timing chart of 1-P6 is shown. The high frequency rectangular wave signal P1 output from the voltage controlled oscillator 17 is supplied to the amplifier 18
Is input to the CK port of the first D flip-flop circuit 18a and the CK of the second D flip-flop circuit 18b.
A signal obtained by inverting the phase of the rectangular wave signal P1 by the inverter circuit 18c is input to the port.

In addition, the second D flip-flop circuit 18b
The signal of the inverted output port of the first D flip-flop circuit 18a is input to the D port of the first D flip-flop circuit 18a, and the signal of the output port Q of the second D flip-flop circuit 18b is input to the D port of the first D flip-flop circuit 18a. Is inputted and the signal P2 at the output port Q of the first D flip-flop circuit 18a, which is a high frequency signal applied to the single-layer winding coil 4 via the resistor 10, is data at the rising edge of the high frequency rectangular wave signal P1. The updated signal becomes a signal obtained by dividing the signal P1 by ½, and is input to the exclusive OR circuit 14c via the inverter 14a. On the other hand, the signal P3 at the output port Q of the second D flip-flop circuit 18b is equal to the signal P1. It is updated at the trailing edge and becomes a signal having the same frequency as the signal P2 but a phase difference of 90 °.

A signal P4 applied to the connection point between the resistor 10 and the single-layer winding coil 4, that is, the single-layer winding coil 4 is input to the input of the exclusive OR circuit 14c, and the signal is input to the other input. The phase of P3 is inverted and input by the inverter 14b, and the phases of both are compared. Where resistance 1
0 and the high-frequency signal P appearing at the connection point of the single-layer winding coil 4
4 is sinusoidal as shown in FIG. 7, its direct current level is determined by the operational amplifier 20 and the variable resistor 21.
By adjusting so that the determination level is 4a, the sine wave signal P4 is waveform-shaped to obtain the rectangular wave P5.

At the frequency at which the LC circuit of the sensor section A resonates, the output rectangular wave P5 of the inverter 14a has an opposite phase to the rectangular wave P2 applied to the resistor 10, and the output of the second D flip-flop circuit 18b. It becomes a rectangular wave signal whose phase is 90 ° out of phase with the signal P3 of the port Q. Therefore, the output of the exclusive OR circuit 14c is a rectangular wave with a duty of 50% when the phase difference between the signals P2 and P4 appearing at both ends of the resistor 10 is 0 °, that is, when the LC circuit of the sensor unit A resonates. It becomes P6, and the duty becomes 50% or less or 50% or more at frequencies other than the resonance frequency, and becomes a rectangular wave having a duty corresponding to the phase difference between the signals P2 and P4.

The output P6 of the exclusive OR circuit 14c is input to the low pass filter 15, and its DC output is the resistance 10
Corresponding to the phase difference between the high-frequency voltage signals P2 and P4 appearing at both ends of. The output of the low-pass filter 15 is input to the comparison integrator 16, and the variable resistor 22 causes the resistance 10
When the phase difference between the signals P2 and P4 appearing at both ends is 0 °, the difference with the voltage V _ref adjusted to be equal to the DC level output by the low-pass filter 15 is integrated, and the integrated value, that is, the comparison The output of the integrator 16 is the voltage controlled oscillator 1
7 and the oscillation frequency is controlled.

As described above, since the phase locked loop for controlling the output frequency of the voltage controlled oscillator 17 is formed so that the phase difference between the high frequency voltage signals P2 and P4 appearing across the resistor 10 becomes 0 °, the voltage control is performed. The frequency output f _{out obtained} by dividing the frequency of the oscillator 17 by the frequency divider 19 is a function that monotonically decreases with respect to the permittivity ε of the fuel, that is, the methanol content shown in FIG. Further, the output of the comparison integrator 16 input to the voltage controlled oscillator 17 is also output as the voltage output V _out .

[0020]

However, in the above-mentioned conventional apparatus, when the permittivity of the fuel suddenly changes so much that the control of the phase locked loop cannot catch up, the phases of the signals P2 and P4 are different, and as shown in FIG. Since the impedance of the LC resonance circuit becomes small, the amplitude of the signal P4 becomes small, and the decision level of the inverter 14a that shapes the waveform of the sinusoidal high frequency signal P4 and the operational amplifier 2 for the single-layer winding coil 4 are provided.
0 and the DC level given by the variable resistor 21 are slightly different, and the signal P4 does not cross the judgment level as shown in FIG. 8, and as a result, waveform shaping becomes impossible like the signal P5 of FIG. Become.

In such a state, the phase comparator 14
8, the output P6 becomes a signal with a duty of 50% simply by inverting the signal P3, and the output of the low pass filter 15 becomes the same as the output when the control of the phase locked loop is established. However, there was a problem that the control of the phase-locked loop failed and a signal different from the true permittivity of the fuel was output. Further, in the case of manufacturing a large number of this device, the DC level applied to the single-layer winding coil 4 had to be individually adjusted according to the determination level of the inverter 14a.

The present invention has been made to solve the above problems, and can restore a normal control state even when the permittivity of fuel suddenly changes so that the control of the phase-locked loop cannot catch up. It is an object of the present invention to obtain a fuel dielectric constant detection device that is suitable for mass production.

[0023]

A fuel dielectric constant detecting device according to the present invention is provided with high frequency applying means for applying a rectangular high frequency wave to a detecting coil via a resistor and a signal at a connection point between the detecting coil and the resistor. , A waveform shaper that outputs a rectangular wave by comparing with a predetermined comparison level, a phase comparator that detects the phase difference between the output of the high-frequency applying means and the output of the waveform shaper, and the output of the phase comparator has a predetermined value Control means for controlling the output frequency of the high frequency applying means, duty detecting means for detecting the duty of the output of the waveform shaper, and a DC voltage to the detecting coil so that the output of the duty detecting means becomes a predetermined value. Bias control means for supplying is provided.

[0024]

According to the present invention, the output duty of the waveform shaper for shaping the sinusoidal high-frequency voltage signal appearing at the connection point between the detection coil and the resistor into a rectangular shape is detected, and the output duty is detected so as to become a predetermined value. DC voltage is supplied to the coil.

[0025]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 shows the configuration according to this embodiment, and the sensor unit A
Is exactly the same as before. In the detection circuit section B, 11
Is a waveform shaper connected to the connection point between the single-layer winding coil 4 and the resistor 10, 12 is a low-pass filter to which the output of the waveform shaper 11 is connected, 13 is an output of the low-pass filter 12, and a waveform When the output duty of the shaper 11 is 50%, a predetermined reference voltage V _ref corresponding to the voltage output by the low-pass filter 12 is connected, and a DC level is applied to the single-layer winding coil 4 via the lead 4b. Bias control means, 1
Reference numeral 4 denotes a phase comparator to which the output side of the waveform shaper 11 and the connection side of the resistor 18 to the amplifier 18 are connected, and the other configurations are the same as those of the prior art.

Next, FIG. 2 shows a concrete example of the above-mentioned constitution, and a concrete example of the case where the phase current loop is formed so that the phase difference of the high frequency voltage signals appearing at both ends of the resistor 10 becomes 0 ° like FIG. And many parts are common. The difference from the conventional example will be mainly described below. FIG. 3 is FIG.
The signal waveform of each part of is shown.

The high frequency rectangular wave signal P1 output from the voltage controlled oscillator 17 is the first D flip-flop circuit 18a.
Of the inverter circuit 1 to the CK port of the second D flip-flop circuit 18b.
The signal whose phase is inverted at 8c is input. The Q output P2 of the first D flip-flop circuit 18a is connected to the single-layer winding coil 4 via the resistor 10, and the Q output P3 of the second D flip-flop circuit 18b has a phase difference of 90 ° with respect to the signal P2. It becomes a high frequency rectangular wave with.

The connection point between the resistor 10 and the single-layer winding coil 4, that is, the signal P4 applied to the single-layer winding coil 4 is TTL or CM.
The waveform is shaped into a rectangular wave by the waveform shaper 11 which is an inverter circuit which is one of the logic circuits such as the OS. The output is switched between the H and L levels if the input is equal to or higher than the determination level V _th. If it is less than or equal to _th , it becomes H and V _th cannot be adjusted.

Therefore, in the prior art shown in FIG. 6, the waveform was shaped by supplying a voltage corresponding to V _th to the lead 4b of the single-layer winding coil 4 by the operational amplifier 20 as the DC component of the signal P4. In the example, the output of the waveform shaper 11 is converted into a DC voltage corresponding to the duty by the low pass filter 12, and the low pass filter 12
And the voltage V _ref corresponding to the voltage output by the low-pass filter 12 when the output duty of the waveform shaper 11 is 50% is integrated by the proportional integrator 13a in the bias control means 13, The integrated value is supplied to the lead 4b of the single-layer winding coil 4.

For example, when the signal P4 is shown by the broken line in FIG. 3, the output P5 of the waveform shaper 11 is always at the H level as shown by the broken line. Therefore, the low pass filter 12
3 is always at the H level, and this H level is input to the comparator / integrator 13a. Therefore, the output of the bias control means 13 is increased, which also increases the DC level of the signal P4 and the entire waveform shown in FIG. Direction, and finally controlled to the level shown by the solid line. Therefore, the output P5 of the waveform shaper 11 is always a rectangular wave with a duty of 50% as shown by the solid line.

Exclusive OR circuit 14c of the phase comparator 14
Is formed of TTL, CMOS, or the like, and the signal P3 is input to one of the inputs via the inverter circuit 14b, the output P5 of the waveform shaper 11 is input to the other input, and both signals are compared in phase. .. When the LC circuit of the sensor unit A is excited at a frequency other than the resonance frequency, the signal P
5 and the signal P2 have a phase difference of not 0 ° but signals P5 and P3.
Since the phase difference between the two is not 90 °, the exclusive OR circuit 14
The duty of the output P6 of c is not 50%.

Therefore, the signal P6 is passed through the low pass filter 15
To the comparator / integrator 16 to integrate the difference from the reference voltage V _ref corresponding to a phase difference of 0 °, and the integrated value is input to the voltage controlled oscillator 17 to determine its oscillation frequency. When controlled, the phase of the signal P4 advances to the left in FIG. 3, and along with this, the waveform of the signal P6 also changes as shown by the arrow, and finally the phase feedback control as shown in FIG. 7 is established.

In the above embodiment, the bias control means 13 obtains the voltage V _ref corresponding to the duty of 50% by dividing the voltage _Vref from the power source by the variable resistor 23. However, other signals with the duty of 50%, for example, the signal P2. Is obtained through a low-pass filter equivalent to the low-pass filter 12, and the DC voltage is supplied to the single-layer winding coil 4 by the bias control means 13 so that this voltage becomes equal to the output voltage of the low-pass filter 12. The voltage applied may be controlled. Further, although the present apparatus was used to detect the methanol content in methanol-blended gasoline, it can be widely used to detect the dielectric constant in other liquids.

[0034]

As described above, according to the present invention, the phase difference between the voltages across the resistor connected in series with the detection coil is detected, and the phase difference is detected through the resistor so that the phase difference becomes a predetermined value. Controls the output frequency of the high-frequency applying unit that applies a high frequency to the coil, and shapes the voltage at the connection point between the detection coil and the resistor into a rectangular wave with a waveform shaper when detecting the phase difference, and the duty of this rectangular wave is a predetermined value. A DC voltage is applied to the detection coil so that the output duty of the waveform shaper is controlled to a predetermined value, and the permittivity of the fuel suddenly changes so that the control of the phase-locked loop cannot catch up, and the phase difference is detected. Even if the normal control is no longer performed, the normal control state can be quickly recovered, and the accuracy of the dielectric constant detection can be improved. Further, even when a large amount of this device is manufactured, it is not necessary to adjust the DC voltage supplied to the detection coil in accordance with each characteristic of the waveform shaper, and a device suitable for mass production can be obtained.

[Brief description of drawings]

FIG. 1 is a block diagram of an apparatus according to the present invention.

FIG. 2 is a more specific configuration diagram of the device of the present invention.

FIG. 3 is a time chart showing operation waveforms of the configuration of FIG.

FIG. 4 is an equivalent circuit diagram and a frequency characteristic diagram of a sensor unit of a conventional device.

FIG. 5 is a configuration diagram of a conventional device.

FIG. 6 is a more specific configuration diagram of a conventional device.

FIG. 7 is a time chart showing the operation of the configuration of FIG.

FIG. 8 is an explanatory diagram of an output error of the conventional device.

[Explanation of symbols]

 A sensor section 1 insulating tube 2 fuel passage 3 conductive electrode 4 single layer winding coil B detection circuit section 10 resistance 11 waveform shaper 12 low pass filter 13 bias control means 14 phase comparator 16 comparator / integrator 17 voltage controlled oscillator

Claims (1)

[Claims] 1. A detector provided in a fuel passage, the capacity of which changes in accordance with the dielectric constant of the fuel, a detector coil connected substantially in parallel with the output of the detector, and a detector coil connected in series. A resistor, a high-frequency applying means for applying a rectangular high-frequency wave to the detection coil via this resistor, and a signal at the connection point between the detection coil and the resistor are input, and a waveform shaping that outputs a rectangular wave in comparison with a predetermined comparison level. A phase comparator that detects a phase difference between the output of the high frequency applying means and the output of the waveform shaper, and a control means that controls the output frequency of the high frequency applying means so that the output of the phase comparator has a predetermined value. An output rectangle of the waveform shaper; a duty detecting means for detecting the duty of a high frequency wave; and a bias control for applying a DC voltage to the side of the detecting coil to which the resistor is not connected so that the output of the duty detecting means becomes a predetermined value. A fuel dielectric constant detecting device comprising a control means, and detecting the dielectric constant of the fuel by the output frequency of the high frequency applying means or the control output of the control means.