JPH04359889A - Power supply for electronic oven - Google Patents

Abstract

PURPOSE:To stably control an output power of a magnetron by a simple constitution. CONSTITUTION:A half bridge converter, containing a high frequency transformer 34 comprising a series connection of switching elements 20a and 20b, a series connection of resonance condensers 26a and 26b, a primary winding 34a, and a secondary winding 34b, is driven by a 200V commercial power supply 14. A current transformer 44 is coupled to a current path containing a connecting wire 22 for connecting the switching elements 20a and 20b so as to detect a current in the primary winding 34a. A control circuit 12 compares the voltage equivalent to an average value of a current detected by the current transformer 44 with the voltage set by a microcomputer, thereby switching elements 20a and 20b are controlled by the frequency proportional to the differential voltage therebetween A high frequency high voltage induced in the secondary winding 34b is applied in a magnetron 36 via a voltage doubler rectifier circuit.

Description

[Detailed description of the invention]

0001

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply device for a microwave oven. More particularly, the present invention relates to a novel microwave oven power supply device suitable for relatively high voltage commercial power supplies, such as 200V or 220V.

0002

[Prior Art] An example of an inverter circuit that can be used as a power supply device for a microwave oven was released on March 1, 1977.
Japanese Unexamined Patent Publication No. 52-35903 (H
05B6/66) or Japanese Patent Application Laid-open No. 191290 (H05B6/68) published on October 30, 1984. This conventional technology is a power supply device using a voltage resonance inverter.
The resonance time constant is determined by the leakage inductance of the high frequency transformer, the inductance of the primary and secondary windings of the high frequency transformer, and the resonant capacitor. In such a voltage resonance type inverter circuit, fluctuations in the input voltage of the main circuit (power supply voltage of the commercial power supply) directly appear as fluctuations in the resonant voltage. Therefore, as the switching elements constituting the inverter, it is necessary to use switching elements with a large withstand voltage in anticipation of such fluctuations in the resonance voltage. For example, if the input voltage is 100V, the peak voltage is about 141V (=100√2), and a resonant voltage of about 600V is applied to the switching element at the rated input. Therefore, since the resonant voltage varies according to the variation of the input voltage, in this case, for example, 900V
A switching element with a rating of 50A is used.

On the other hand, recently, it has been proposed to use a 200V commercial power source to increase the capacity and speed of cooking appliances such as microwave ovens. When the input voltage of the above-mentioned inverter circuit becomes 200V, a switching element having a rating of about 1800V30A is required. However, a switching element that operates in a high frequency range and has such a high breakdown voltage in a power supply device for a microwave oven has not yet been put into practical use. Therefore, this JP-A-52-3
The inverter circuit disclosed in Japanese Patent Laid-open No. 5903 or Japanese Patent Laid-Open No. 59-191290 cannot be used as a power supply device for a microwave oven driven by a 200V commercial power source.

[0004] On the other hand, Japanese Patent Laid-Open No. 101962 (H02M) published on April 13, 1990
3/28, 3/335) includes a series circuit of two switching elements, a series circuit of two resonant capacitors, and a series circuit of two switching elements.
A series circuit of two feedback diodes is connected to a DC power supply, and two
A half-bridge converter is disclosed in which a series circuit including a primary winding of a transformer and a first resonant inductor is connected between a connection point between two switching elements and a connection point between two resonant capacitors. This conventional half-bridge converter is mainly used in a stabilized power supply circuit, and such a half-bridge converter is not currently used as a power supply device for a microwave oven.

[0005]

[Problem to be solved by the invention] JP-A-2-10196
Although it is conceivable to use the half-bridge converter disclosed in Publication No. 2 as a power supply device for a microwave oven, this conventional half-bridge converter cannot be used as it is as a power supply device for a microwave oven. That is,
In the conventional technology described above, in order to control the output voltage of the half-bridge converter, the switching frequency of the two switching elements is controlled by the error voltage between the output voltage of the load and the reference value, but the magnetron used in the microwave oven To control the output power of the magnetron, since it is a constant voltage load to the bridge converter,
What is necessary is to detect the output current. In this case, a current transformer that detects the output current of the magnetron is provided on the secondary side of the high-frequency transformer, and the output of the current transformer is input to the control circuit. In order to isolate the next side and the control circuit, a current transformer with a large creepage distance and spatial distance is required. Therefore, the power supply device becomes large and expensive. Furthermore, in this conventional half-bridge converter, a resonant inductor is connected in series to the primary winding, but in order to obtain a power supply for a microwave oven with a magnetron output power of approximately 800 W at a resonant frequency of 50 kHz, an inductance of approximately 10 μH is required. It is necessary to use a large resonant inductor with a Therefore, if the half-bridge converter disclosed in Japanese Patent Application Laid-Open No. 2-101962 is used as is, the microwave oven power supply device will become large and expensive.

[0006] Therefore, the main object of the present invention is to provide a new power supply device for a microwave oven. Another object of the present invention is to provide a power supply device for a microwave oven that uses a half-bridge converter and can be made small and inexpensive.

[0007]

[Means for Solving the Problems] Simply put, the present invention provides a first series connection of first and second switching elements that are driven by a commercial power source, and a first series connection of first and second switching elements connected in parallel to the first series connection. a second series connection of the first and second resonant capacitors, and a primary winding connected between the connection point of the first series connection and the connection point of the second series connection; Magnetically coupled to supply voltage to magnetron 2
A half-bridge converter including a high-frequency transformer having a secondary winding, a set voltage output means for outputting a set voltage that commands the desired output power of the magnetron, a current detection means for detecting the current flowing in the primary winding, and a current detection means averaging means for averaging the current detected by the averaging means; error voltage output means for outputting an error voltage corresponding to the difference between the voltage obtained by the averaging means and the set voltage; This is a power supply device for a microwave oven, including switching control means for controlling first and second switching elements.

[0008]

[Operation] Commercial power is applied as input voltage to the half-bridge converter through a diode bridge. For example, bipolar transistors, field effect transistors or IGBTs (Insulated Gate
BipolarTransistor)
and the second switching element are given first and second switching signals having mutually different phases from the switching control means, so that the first and second switching elements are alternately turned on or off, and the half bridge The converter oscillates with a resonant time constant determined by the primary leakage inductance of the high frequency transformer and the first and second resonant capacitors. The first and second switching elements are connected in series by a connecting line, and a current transformer functioning as a current detecting means is magnetically coupled to the connecting line. This current transformer is connected to the first and second
The current flowing through the switching element, that is, the current flowing through the primary winding is detected. The output of the current transformer is inputted to an averaging means such as a bandpass filter or a lowpass filter via a diode bridge, for example. The voltage has a magnitude corresponding to the difference between a voltage having a level corresponding to the average current voltage obtained from the averaging means and a set voltage that commands a desired output voltage of the magnetron from a set voltage output means such as a microcomputer. The error voltage is output from the error voltage output means. The switching control means includes, for example, a V/F conversion means, and outputs first and second switching signals having a frequency proportional to the magnitude of the error voltage but constant on-time,
and the second switching element.

[0009]

According to the present invention, the output power of a half-bridge converter, that is, a magnetron, can be feedback-controlled based on the current detected by the current detection means. Therefore, compared to the case where the half-bridge converter disclosed in JP-A-2-101962 is used as is, a smaller and cheaper power supply device for a microwave oven can be obtained.

The above objects and other objects, features, aspects and advantages of the present invention will become more apparent from the following detailed description of the embodiments, taken in conjunction with the accompanying drawings.

[0011]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 is a circuit diagram showing an embodiment of the present invention. The microwave oven power supply device 10 of this embodiment includes a main circuit controlled by a control circuit 12, which is mainly constituted by a half-bridge converter, and receives 200V AC power from a commercial power supply 14. 200V
The alternating current is full-wave rectified by a diode bridge 16, and a half-bridge converter is connected to the output of the diode bridge 16 via a normal mode choke coil 18.

That is, a series connection of switching elements 20a and 20b is connected to the output of the diode bridge 16 via a normal mode choke coil 18. In this embodiment, IGBTs are used as switching elements 20a and 20b, respectively. This 2
The two switching elements 20a and 20b are connected to the connection line 22.
connected in series by A series connection of two discharge resistors 24a and 24b and a series connection of two resonant capacitors 26a and 26b are connected in parallel to the series connection of switching elements 20a and 20b, respectively. At the same time, a ringing choke converter 28 that generates power for the control circuit 12 is connected to the output of the diode bridge 16 . A diode 30a is connected in parallel to the switching element 20a, and a series connection of a diode 30b and a current detection resistor 32 is connected in parallel to the switching element 20b.

Two switching elements 20a and 20
The primary winding 3 of the high frequency transformer 34 is connected between the series connection point of the resonant capacitors 26a and 26b and the series connection point of the resonant capacitors 26a and 26b.
4a is connected. Two secondary windings 34b and 34c are magnetically coupled to this primary winding 34a so that the energy stored in the primary winding 34a is transmitted. Note that the diodes 30a and 30b described above are connected to the primary winding 3.
It functions to return to the commercial power supply 14 the energy stored in the coil 4a but not transmitted to the secondary windings 34b and 34c. The secondary winding 34b supplies high voltage between the heater or cathode and anode of the magnetron 36,
Secondary winding 34c supplies heater voltage to the cathode of magnetron 36. That is, a voltage doubler rectifier circuit consisting of a diode 38 and a capacitor 40 is connected to the secondary winding 34b, and the output voltage of this voltage doubler rectifier circuit is applied between the cathode and anode of the magnetron 36. Further, the heater of the magnetron 36 is connected to the secondary winding 34c through a high frequency filter 36a built in the magnetron 36.

A potential transformer 42 is connected to the input of the commercial power supply 14 , that is, the diode bridge 16 , and the output of the potential transformer 42 is given to the control circuit 12 . Further, as shown in FIG. 1, a current transformer 44 is magnetically coupled to the once-wound connecting wire 22, and the output of this current transformer 44 is also given to the control circuit 12. By winding the connecting wire 22 as shown in FIG. 1, the current transformer 44 can individually detect the current flowing through the switching element 20a or the switching element 20b, and the current flowing through the switching elements 20a and 20b can be superimposed. The total current generated can also be detected. That is, the current of the switching element 20a flowing in a part of the path of the connection wire 22 is in the same direction as the current of the switching element 20b flowing between the primary winding 34a and the switching element 20b, so that the current transformer 44 is When commonly magnetically coupled to the two current path sections, current transformer 44 can detect the total current flowing through switching elements 20a and 20b. Moreover, the switching element 20a
and 20b are turned on alternately, so that the current transformer 44 can detect individual currents during each on period.

In the embodiment shown in FIG. 1, a cooling fan 46 is disposed close to the magnetron 36, and this cooling fan 46 is driven or stopped by a fan driver 48. The main circuit of the microwave oven power supply device 10 of the embodiment shown in FIG.
It includes a half-bridge converter similar to that disclosed in Publication No. 101962. That is, the switching elements 20a and 20b are alternately turned on or off by the switching signals Q1 and Q2 from the control circuit 12, so that the primary winding 34a of the high frequency transformer 34
A high frequency current as shown by the solid line in FIG. 2 or 3 is generated. This high frequency current is multiplied by the high frequency transformer 34b, and a high frequency high voltage is induced in the secondary winding 34b. The high frequency high voltage induced in the secondary winding 34b is converted into a DC high voltage by the voltage doubler rectifier circuit, and the DC high voltage is applied between the cathode and anode of the magnetron 36, thereby causing the magnetron 36 to oscillate. do.

Note that in a steady state, that is, in a normal oscillation state, the primary winding 34a of the high frequency transformer 34 is
Alternatively, a primary current as shown by the solid line in FIG. 3 flows. Further, a current flows through the resonant capacitor 26a or 26b as indicated by the dashed line in FIG. 2, and a voltage as indicated by the dashed dotted line in FIG. 2 is generated in the resonant capacitors 26a and 26b. The voltage of the primary winding 34a of the high frequency transformer 34 is shown by the dotted line in FIG.

The control circuit 12 shown in FIG. 4 includes a microcomputer 50, which controls the cooking time, the output power of the magnetron 36, etc. As shown by the solid line in FIG. 2 or 3, the high frequency transformer 34
A primary current flows through the primary winding 34a, and this primary current is detected by the current transformer 44 shown in FIG. The output of the current transformer 44 is applied to a diode bridge 52, and the output of the diode bridge 52 is applied to the (-) input of a comparator 56 through a bandpass filter 54. The bandpass filter 54 extracts harmonic components of the current detected by the current transformer 44. In this embodiment, the pass band of the bandpass filter 54 is set to 120 Hz, which is twice the frequency of the commercial power supply 14 shown in FIG. 1, for example, 60 Hz. That is, the bandpass filter 54 includes switching elements 20a and 20
The second harmonic component of the power supply frequency is extracted from the current of b, that is, the current of the primary winding 34a. The output of the bandpass filter 54 is averaged by a smoothing circuit 55 consisting of a resistor and a capacitor. The frequency spectrum of the output of the current transformer 44 is shown in FIG. 5, and the 120 Hz component having the maximum amplitude in FIG. It has a correlation with electric current. Therefore, the harmonic components of the output of the current transformer 44 are transferred to the bandpass filter 5.
4 and averaged by the smoothing circuit 55, the input power of the half-bridge converter can be equivalently detected. In this embodiment, the half-bridge converter is feedback-controlled in accordance with the output of the current transformer 44.

In this way, the bandpass filter 54
The harmonic components of the commercial power supply 14 are extracted and averaged by the smoothing circuit 55, and a voltage corresponding to the average current is applied from the smoothing circuit 55 to the (-) input of the comparator 56. The same (-) input of the comparator 56 is given a set voltage Vs corresponding to the output power of the magnetron 36 that matches the cooking conditions set by the user from the microcomputer 50, and the (+) input of the comparator 56 is given ) input has reference voltage 57
is input. Therefore, the comparator 56 outputs an error voltage corresponding to the difference between the output voltage from the smoothing circuit 55 and the set voltage Vs. This error voltage is applied to the V/F conversion circuit 60 through the soft start circuit 58. In this embodiment, an integrated circuit "GP605" manufactured by GENNUM is used as the V/F conversion circuit 60. A functional block diagram of this integrated circuit "GP605" is shown in FIG.

The integrated circuit "GP605" is an integrated circuit developed for switching control of high frequency power supply equipment, and when a voltage is applied to the 13th terminal in FIG. 7, from the 8th terminal and the 6th terminal, Two switching control signals are obtained with frequencies proportional to the magnitude of the applied voltage. Therefore, in this example, the integrated circuit “GP60
The output voltage of the comparator 56 is applied to the 13th terminal of the 5''.The two switching control signals QA and QB output from the 8th and 6th terminals are shown in FIG. The on-time TON of both switching control signals QA and QB is constant, and the frequency is proportional to the magnitude of the input voltage, that is, the output voltage of the comparator 56, as shown in the graph of FIG.

Note that this integrated circuit "GP605" outputs two switching control signals having a phase difference of 180 degrees from each other as shown in FIG. 8 when the signal SEO applied to the 10th terminal is at a low level. However, the signal SE
When O is at high level, the two switching control signals are in phase. A switching control signal QA as shown in FIG. 8 from the V/F conversion circuit 60 is given to the driver 62 via an AND gate 61, and the switching control signal Q
B is given to the driver 62 as is. driver 62
Although the details are not shown, the switching control signal QA
and outputs switching signals Q1 and Q2 of a predetermined voltage, for example 16V, in synchronization with QB. As described above, the switching signals Q1 and Q2 are applied to the gates of switching elements 20a and 20b, respectively. The driver 62 is an IR (Inter
A high-side/low-side switch, such as the integrated circuit "IR2110" manufactured by International Rectifier Corporation, may be used. However, as the driver 62, a circuit configuration as shown in FIG. 10 may be used.

Driver 62 shown in FIG. 10 includes photocouplers 621 and 622 driven by switching control signals QA and QB, respectively, and the outputs of photocouplers 621 and 622 are connected to switching signal Q through inverting buffer amplifiers 623 and 624, respectively.
1 and Q2. That is, when the switching control signal QA or QB is at a high level, that is, during the off period, the photocoupler 621 or 622 is not driven, and when the switching control signal QA or QB is at a low level, that is, during the on period TON, the photocoupler 621 or 622 is not driven.
1 or 622 is driven. When the photocoupler 621 or 622 is driven, the input of the inverting buffer amplifier 623 or 624 becomes low level, which is output via the inverting buffer amplifier 623 or 624 as a high level switching signal Q1 or Q2. The switching elements 20a and 20b are turned on alternately by the switching signals Q1 and Q2, and the half-bridge converter performs an oscillating operation, but since the detailed operation is well known, a detailed explanation thereof will be omitted.

The soft start circuit 58 shown in FIG.
This is a circuit for gradually increasing the input voltage of the /F conversion circuit 60. Although the aforementioned integrated circuit "GP605" also incorporates a soft start circuit, the delay time is short and its operation is different from the operation described later, so in this embodiment,
The built-in soft-start circuit is not used. The soft start circuit 58 gradually increases the oscillation frequency of the switching elements 20a and 20b after the magnetron 36 starts oscillating.
Gradually increase the output power of. Specifically, as shown in FIG. 11, the soft start circuit 58 includes a diode 581.
and a series connection of a resistor 582 and a diode 581
and a transistor 583 at the series connection point of the resistor 582.
A resistor 584 is connected to the emitter of the transistor 583, and a capacitor 585 is connected to the collector of the transistor 583. Therefore, the error voltage from comparator 56 is applied through resistor 584 to the emitter of transistor 583. Then, when the error voltage reaches a predetermined base voltage, the transistor 583 is turned on and the capacitor 585 is charged with the error voltage. Therefore, the error voltage from comparator 56 is delayed according to a time constant determined by resistor 584 and capacitor 585, and V/F
The signal is applied to the conversion circuit 60.

A transistor 586 is connected in parallel to the capacitor 585, and a control signal from the activation control circuit 66 (FIG. 2) is applied to the base of this transistor. Since this control signal is at a high level at startup, the collector of transistor 583 is grounded by transistor 586. Therefore, at startup, a voltage of approximately 0V is applied to the V/F conversion circuit 60, the frequency of the switching control signals QA and QB, that is, the switching signals Q1 and Q2 is low, and therefore the primary winding 34a (
The current in FIG. 1) is small, and the high frequency voltage induced in the secondary winding 34b is minimized.

Thereafter, when the magnetron 36 has started up sufficiently, the transistor 58 is activated from the startup control circuit 66.
The control signal given to the base of 6 becomes low level,
Since the transistor 586 is turned off, the error voltage from the comparator 56 passes directly through the soft start circuit 58 to V
/F conversion circuit 60. Note that the diode 587 in FIG. 11 forms a discharge path for the capacitor 585.

In such a steady state, the primary current of the primary winding 34a is detected by the current transformer 44, harmonic components are extracted by the bandpass filter 54, and averaged by the smoothing circuit 55. The current output power of the magnetron 36 is determined by the output power of the magnetron 36 commanded by the set voltage Vs from the microcomputer 50.
When the output power is larger, the output voltage (average value voltage) of the smoothing circuit 55 becomes larger than the set voltage Vs, so the error voltage from the comparator 56 becomes smaller. Therefore, the switching control signals QA and QB from the V/F conversion circuit 60 have relatively low frequencies as shown in FIG. 9, and the frequencies of the switching signals Q1 and Q2 also become low. Therefore, the off time of switching elements 20a and 20b becomes longer, and the oscillation frequency of the half-bridge converter becomes lower, so that the primary current also becomes smaller, and the output power of magnetron 36 becomes smaller. Conversely, when the current output power of the magnetron 36 is smaller than the output power of the magnetron 36 commanded by the set voltage Vs from the microcomputer 50, the output voltage of the smoothing circuit 55 becomes smaller than the set voltage Vs. , the error voltage from comparator 56 increases. For this purpose, the V/F conversion circuit 60 outputs relatively high frequency switching control signals QA and QB. Therefore, the oscillation frequency of the half-bridge converter increases, and the primary current increases. In this manner, in this embodiment, the magnitude of the primary current of the primary winding 34a of the high frequency transformer 34 is detected by the current transformer 44, and a current feedback loop is configured to perform power control. Therefore, power control, which was difficult with the circuit disclosed in the above-mentioned Japanese Patent Application Laid-Open No. 2-101962, becomes possible.

Referring to FIG. 4, the output voltage of potential transformer 42 shown in FIG. 1 is applied to diode bridge 64. The outputs of the two diode bridges 52 and 64 are both provided to a startup control circuit 66. This startup control circuit 66 performs an intermittent oscillation operation by the switching element 20a until the magnetron 36 starts oscillating operation, operates the soft start circuit 58 after the magnetron 36 oscillates, and changes from the above-mentioned intermittent oscillation operation to normal oscillation operation. to be transferred to Specifically, the startup control circuit 66 includes a diode bridge 64 as shown in FIG.
The comparator 661 receives the output of the comparator 661 at one input thereof, and the other input of the comparator 661 is provided with a reference voltage 662. The output of the comparator 661 is then inverted and applied to the other input of the AND gate 61. Comparator 661
A transistor 663 is connected between the output and ground. Also, the output of the diode bridge 52 is connected to the comparator 6.
A reference voltage 665 is applied to the other input of the comparator 664. The output of comparator 664 is provided as a trigger input to monostable multivibrator 666, and the output of this monostable multivibrator 666 is provided to the reset input of RS flip-flop 667. A start command signal from the microcomputer 50 (FIG. 4) is applied to the set input of the RS flip-flop 667. The inverted output of the RS flip-flop 667 is applied to the base of the aforementioned transistor 663. Note that this inverted output is also given to the fan driver 48 shown in FIG.

Comparator 661 is potential transformer 42
The comparator 664 determines whether the voltage of the current transformer 44, that is, the input voltage of the main circuit, exceeds a certain value, and the comparator 664 determines whether the output of the current transformer 44, that is, the primary current, exceeds a certain value. That is, at startup, the RS flip-flop 667 is set by a command signal from the microcomputer 50, so its inverted output becomes a low level. Therefore, the transistor 663 is turned off, and the output voltage from the diode bridge 64, that is, the commercial power supply 1
4 exceeds a certain value, the output of the comparator 661, that is, the input of the AND gate 61 becomes low level, so that the switching control signal QA from the V/F conversion circuit 60 is not provided to the driver 62. Therefore, at startup, the switching signal Q1 from the driver 62 is not output, and only the switching signal Q2 is output. Therefore, one switching element 20a constituting the half-bridge converter is turned off when the input voltage of the main circuit reaches a predetermined value or more during startup, and the resonance operation by this switching element 20a is stopped. Therefore, the secondary windings 34b and 34 of the high frequency transformer 34
In c, as shown in FIG. 13, no high frequency high voltage is induced while the input voltage of the main circuit exceeds a certain level. Therefore, at startup, the secondary winding 34 of the high frequency transformer 34
The voltage appearing at b can be reduced.

When the magnetron 36 starts oscillating, the primary current increases, and when the output voltage of the diode bridge 52 exceeds the reference voltage 665, the output of the comparator 664 becomes high level, thereby causing the monostable multivibrator 6
66 is triggered and accordingly the RS flip-flop 667
is reset, and its inverted output becomes high level. Therefore, transistor 663 is turned on and the input of AND gate 61 is fixed at a high level. As a result, the switching control signal QA is directly applied to the driver 62, and the switching elements 20a and 20b perform normal oscillation operations.

Assuming that the reference voltage 662 of the comparator 661 is set to a voltage corresponding to 220V of the commercial power supply 14, 2
The output voltage of the next winding 34b is approximately 8800V (=220×2
0x2 (where "20" is the turns ratio of the high frequency transformer 34). Therefore, the surge voltage that occurs when the magnetron 36 (FIG. 1) is not oscillating can be reduced, and therefore the withstand voltage of the high frequency transformer 34, diode 38, and capacitor 40 can be reduced.

If such a starting control circuit 66 is not used, when the magnetron 36 is not oscillating, the 1
A voltage of 1200V (=280x20x2) is applied to the secondary winding 3.
Occurs in 4b. This voltage is the rating of the magnetron 36 (1
0kV). Therefore, the startup control circuit 66
If this is not used, the rating of the magnetron 36 will have to be increased, and the dielectric strength of other parts will also have to be increased. however,
Since the high voltage at startup can be suppressed by the startup control circuit 66, the dielectric strength of the high frequency transformer 34 and the like can be reduced, resulting in a less expensive microwave oven power supply device.

When the embodiment of FIG. 12 is used, the secondary winding 3
Since an intermittent voltage as shown in FIG. 13 is also induced in the secondary winding 34c, the heating time of the heater heated by the voltage of the secondary winding 34c becomes longer.
6 will take longer to start up. Therefore, in order to deal with this new problem, in this embodiment, the fan driver 48 (FIG. 1) is controlled by the output of the RS flip-flop. That is, a start command signal from the microcomputer 50 is applied to the set input of the RS flip-flop 667, and therefore, in response to this signal, the inverted output of the RS flip-flop 667 becomes low level. While the inverted output of the RS flip-flop 667 is at a low level, that is, at startup, the fan driver 48 stops the cooling fan 46. Therefore, even if the heater voltage of the magnetron 36 is small, it can be sufficiently heated relatively quickly. Therefore, by intermittently operating the half-bridge converter at startup, it is possible to prevent the startup time from increasing.

In the embodiment shown in FIG. 14, the switching control signal QA from the V/F conversion circuit 60 is also applied to the toggle flip-flop 76, so that the output of the toggle flip-flop 76 is switched as shown in FIG. It becomes high level or low level for each control signal QA. This toggle flip-flop 76 is given, for example, the output (or its inversion) of the comparator 661 shown in FIG. Ru. The output of this toggle flip-flop 76 is applied as a switching signal to the 10th terminal of the integrated circuit "GP605" shown in detail in FIG.

During the period when the switching signal is at high level, V/F
The conversion circuit 60, ie, the integrated circuit "GP605" operates in single mode rather than dual mode. Therefore, during this high level period, switching control signals QA and QB become in-phase signals as shown in FIG. 15. Therefore, switching elements 20a and 20b
are turned on at the same time, and the diode bridge 16, common mode choke coil 18 and switching element 20a
Current flows through the paths 20b and 20b, and energy is stored in the common mode choke coil 18. Then, in the subsequent low level period, the switching control signal QA
The switching element 20a is turned on in synchronization with the switching control signal QB, and the switching element 20a is turned on in synchronization with the switching control signal QB.
Since 0b is turned off, the output of the diode bridge 16 and the energy stored in the common mode choke coil 18 are provided as the input voltage of the half-bridge converter. Even if an input voltage is applied such that the voltage applied to the magnetron 36 becomes less than a predetermined value (for example, 3.8 kV to 4.0 kV), no current flows through the magnetron 36, and the current flows from the primary side of the high frequency transformer 34 to the secondary side. No energy is transferred to the side. However, as described above, the energy stored in the common mode choke coil 18 increases the input voltage, and the start and end of oscillation of the magnetron 36 are advanced and delayed as shown by dotted lines in FIG. Therefore, the primary current increases, and according to the embodiment shown in FIG. 14, an improvement in the power factor can be expected.

FIG. 17 shows a detection circuit 78 for detecting breakdown of the diode 38 constituting the voltage doubler rectifier circuit connected to the secondary winding 34b of the high frequency transformer 34.
is shown. This detection circuit 78 includes a comparator 80, and the output from the diode bridge 52 shown in FIG.
0 is applied to one input of the comparator 80, and a reference voltage is applied to the other input of the comparator 80. Therefore, the comparator 80 detects whether the output of the diode bridge 52 exceeds a certain level. The output of this comparator 80 is applied to a trigger input (fifth terminal) of a retriggerable monostable multivibrator 82. By the action of the AND gate 84, the retriggerable monostable multivibrator 82 outputs a continuous signal at a high level or a low level when a subsequent trigger input is applied within a certain period of time T1. The above-mentioned fixed time T1 is determined by the resistor R and capacitor C shown in FIG.

When the diode 38 is in a normal state, a primary current shown in FIG. 18(A) flows through the primary winding 34a of the high frequency transformer 34. Therefore, diode bridge 5
The output of No. 2 becomes as shown in FIG. 18(B), and exceeds the reference voltage only when the primary current is a positive half wave. therefore,
The output of the comparator 80 is as shown in FIG. 18(C), and the period of high level is longer than the fixed time T1. Therefore, the ninth retriggerable monostable multivibrator 80
The output from the number terminal becomes a high level continuous signal as shown in FIG. 18(G). In addition, FIG. 18(E) and FIG.
8(F) indicates one input and output of the AND gate 84, respectively.

When the diode 38 is broken, a primary current shown in FIG. 19A flows through the primary winding 34a of the high frequency transformer 34. Therefore, the output of the diode bridge 52 becomes as shown in FIG. 19(B), and exceeds the reference voltage every half wave of the primary current. Therefore, comparator 8
The output of 0 is as shown in FIG. 19(C), and the period of high level is shorter than the fixed time T1. Therefore, the output from the ninth terminal of the retriggerable monostable multivibrator 80 becomes a low level continuous signal as shown in FIG. 19(G). In addition, FIG. 19(E) and FIG. 19(F)
represent one input and output of the AND gate 84, respectively. A low-level continuous signal from the retriggerable monostable multivibrator 80 is given as an abnormality detection signal to, for example, the microcomputer 50 (FIG. 4).

In place of the detection circuit 78 shown in FIG.
The circuit shown in 0 may also be used. In the embodiment shown in FIG. 20, the terminal voltage of the detection resistor 32 shown in FIG.
is given to one input. A reference voltage 88 is applied to the other input of the comparator 86, and the comparator 86 operates as shown in FIG.
Similarly to the comparator 80, the current level of the negative half wave of the primary current is detected. Therefore, as shown in FIG. 19A, when the current level of the negative half wave of the primary current exceeds a certain value due to a failure of the diode 38, a detection signal is output from the comparator 86.

In the embodiment shown in FIG. 21, the magnetron 3
The heater No. 6 is heated not only by the output of the secondary winding 34c of the high frequency transformer 34 but also by the output of the ringing choke converter 28 provided through the ring core 90. That is, the output line of the ringing choke converter 28 shown in detail in FIG. 22 is wound around the ring core 90. The ring core 90 has another winding connected to the cathode of the magnetron 36 through a diode 92 . Therefore, the output of the ringing choke converter 28 induced through the ring core 90 is applied to the heater together with the output of the secondary winding 34c. However, the heater current of the magnetron 36 may be supplied only from the ringing choke converter 28. Note that the capacitor 94 is a smoothing capacitor. In addition, the ringing choke converter 28
As shown in FIG. 22, the other outputs are used as a power source for the control circuit 12 (FIG. 1 or FIG. 4) and as a voltage source for the driver 62 (FIG. 4), respectively.

According to the embodiment shown in FIG. 21, the following effects can be expected. That is, as shown in FIG. 1 or FIG. 21, the magnetron 36 has a built-in high frequency filter 36a, and the impedance of the high frequency filter 36a is 2πfL (L is the inductance of the high frequency filter).
It changes with Therefore, since the oscillation frequency of the half-bridge converter is changed to control the output power of the magnetron 36, it is difficult to always flow a constant heater current. Therefore, if the heater current is also supplied from another power source that is essential to the half-bridge converter, such as the ringing choke converter 28, a constant current can be passed through the heater of the magnetron 36 regardless of the operating frequency, and the magnetron 36 can be stabilized. You can expect this behavior.

Note that the overcurrent detection circuit 68 shown in FIG. 4 outputs a high level signal when it detects an excessive primary current, and the abnormal voltage detection circuit 70 outputs a high level signal when it detects an excessive or insufficient power supply voltage. Outputs the signal. The outputs of overcurrent detection circuit 68 and abnormal voltage detection circuit 70 are applied to flip-flop 74 through OR gate 72 . Therefore, when an abnormal state occurs in the half-bridge converter, the signal from the flip-flop 74 is applied to the first terminal (see FIG. 7) of the integrated circuit "GP605". Therefore, the operation of the V/F conversion circuit 60 is stopped, the switching control signals QA and QB both remain at high level, and the operation of the half-bridge converter is stopped.

Note that the bandpass filter 54 as harmonic component extracting means and the smoothing circuit 55 as averaging means shown in FIG. 4 may be replaced with a low-pass filter.

[Brief explanation of drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a waveform diagram showing currents or voltages at various parts in the embodiment of FIG. 1;

FIG. 3 is a waveform diagram showing currents or voltages at various parts in the embodiment of FIG. 1;

FIG. 4 is a block diagram showing details of the control circuit of the embodiment of FIG. 1;

FIG. 5 is a graph showing a frequency spectrum representing harmonic components of the primary current of the high-frequency transformer of the embodiment of FIG. 1;

FIG. 6 is a graph showing the relationship of the 120 Hz component to the input current of the half-bridge converter.

FIG. 7 is a functional block diagram showing an example of the V/F conversion circuit of the embodiment of FIG. 4;

8 is a waveform diagram showing switching control signals from the V/F conversion circuit shown in FIG. 7. FIG.

9 is a graph showing the relationship between the frequency of the switching control signal and the input voltage of the V/F conversion circuit shown in FIG. 7. FIG.

FIG. 10 is a circuit diagram showing an example of the driver of FIG. 4;

FIG. 11 is a circuit diagram showing an example of the soft start circuit of FIG. 4;

FIG. 12 is a circuit diagram showing an example of the activation control circuit of FIG. 4;

13 is a waveform diagram of a primary current showing intermittent oscillation operation achieved by the start-up control circuit of FIG. 12; FIG.

14 is a circuit diagram showing a modification of the V/F conversion circuit of FIG. 4. FIG.

FIG. 15 is a waveform diagram showing the operation of a modification of FIG. 14;

FIG. 16 is a waveform diagram showing the operation of a modification of FIG. 14;

FIG. 17 is a circuit diagram showing an example of a diode breakdown detection circuit.

18 is a waveform diagram showing the operation of the diode breakdown detection circuit of FIG. 17 during normal operation.

19 is a waveform diagram showing the operation of the diode breakdown detection circuit of FIG. 17 during an abnormality.

FIG. 20 is a circuit diagram showing another example of a diode breakdown detection circuit.

FIG. 21 is a circuit diagram showing a modified example of a magnetron heater circuit.

22 is a circuit diagram showing a ringing choke converter according to a modification of FIG. 21; FIG.

[Explanation of symbols]

10
...Power supply device 12 for microwave oven
...control circuit 14
...Commercial power supply 16

...Diode bridge 18
...Common mode choke coils 20a, 20b
...switching element 22
...Connection lines 24a, 24b
...Discharge resistance 26
a, 26b...Resonance capacitor 28
...Ringing choke converter 30a, 30b
...diode 32
...Detection resistor 34
...High frequency transformer 34a
...Primary windings 34b, 34c
...Secondary winding 36
...Magnetron 38
...High voltage diode 40
...High voltage capacitor 42
…Potential transformer 44
…Current transformer 46
…Cooling fan 48
...Fan driver 50
...Microcomputer 52, 64
...Diode bridge 54
... Bandpass filter 56, 661, 664, 80, 86... Comparator 58

…Soft start circuit

Claims (8)

[Claims] 1. A first series connection of first and second switching elements driven by a commercial power source, and a second resonant capacitor of first and second resonant capacitors connected in parallel to the first series connection. a primary winding connected between a connection point of the first series connection and a connection point of the second series connection, and a primary winding magnetically coupled to the primary winding to supply voltage to the magnetron. a half-bridge converter including a high-frequency transformer having a secondary winding, set voltage output means for outputting a set voltage that commands a desired output power of the magnetron, and current detection means for detecting the current flowing through the primary winding. , averaging means for averaging the current detected by the current detection means, error voltage output means for outputting an error voltage corresponding to the difference between the voltage obtained by the averaging means and the set voltage, and the error A power supply device for a microwave oven, comprising switching control means for controlling the first and second switching elements based on the magnitude of voltage. 2. The switching control means has a frequency that varies depending on the magnitude of the error voltage, but a constant on time of 2.
2. The power supply device for a microwave oven according to claim 1, further comprising voltage/frequency conversion means for outputting two switching signals. 3. The first series connection includes the first and second series connections.
the current detection means is coupled to the connection line to connect the first and second switching elements;
3. The power supply device for a microwave oven according to claim 1, further comprising a current transformer for detecting a current flowing through each of the switching elements. 4. The connecting line includes a first portion in which a current in a first switching element flows in the same direction as a current in a second switching element, and the current transformer includes a first portion of the connecting line and The power supply device for a microwave oven according to claim 3, wherein the power supply device for a microwave oven is coupled to a second portion of the second switching element through which current flows. 5. Further comprising a stop means for controlling the switching control means to stop switching operations of the first and second switching elements when the current detected by the current transformer exceeds a predetermined value.
The power supply device for a microwave oven according to claim 1. 6. The averaging means includes harmonic component extracting means for receiving the output of the current transformer and extracting harmonic components of the fundamental wave of the commercial power source. Power supply device for microwave oven. 7. The harmonic component extracting means extracts two of the fundamental waves.
7. The power supply device for a microwave oven according to claim 6, further comprising second harmonic extracting means for extracting a component having twice the frequency. 8. The power supply device for a microwave oven according to claim 6, wherein said harmonic component extraction means includes a filter circuit.