JPH02239589A - High-frequency heating device - Google Patents

Abstract

PURPOSE:To prevent the reduction of inductance of the winding of a transformer and a thermal breakdown of the component element of an inverter by controlling to make the temperature of the core of the transformer to compose a part of the inverter less than the preset temperature near the lowest temperature to increase the power loss of the converter. CONSTITUTION:When the temperature of a core 121 is increased extensively by generating a variation of the surrounding temperature ambiance or a shortage of cooling and the like owing to the voltage variation or the motor life, the power loss of a magnetron driving converter 104 is increased, the anode current is reduced, and the output of the secondary current return device 111 is also reduced. As a result, a starting time width control circuit 14 is going to adopt a larger W1s rapidly, but since the detected voltage of a temperature detecting device 112 is increased, a comparison circuit 17 outputs a subtraction signal, and the ON-time width output from an ON-time width control circuit 13 is suppressed. Consequently, the ON-pulse width is suppressed, the temperature rise of the magnetron driving converter 104 is also suppressed, and the tendency of a reckless heating can be prevented.

Description

DETAILED DESCRIPTION OF THE INVENTION Field of Industrial Application This invention relates to the improvement of a high-frequency heating device for performing so-called dielectric heating such as a microwave oven. This invention relates to the improvement of a high-frequency heating device configured to generate high-frequency power using a step-up transformer to drive a high-frequency generating means such as a magnetron. Conventional technology Conventionally, the magnetron drive transformer used in this type of high-frequency heating device uses a so-called power ferrite material, which has low power loss in the high frequency band and high specific resistance. The reason why such a power ferrite material is used is that a high specific resistance is advantageous because it reduces eddy current loss in the core. The power loss in the high frequency band of the ferrite core is expressed by the following equation. P(L)=P(h)+P(e)=(κ(b)*f
1κ(e) * f”) Book B(m) * V(e)
...-<1) However, P(L): bower loss, P(h): hysteresis loss, P(e): eddy current loss, K(h): hysteresis loss constant, K(e): eddy current loss Constant, f: frequency, B(m): magnetic flux density B to the m power (m constant number). V(
e): Effective volume of the core. From this first equation, hysteresis loss is proportional to frequency f, and eddy current loss is proportional to the square of frequency f. It turns out that Therefore, in the case of high-frequency drive, eddy current loss contributes more to power loss than hysteresis loss. Therefore, to reduce eddy current loss, bower ferrite material with high resistivity is used. Bower ferrite material has temperature characteristics as shown in Figure 9. When the operating temperature is low, even if the temperature rises, it is not a problem because the power loss decreases. However, if the operating temperature becomes too high, the magnetic properties of the power ferrite material deteriorate, resulting in increased power loss. That is, the operating temperature is at the temperature TI in FIG. 9 (for example, 130
℃), power loss continues to increase as the temperature rises, and the core temperature continues to rise. When entering such a thermal cycle, the core temperature instantly exceeds the Curie point and the magnetic permeability decreases dramatically. The inductance of the transformer winding is L=μ*r*a”*11”*a −(
2) Where, μ: magnetic permeability, π: pi, a: distance from the magnetic core to the winding, n: number of turns per unit length, α: Nagaoka coefficient. As expressed by Conventionally, the core size and
The safety design included large margins for winding diameter, cooling conditions, etc. Therefore, the size of the transformer is large and a large cooling fan must be installed to lower the operating temperature, which naturally makes it extremely difficult to create an inverter-type high-frequency heating device with high output (with large self-heating). Met. Problems to be Solved by the Invention An object of the present invention is to suppress the temperature rise in the core of a transformer,
It is an object of the present invention to provide a high-frequency heating device that prevents a decrease in the inductance of the winding of the transformer, thereby reliably preventing thermal destruction of the components of the inverter. Means for Solving the Problems The present invention comprises: a rectifier and smoothing circuit that rectifies and smoothes commercial AC power to obtain DC power; a l&frequency generating means; and a circuit that is energized by the DC power of the rectifier and smoother circuit to generate high-frequency power. an inverter that boosts the high frequency power of the inverter and supplies it to the high frequency generation means; a temperature detection means that detects the temperature of the core of the transformer; , the lowest temperature T at which the power losses of the transformer increase with increasing temperature of its core.
This is a high-frequency heating device characterized by including means for controlling the output of the inverter by $1 so that the temperature is below a predetermined temperature around l. According to the present invention, the temperature of the core of the transformer constituting a part of the inverter is detected by the temperature detection means, and as the temperature of the core increases, the temperature of the core and therefore the transformer increases. The output of the inverter is controlled by the vsm means so that it is below a predetermined temperature near the lowest temperature Tl at which the power loss increases. Therefore, the temperature of the core is suppressed to the predetermined temperature range, thereby preventing the inductance of the transformer winding from decreasing. This prevents excessive current from flowing through the components that make up the inverter, such as the semiconductor switching elements connected to the inverter's resonant circuit, thereby reliably preventing thermal damage. The predetermined temperature may be a temperature at which the minimum temperature T1 has not yet been reached, or may be a value equal to or higher than the minimum temperature T1 near the minimum temperature Tl. This makes it possible to create a compact high-frequency heating device with an inverter that has higher output and higher reliability than conventional devices. Embodiment FIG. 1 is a block diagram showing an embodiment of the present invention.
The power from the commercial AC power supply 100 is supplied to the rectifier and smoothing circuit 101.
After being rectified by , it is smoothed and DC power is obtained. The inverter 102 includes a step-up transformer 104 for driving the magnetron M1, which is a high-frequency generating means,
A resonant capacitor 103 is connected in parallel to the primary winding ml17, and a semiconductor switching element 105 realized by a transistor or the like is connected in series to the transformer 104.
is connected. Primary winding 117 and resonant capacitor 103
may be connected in series to form a resonant circuit. The drive circuit 8107 drives the semiconductor switching element 105. The control circuit 108 is the output setting circuit 1
09, a signal representing the current corresponding to the output of the magnetron M1 from the secondary current feedback means 111 that detects the current in the secondary winding 119 of the transformer 104 that drives the magnetron M1, and the signal representing the current corresponding to the output of the magnetron M1. A signal from the voltage return means 106 to which the output of the auxiliary winding 120 of the transformer 104 is given, and a signal from the temperature detection stage 122 fixed directly to the core 121 of the transformer 104 or provided in the vicinity of the core 121. The width of the pulse given to the drive circuit 107 is controlled in response to the signal. Drive circuit 1 that drives magnetron M1
10 doubles and rectifies the output of the secondary winding 119 of the transformer 104, and includes a capacitor 112 and a high voltage diode 113. The heater winding 118 of the transformer 104 is connected to the filament terminal of the magnetron M1, and capacitors 115 and 116 for preventing radio wave leakage noise are connected between the filament terminal and the anode. In this way, the magnetron circuit 114 is constructed.When the magnetron M1 is in steady oscillation, the anode and cathode positions of the five magnetrons M1 are constant, so the output is constant Ili! I! If you try to
Anode current, i.e. secondary winding 119 of transformer 104
It is sufficient to control the current by a constant m. an input from the voltage output means 106, a setting input from the output setting circuit 109, and 2
Input from next current feedback means 111 and temperature detection means 12
2, the control circuit 108 provides the drive signal shown in FIG. 2(1) to the drive circuit 107. Based on the input from the output setting circuit 109 and the secondary current feedback means 111 (comparison of the output setting value of the immediately preceding power cycle and the average secondary current), the control circuit 108 determines the rise-on time width W1, Outputs an on signal. Then, the semiconductor switching element 105 becomes conductive, and the second r:l(2)
The collector current indicated by the broken line l1 in the middle is supplied to the primary winding 117 of the transformer 104. When the fl41M circuit 108 outputs an off signal, the semiconductor switching element 105 becomes non-conductive, and the resonant capacitor 103 and the primary winding 1
17 constitute a resonant circuit, and a resonant voltage appears in the collector voltage of the semiconductor switching element 105, which is a transistor, as shown by the solid line I2 in FIG. 2(2). That is, the collector of the transistor, which is the semiconductor switching element 105, is connected to the connection point between the primary winding 117 and the capacitor 103, and its emitter is grounded. In FIG. 2(1), the period W2 at which the semiconductor switching element 105 shuts off is a substantially constant value depending on the resonant frequency of the inverter 102, and the leading *fmW1 controls the output of the magnetron M1. Changed for. The inverter 102 has a frequency of 20kHz to 100K
It operates at a frequency of about Hz, and as shown in Figure 3, it operates at a frequency of about 50Hz.
The collector voltage waveform of the semiconductor switching element 105 is shown at a power supply cycle of 2 or 60 Hz. In the case of high output as shown in Fig. 3 (1), the area near the valley of the power supply voltage rises with a wide ON closing width, and both! When the voltage feedback means 106 detects that the voltage has reached a predetermined value, the control circuit 108 sequentially decreases and increases the on-time width, and adjusts the on-time width so that the collector voltage waveform becomes flat at the peak of the power supply voltage. control. Then, when the power supply voltage decreases, the collector voltage does not rise to the specified value even with a wide on-time width, and the wide on-time width continues until the valley. This maximum on-time width varies depending on the output setting value, and the on-time width of the rise at low output (Fig. 3 (2)) is narrower than that at high output. Therefore, even when the power supply voltage reaches a high peak, the collector voltage does not decrease to a predetermined value, and the control circuit 108 only outputs ON signals with the same ON time width. The off time width is determined by the resonance capacitor 103 and the primary winding. It is determined by the resonance frequency of the line 117 and is approximately constant.
! ). Voltage feedback 1,000 steps indicates that the resonant voltage has fallen.
When detected by the signal from 06 (synchronous detection), the control circuit 1
08 determines that the off time has ended and outputs the next on signal. The upper limit value of the resonant voltage mentioned above is a value that is determined by the design of the component 1g of the inverter 102 (mainly the rating of the live conductor switching f: part 105), and the larger the upper limit value is, the higher the output can be obtained. It is. Furthermore, even under the same upper limit value, the wider the on-time width of the rising edge, the faster the resonance voltage rises, the larger the output power, and the higher the output. When the semiconductor switching element 105 turns non-conducting, the magnetron drive transformer 104
The next winding 119 is the magnetron drive circuit 1. The capacitor 112
Charge. Then, when the semiconductor switching element 105 is reversed to conduction, the output voltage of the secondary winding 119 is reversed, and the positive voltage of the magnetron M1 is reversed. A high voltage is applied to the anode (anode), and the charge stored in the capacitor 112 is released as an anode current. Heater winding 1 of transformer 104
18 generates an alternating current voltage in synchronization with the switching operation of the semiconductor switching element 105, and supplies power to the filament terminal of the magnetron M1. The leakage inductance of the secondary winding 119 suppresses the peak value of the anode current, but the wider the on-state leap width of the control circuit 108, the greater the resonance energy, and a high voltage higher than the emission voltage (oscillation voltage) of the magnetron M1 is applied. Since the conduction time is expanded, the effective value of the anode current increases. Therefore, the heating output power of the fourth magnetron M1 with the larger output setting value also becomes larger. However, if insufficient cooling occurs due to changes in the ambient temperature environment, voltage fluctuations, or the life of the motor, the temperature of the core 121 becomes extremely high, increasing power loss in the transformer 104, resulting in insufficient output. , secondary current feedback means 11
1 decreases, the control circuit 108 tries to give the semiconductor switching element 105 an increasingly wider on-time width. However, the temperature detection stage 122 provided directly on or near the core 121 detects that the temperature of the core 121 is at a predetermined level! When it is detected that the temperature exceeds (temperature Tl in FIG. 9 or a predetermined temperature T 1 a in the vicinity thereof), a suppressing sl signal is given to the @control circuit 108, and the control circuit 108 reduces the leap width at the time of ON. Let. This suppresses heat generation in the magnetron drive transformer 104, prevents thermal runaway, and prevents thermal destruction of the live conductor switching element 105, although the heating output decreases. It goes without saying that a failure alarm or display is issued based on the output of the temperature detector 122, which allows the user to immediately take repair measures to eliminate the cause of the temperature rise. In this way, a highly reliable high-frequency heating device can be constructed. The present invention will be described in detail below based on the embodiment shown in FIG. 4, which shows a more specific l'lI. Note that this invention is not limited by this. FIG. 4 is a circuit diagram of an embodiment in which the secondary winding 119 of the transformer 104 is configured to have leakage inductance. Functional parts that are the same as those in Figure 1 are given the same numbers and explained. A rectifying and smoothing circuit 10 is connected to a commercial power supply 100 via a switch 2.
1 is connected. The 11th flow smoothing circuit 101 is constructed by connecting a choke coil 4 and a smoothing capacitor 5.6 to the output terminal of the rectifier bridge 3. A parallel resonant circuit consisting of the primary winding 117 of the magnetron driving transformer 104 and the resonant capacitor 103 is connected to the DC output terminal of the Il current smoothing circuit 101, and the primary winding 11 of the transformer 104 is
7 and the semiconductor switching element 105 are connected, and the damper diode 7 is connected to the semiconductor switching element 10.
It is connected in reverse to the collector-emitter connection of 5. A magnetron drive circuit 110 that drives the magnetron M1 is composed of a capacitor 112 that doubles the output of the high voltage secondary winding 119 of the transformer 104 by a half-wave voltage, and a high voltage diode 113. This high-voltage secondary winding 119 has a leakage inductance of 2, and has a choke effect on the secondary current. The heater winding 118 of the transformer 104 is a magnetron M]
It is connected to the filament terminal of #. Furthermore, capacitors 115 and 116 for preventing radio wave leakage noise are built in between the filament terminal and the anode of the Magnetoron M1. @The control circuit 108 creates a DC power supply (+Vc=+5V) with the DC power supply circuit 9 and connects the semiconductor switching element 1.
In order to perform high-frequency switching of 05, an on-off pulse signal is generated by the pulse generation circuit l1, and outputted to the drive circuit 107 by the isolation transformer 8. The ground potential of the DC voltage circuit 9 is the same as the anode potential of the magnetron M1. Secondary winding 119 of transformer 104
is connected to the secondary current feedback stage 111. The anode current corresponding to the output C of the magnetron M1 is detected by the voltage across the resistor 35. Referring to FIG. 5, the synchronous circuit 10 described later is shown in FIG. 5(1).
A synchronizing signal shown by is given to the pulse generation circuit 11 and the on-time counter 12. The on-time power counter 12 responds to the signal from the on-time width control circuit 13 and outputs the synchronous circuit 10.
When receiving the pulse f,t L, the predetermined time W1
Finally, the off-pulse shown in Figure 5 (2) is derived. In this way, the pulse generating circuit 11 is configured as shown in FIG. 5(3).
Derive a pulse with on-time W1 as shown in .
The off time W2 is a value determined by the resonance frequency of the primary winding 117 of the transformer 104 and the capacitor 103, as described above. When ON, E W only 1, transistor 10
5 is conductive. Only the phase in which the t current flows to the anode of the magnetron M1 is inputted to the diode 36, smoothed by the resistor 3"l and the capacitor 38, and the average value of the anode current is input to the comparison circuit 1.
Enter 5. The rise-on time width control circuit 14 determines the rise-on time IWI of the swiss-pulse according to the output of the comparison circuit 15 which compares the value with the input from the output setting circuit 109. The anode current of Magmotoron M1 is 20kHz to 100kHz.
k}{z is turned on and off, but the secondary current feedback means 111 uses the time constant of the resistor 37 and capacitor 38,
It is converted into a DC voltage and input to the comparator circuit 15. Let us assume that the input voltage is Vi (rn). A value preset in the rising on time width control circuit 14 is selected as the initial rising on time width W1 at the start of operation. Then, the first power blackout [(4th station in Figure 4 is bridge 3
The average value data of the anode current (120 Hz or 100 H2 since the full wave l1 is flowing at 1) is input from the secondary power supply feedback means 111 to the comparator circuit 15 as Vi (1). If it is smaller than the input DC voltage (denoted as ■.) from the output setting circuit 109, the next rising on time width W1 is set to a value larger than the previous one by the rising on time width Il control circuit 14. do. On the contrary, Vi (n)
But ■. , a value smaller than the previous W1 is selected as the next W1. That is, once every power cycle interval, the signal from the comparison circuit [15 is read by the rising on time #4f#s control circuit 14, and the result is reflected in the rising on time width of the next 1i power cycle. 31
16, the input voltage V i (ri) is the reference DC voltage from the output setting circuit 109. When the voltage is lower than the reference voltage (2), the output of the comparison circuit 15 is at a high level as shown by the broken line l3, and the reference voltage (2) is at a high level. When the above is the case, the output becomes a low level, and the times tl and t2 are as follows.
When the core temperature of the U-magnetron drive transformer 104, which indicates the timing at which the control circuit 14 reads the output of the comparison circuit 5, increases, power loss increases and Vi(n) becomes low F, resulting in a large W1. will now be selected. The auxiliary winding 120 outputs a voltage similar to the primary winding 117 of the magnetron drive transformer 104! Voltage feedback means 10
Import into 6. The detection voltage of the off phase (flyback phase) of the magnetron M1 is converted into a pulse signal of +Vc or less with reference to the ground potential by resistors 22, 23, 26 and clamper diodes 24, 25 and input to the synchronization circuit 10. The capacitor 27 is for noise removal.The detection voltage of the same 'flyback phase' is connected to the diode 28, the resistor 29,
32, a capacitor 30, and a differential offset circuit including a Zener diode 31.
It is input to the comparator circuit 16 as the Vce control signal of 05.
At the same time, the overvoltage detection circuit 18 is divided by resistors 33 and 34.
Enter. The comparator circuit 16 detects that when the Vc e@ control signal exceeds a predetermined DC voltage divided by the resistor 39.40,
The subtraction signal is output to the on-time width control circuit 13. The capacitor 30 of the differential offset circuit has a smaller impedance than the resistor 29, thereby improving responsiveness. The Zener diode 31 functions to create a DC bias. As mentioned above, the similar voltage shown in FIG. The output voltage shown in FIG. 7(2) of the comparator circuit 16 is input to the on-time width control circuit 13 as a brake action that controls the on-time leap width. VJ than the divided voltage v1 by resistor 39.40
(collection of pulse voltages from 20 kHz to 100 kHz) becomes larger, the on-time width control circuit 13 suppresses the on-time width W1 as a braking action. Transformer 104
As the core temperature increases, power loss increases and inductance decreases, so Vj decreases and the effect of suppressing W1 decreases. A signal from the temperature detection means 122 that detects the temperature of the core 121 of the magnetron drive transformer 104 is input as a voltage to the comparator circuit 17 by a resistor 19. If the temperature of the core 121 rises abnormally, the detected voltage becomes larger than the partial voltage of the resistor 20.21, and the comparison circuit 1
7 outputs the subtraction signal to the leap width IIIfJ4 circuit 13 when on. While the power supply voltage is low (near the valley of the power supply), the comparison circuit 16 does not output the subtraction signal, and the on-time leap width control circuit 13 changes the output of the rising on-time width control circuit 14 to the on-time width W1. Adopted as. That is, the on-time counter 12 measures the time width W1 in synchronization with the synchronization signal of the synchronization circuit 10, and the pulse generation circuit 11 generates a switching pulse with the on-time width W1 through the isolation transformer 8. Output to the drive circuit 107. When the output of the pulse generating circuit 11 becomes high level, the semiconductor switching element 105 becomes conductive, and when the output becomes low level, the semiconductor switching element 105 becomes non-conductive. When the Vce of the semiconductor switching element 105 rises from the valley of the power supply with the on-time width W1 and reaches the control level, the Vce control signal of the voltage feedback means 106 increases, and the Vce control signal of the voltage feedback means 106 increases.
outputs the subtraction signal. In response to this, the on-time width control circuit 13 gradually reduces the on-time width W1, and the pulse generating circuit 11 outputs an on-pulse such that Vce of the semiconductor switching element 105 reaches a predetermined value. The capacitor 30 of the differential off-set circuit of the voltage feedback means 106 provides a quick control response, and the Zener diode 31 controls the center portion of the collector-emitter leap voltage Vce waveform in FIG. 3 to be slightly rounded. When the power supply voltage decreases, even if the on-time width returns to W1, Vce continues to decrease, and the subtraction signal of the comparator circuit 16 is no longer output, and the on-time width remains at W1 and advances to the valley of the power supply. When the set output is small, the output of the comparator circuit 15 is inverted when the average secondary current input is at a low level, so the rising on-time leap width@control circuit 14 sets the on-time leap width W1 narrower than in the case of a high output setting. Adopted as. As a result, as shown in Fig. 3 (1) and Fig. 3 (2), the Vce waveform changes according to the set output, and the power is controlled.In the rise-on leap width control circuit 14, when the power is turned on, A predetermined long on-time width W 1 s is set, and at this time, when the output of the comparison circuit 15 is at a high level, that is, when the set output of the output setting circuit 109 is larger than the output of the current feedback circuit 111, When setting for a long time, leave the leap W 1 s.When the output of the comparator circuit 15 becomes a low k level, that is, the output of the current feedback circuit 111 becomes the output setting circuit 1.
When the output exceeds the set output of 09, the on-time leap width Wl
The control circuit 14 operates to subtract a predetermined constant value from s at the power supply cycle. When the output of the comparison circuit 15 becomes high level again, the stabilization circuit 14 adds a predetermined value to the calculated on-time width at the power supply cycle. The on-time width control circuit 13 corrects the on-time width at the cycle of the oscillation frequency of the inverter 102 in response to the outputs of the comparison circuits 16 and 17. That is, the on-time leap width is determined by the rising on-time width control circuit 14, and when the output of the comparison circuit 16 is at a high level, that is, when the output voltage of the auxiliary winding 120 is relatively high, the on-time leap width is determined by the rising on-time width control circuit 14. Inverter 10 to shorten the width
The on-time width is set short in the period of the oscillation frequency of 2, so that the output of the comparator circuit 16 is at a low level, that is, the output voltage of the auxiliary winding 120 is reduced. When the output of the comparison circuit 16 becomes low level, the on-time width is increased at the cycle of the oscillation frequency of the inverter 102 so that the on-time width increases. When the overvoltage detection circuit 18 detects that the inverter 102 oscillates abnormally and the Vce of the semiconductor switching element 105 (L, therefore, the flyback voltage of the magnetron drive transformer 1104) increases abnormally.
The output of the pulse generation circuit 11 is stopped instantly, and the oscillation of the inverter 102 is stopped. Then, the body is stopped for a certain period of time (for example, lmsec) and restarted automatically. In this way, overvoltage damage to components such as the semiconductor switching element 105 is prevented. Inverter 1 is controlled by the output of control circuit 108.
When the switching operation of 02 starts, the magmotoron M
A heater current flows through the first filament terminal, raising the filament temperature, and then an anode current begins to flow, and microwaves are radiated into the chamber of the high-frequency heating device. Here, if the temperature of the core 121 becomes extremely high due to changes in the ambient temperature environment, voltage fluctuations, insufficient cooling due to the life of the motor, etc.
The power loss increases, the anode current decreases, and the output of the secondary current feedback means 111 also decreases. Therefore, the rising ON time width control circuit 14 tries to adopt a larger W 1 s, but since the detection voltage of the temperature detection means 122 increases, the comparison circuit 17 outputs a subtraction signal, and the ON time width control circuit The ON time range output by 13 is suppressed. As a result, the on-pulse width is suppressed, the temperature rise of the magnetron drive transformer 104 is also suppressed, and thermal runaway can be prevented. That is, overcurrent damage to the semiconductor switching element 105 caused by thermal runaway can be prevented. As mentioned above, when the core temperature increases,
Comparison circuits 15 and 16 have no suppressing effect and proceed to thermal runaway. Therefore, the temperature detected by the core temperature detection means 122 is converted into a voltage and input to the comparison circuit 17. A value T near the operating temperature T1 at which the power loss of the core becomes so large that it cannot be ignored.
1 a, reference voltage (determined by resistors 20 and 21) V2
Design to exceed. Then, the comparator circuit 17 outputs a suppression signal to the on-time width control circuit 13, so the primary side excitation current of the magnetron drive transformer 104 decreases.
As a result, heat generation in the core is suppressed. FIG. 8 shows the operation of the comparison circuit 17, and shows the temperature detection stage 12.
When the voltage Vk corresponding to the temperature of 2 is less than the divided reference voltage 2 of the resistor 2021, the output is at a low level as shown by the broken line, and when the voltage Vk is equal to or higher than the reference voltage 2, The output of the Hikyo circuit 17 is at a high level, and the output of this comparison circuit 17 is the same as the above-mentioned. The on-time question width s4 is given to #l13 as follows. According to the embodiment described above, when the core temperature of the magnetron drive transformer is about to rise due to changes in the ambient temperature environment, power supply voltage fluctuations, deterioration of motor life, etc., the core temperature of the magnetron drive transformer increases. A level is determined and the inverter output is suppressed to prevent it from exceeding that level.
Moreover, it is possible to prevent thermal runaway and furthermore, damage to semiconductor switching elements due to excessive Ti. This means that the thermal design margin of the magnetron drive transformer can be made smaller than before, and in order to obtain the same output, a much smaller transformer is required than before, and if the same size transformer is used. This makes it possible to realize a high-temperature heating device with much higher output than conventional methods. Effects of the Invention As described above, according to the present invention, the temperature of the transformer core is
This prevents the winding inductance of the transformer from decreasing, making it possible to reliably prevent thermal damage to the components that make up the inverter. It is possible to realize a high-frequency heating device equipped with an inverter with a higher output and higher reliability than conventional ones.

[Brief explanation of drawings]

FIG. 1 is a block diagram of an embodiment of the present invention, the second part is a waveform diagram for explaining the operation of the control circuit 108, and the third part is a waveform diagram for explaining the operation of the control circuit 108.
is a diagram showing the collector voltage waveform of the semiconductor switching element 105, FIG. 4 is a block diagram showing a specific configuration of an embodiment of the present invention, and FIG. 5 is a diagram for explaining the basic operation of the control circuit W8108. Waveform diagrams, FIG. 6 is a diagram for explaining the operation of the comparison circuit 15, FIG. 7 is a diagram for explaining the operation of the comparison circuit 16, and FIG. 8 is a diagram for explaining the operation of the comparison circuit 17. , FIG. 9 is a graph showing the temperature characteristics of the bower ferrite material that constitutes the core 12l of the transformer 104. 100... commercial power supply, 101... rectifier/smoothing circuit,
102... In'verter, 103... Resonant capacitor, 104... Magnetron drive transformer, 105
... Cliff, conductor switching element. 106... Four stages of voltage/voltage feedback, 107... Drive circuit, 108... III control circuit, 109... Output setting circuit, 110... Magnetron drive circuit, 111, . Secondary current feedback means, 11
2... Capacitor, 113... High voltage diode, 1
15,1. ．． 16... Radio wave leakage prevention capacitor with built-in magnetron, 121... Core of transformer 104, 122...
...Temperature detection means, M1... Magnetron agent Patent attorney Keiichiro Nishikyo

Claims (1)

[Claims] A rectifying and smoothing circuit that rectifies and smoothes commercial AC power to obtain DC power; a high-frequency generating means; an inverter that is energized by the DC power of the rectifying and smoothing circuit to generate high-frequency power; A transformer that boosts high frequency power and supplies it to the high frequency generation means; a temperature detection means that detects the temperature of the core of the transformer; The minimum temperature T at which the power loss of the transformer increases accordingly.
and means for controlling the output of the inverter so that the temperature is below a predetermined temperature around 1.