JPH04271297A - Induction motor controller - Google Patents

Abstract

PURPOSE:To control the rotational speed of induction motor stably at all times without causing deficiency of torque or overcurrent and regardless of the variation rate of primary frequency command or a machine driven through the induction motor. CONSTITUTION:A noload voltage operating circuit 5 outputs the noload voltage of an induction motor 1. An error current component operating circuit 6 outputs an error current component which goes zero when the actual value of primary flux induced in the induction motor matches with a command value of primary flux determined by the product of an exciting current value and the primary self inductance of the induction motor. A correction voltage operating circuit 7 outputs a correction voltage for bringing the error current component close to zero. A primary voltage command operating circuit 8 outputs a primary voltage command value of the induction motor. A variable frequency power converting circuit controls the actual value of the primary voltage to be applied on the induction motor so that it follows up the primary voltage command value.

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for controlling the primary frequency of an induction motor. [0002] FIG. 8 is a block diagram showing a conventional control device for an induction motor, which is shown in, for example, the literature (Introduction to Power Electronics, Section 8, 3, published by Ohm Publishing Co., Ltd., 1982). It is. In FIG. 8, 1 is an induction motor, 21 is a transistor inverter circuit for driving the induction motor 1 at a variable frequency, 22 is a frequency command generator, 23 is a function generator connected to the frequency command generator 22, and 24 is a transistor inverter circuit for driving the induction motor 1 at a variable frequency. A primary voltage command generating circuit 25 is connected to the frequency command generator 22 and the function generator 23, and a PWM circuit 25 is connected to the primary voltage command generating circuit 24. First, the principle of frequency control of an induction motor by this control device will be explained. FIG. 9 shows a T-shaped equivalent circuit per phase of a known induction motor. In the figure, R1 is the primary resistance, R2 is the secondary resistance, liter 1 is the primary leakage inductance, liter 2 is the secondary leakage inductance, M is the primary and secondary mutual inductance, ω1 is the primary frequency, and ωS is the slip frequency. , V1 is the primary voltage, E0 is the air gap induced voltage, I1 is the primary current, and I2 is the secondary current. First, the air gap magnetic flux Φ0 is the induced voltage E
0 and the primary frequency ω1, and the time integral of the voltage becomes the magnetic flux, so equation (1) holds true. ##EQU1## Note that the current I2r that acts on this magnetic flux Φ0 and generates torque is an effective component of the secondary current I2, that is, an in-phase component with the induced voltage E0. From FIG. 9, I2r is expressed as equation (2). ##EQU2## Furthermore, since the torque Te generated by the induction motor is proportional to the product of the magnetic flux Φ0 and the current I2r, equation (3) holds true. ##EQU00003## where K is a proportionality constant. Next, (1)
By substituting equation (2) into equation (3), equation (4) is obtained. ##EQU4## From equation (4), if E0/ω1 is controlled to be constant, the generated torque Te changes depending on the slip frequency ωS. At this time, the maximum torque Tmax can be calculated by differentiating equation (4) with respect to the slip frequency ωS and setting the numerator to zero.
5) Equation becomes. [Equation 5] Therefore, the maximum torque Tmax is E0/
If ω1 is constant, it is irrelevant to changes in ω1. By the way, in reality, it is not possible to easily detect the induced voltage E0, so the so-called constant V/F control method is usually used, in which the primary voltage V1 is made proportional to ω1 and the value of V1/ω1 is controlled to be constant. , used. In this case, the primary frequency ω
In the region where 1 is low, the voltage drop due to the primary resistance R1 is 1
Since the next voltage V1 cannot be ignored, a method is used in which V1 is increased in advance by a voltage corresponding to R1I1 in a low frequency region. Next, the operation of the control device shown in FIG. 8 will be explained. First, for the above-mentioned reason, the function generator 23 inputs the primary frequency command ω1* output from the frequency command generator 22 based on the functional relationship shown by the solid line in FIG. The amplitude command V1* is output. Next, the primary voltage command generation circuit 24 calculates the equation (6) from the primary voltage amplitude command V1* and the primary frequency command ω1* to apply it to each primary winding of the induction motor. Should 1
The next voltage commands V1u*, V1v*, and V1w* are output. ##EQU6## Next, the PWM circuit 25 turns on a transistor (not shown) constituting the transistor inverter circuit 21 according to these primary voltage commands V1u*, V1v*, V1w*. - A base signal for controlling the off operation is generated, and as a result, the primary voltage commands V1u, V1v, and V1w actually applied to the induction motor 1 are controlled to follow the respective commands. Therefore, the primary frequency command ω1
*It is possible to control the frequency, that is, the rotational speed of the induction motor 1, according to the above. [0017] Since the conventional induction motor control device is configured as described above, when a large torque is required to be generated during rotation, the primary resistance R1
In order to correct the voltage drop due to
It is necessary to set the amplitude command V1* of the next voltage higher in advance by the voltage drop. However, since the value of the primary resistance R1 changes depending on the temperature, it is difficult to accurately correct the voltage drop. Therefore, if the voltage correction amount is smaller than the actual voltage drop, if load torque is constantly applied to the induction motor, the generated torque at low speed rotation will be insufficient, so the induction motor cannot be started, and conversely, When the voltage correction amount is large, a large primary current flows during low-speed rotation, and there is a problem that the operation of the inverter circuit must be stopped in order to protect the inverter circuit from overcurrent. Further, even if the generated torque is the same, if the machines driven by the induction motor are different, the overall moment of inertia will be different, so the rate of change in the rotational speed of the induction motor will be different. Therefore, there is a problem in that unless the rate of change of the primary frequency command ω1* is appropriately adjusted, the induction motor cannot be normally accelerated or decelerated according to ω1*. This invention was made in order to solve the above-mentioned problems, and does not cause problems of insufficient torque or overcurrent during low-speed rotation, and is suitable for machines driven by induction motors and primary frequency command ω1. * The purpose of this invention is to obtain a control device for an induction motor that can always stably control the rotational speed of the induction motor regardless of the rate of change of . Means for Solving the Problems A control device for an induction motor according to the present invention is provided with the following means. (1) a no-load voltage calculation circuit that outputs a no-load voltage command; (2) an error current component calculation circuit that calculates an error current component based on the excitation current command value and the primary current of the induction motor;
(3) A correction voltage calculation circuit that inputs the error current component and calculates a correction voltage; (4) A primary voltage command calculation circuit that outputs a primary voltage command based on the no-load voltage command and the correction voltage. , (5) a variable frequency power conversion circuit that inputs the primary voltage command to control the primary voltage of the induction motor. [Operation] In the present invention, the no-load voltage calculation circuit outputs the no-load voltage of the induction motor. Further, an error current component calculation circuit determines that the actual value of the primary magnetic flux generated inside the induction motor is the command value of the primary magnetic flux given by the product of the excitation current command value and the primary self-inductance of the induction motor. An error current component that becomes zero when they match is output. Next, a correction voltage calculation circuit outputs a correction voltage that brings the error current component closer to zero. Furthermore, a primary voltage command value for the induction motor is outputted by a primary voltage command calculation circuit. The variable frequency power conversion circuit controls the actual value of the primary voltage applied to the induction motor to follow the primary voltage command value. [Embodiment] An embodiment of the present invention will be described below with reference to the drawings. FIG. 1 is a block diagram showing an entire embodiment of the present invention, and numerals 1 and 22 are exactly the same as those of the conventional device. In FIG. 1, 2 is a current detector that detects the primary current flowing through the motor 1, and 3 is a variable frequency power conversion circuit provided upstream of the induction motor 1. This conversion circuit 3 is, for example, a transistor in a conventional device. It is composed of an inverter circuit 21 and a PWM circuit 25. 4 is an excitation current command setter, 5 is a no-load voltage calculation circuit connected to the excitation current command setter 5 and the frequency command generator 22 and outputs a no-load voltage command, 6 is a current detector 2, and an excitation current command setting 7 is an error current component calculation circuit connected to the frequency command generator 22 and the excitation current command value and the primary current of the induction motor 1, and 7 is an error current component calculation circuit 6 and a frequency command. A correction voltage calculation circuit 8 is connected to the generator 22 and calculates a correction voltage, a correction voltage calculation circuit 7, a no-load voltage calculation circuit 5, and a frequency command generator 22.
1 based on the correction voltage and the no-load voltage command.
This is a primary voltage command calculation circuit that outputs a secondary voltage command. FIG. 2 is a block diagram showing the detailed configuration of the above-mentioned no-load voltage calculation circuit 5. As shown in FIG. In FIG. 2, the no-load voltage calculation circuit 5 includes an input terminal 10 connected to a frequency command generator 22, an input terminal 11 connected to the excitation current command setting device 4, and a coefficient multiplier connected to the input terminal 11. 12
and a multiplier 13 connected to the input terminal 10 and the coefficient unit 12.
and an output terminal 14 connected to a multiplier 13. FIG. 3 is a block diagram showing the detailed configuration of the error current component calculation circuit 6 described above. In FIG. 3, the error current component calculation circuit 6 is connected to an input terminal 30 connected to the excitation current command setter 4, input terminals 31 and 32 connected to the current detector 2, and a frequency command generator 22. input terminal 33, coefficient multipliers 34, 35, 36, 47 and 50, adders 37, 45 and 52, V/F converter 38, counter 39, ROM 40, and multiplication type D/A converters 41 to 44, subtracters 46, 48 and 53, multiplier 49, divider 51, and output terminal 54
~56. FIG. 4 is a block diagram showing a detailed configuration of the above-mentioned correction voltage calculation circuit 7. As shown in FIG. In FIG. 4, the correction voltage calculation circuit 7 has input terminals 60, 61 and 63 connected to the error current component calculation circuit 6, and an input terminal 62 connected to the frequency command generator 22.
Coefficient multipliers 64, 68 and 71 and amplifiers 65 and 67
, adders 66, 70, and 72, a multiplier 69, and output terminals 73 and 74. Figure 5 shows
FIG. 2 is a block diagram showing a detailed configuration of the above-mentioned primary voltage command calculation circuit 8. FIG. In FIG. 8, the primary voltage command calculation circuit 8 has input terminals 80 and 81 connected to the correction voltage calculation circuit 7, an input terminal 82 connected to the no-load voltage calculation circuit 5, and a frequency command generator 22. , adders 84, 93 and 96, V/F converter 85, counter 86, ROM 87, multiplier D/A converters 88 to 91, and subtracters 92 and 9.
5, coefficient units 94, 97-99, and output terminals 100-1
It is composed of 02. Now, before proceeding to the explanation of the operation of the embodiment, the control system for the induction motor according to the present invention will be explained. As is well known, the primary voltage V1 applied to the induction motor 1
u, V1v, and V1w can be converted into components V1α and V1β on orthogonal coordinate axes (assumed to be α-β coordinate axes) using the relational expression (7). [Equation 7] However, V1α: α-axis component of the primary voltage,
V1β: β-axis component of the primary voltage Conversely, V1α, V1β can be calculated from equation (7) by using the relational expression (8).
It can be converted to v, V1w. ##EQU8## Similar relational expressions also hold between the primary currents I1u, I1v, I1w and the α-axis component I1α and the β-axis component I1β, respectively, as expressed by equations (9) and ( 10) It is shown by the formula. [Equation 9] [Equation 10] Next, the voltage/current equation of the induction motor on the α-β coordinate axis is expressed by equation (11) as is well known. ##EQU11## However, L1 and L2 are the primary and secondary self-inductances of the induction motor, respectively, I2α and I2β are the α-axis and β-axis components of the secondary current, respectively, and ωm is the rotation speed,
P is a differential operator (=d/dt). Next, (11)
In order to convert the equation into a relational equation on a rotating coordinate axis (referred to as a d-q coordinate axis) rotating at a primary frequency ω1, use (12)
The coordinate rotation equations shown in equations (14) to (14) are used. [Equation 12] [Equation 13] [Equation 13] [Equation 14] [Equation 15] [Equation 15] Substituting equations (12) to (14) into equation (11), V1α , V1β, I1α, I1β, I2α, I2
By eliminating β, equation (16) is obtained. [Equation 16] [Equation 17] Next, the d and q axis components Φ1 of the primary magnetic flux Φ1
d, Φ1q are expressed by the formula (18) as is well known. ##EQU18## Substituting equation (18) into equation (16), I2d
, I2q, equations (19) and (20) are obtained, respectively. [Equation 19] [Equation 20] [Equation 21] Next, the d and q axis components of the secondary magnetic flux Φ2 are (
22) It is shown by the formula. [Formula 22] From equations (18) and (22), I2d, I2
By eliminating q, equation (23) is obtained. ##EQU23## Furthermore, by substituting equation (23) into equation (20) and eliminating I1d and I1q, equation (24) is obtained. [Equation 24] [0051] However, T2 = L2 /R2 ; quadratic time constant Next, when both sides of equation (24) are differentiated, PΦ1 is found on the right side.
Since terms of d and PΦ1q are generated, using equation (19), P
Equation (25) is obtained by eliminating Φ1d and PΦ1q. ##EQU25## Furthermore, by substituting equation (24) into equation (25), eliminating Φ1d and Φ1q, and expressing using a determinant, equation (26) is obtained. [0054] Here, the characteristic equation of equation (26) is (27)
The equation becomes: [Equation 27] The natural angular frequency ωn1 and the damping rate ζn1 are (2
8) It is given by Eq. ##EQU28## Therefore, as ω1 becomes larger, the attenuation rate ζ
Since n becomes small, the responses of PΦ2d, PΦ2q and, in turn, the secondary magnetic fluxes Φ2d, Φ2q become oscillatory. Therefore, in order to suppress this vibration by increasing the damping rate (
Transform equation (26) into equation (29). ##EQU29## Then, the characteristic equation becomes equation (30),
##EQU30## The natural angular frequency ωn2 and the damping rate ζn2 are (3
1) Given by Eq. ##EQU31## Therefore, by adjusting the values of the control gains Kcd and Kcq, the damping characteristics of the responses of PΦ2d and PΦ2q can be improved. Next (29
), in order for PΦ2d, PΦ2q to converge to zero, the term indicated by A must be zero. That is, it is necessary to control V1d and V1q according to equation (32). [Equation 32] Now, on the right side of equation (32), Φ2d, Φ
Since terms related to secondary magnetic flux components such as 2q are included, in order to calculate equation (32), 2q must be calculated in some way.
It is necessary to detect the primary magnetic flux, but here, the primary magnetic flux Φ
1 is controlled constant according to the set value, (
33) Assuming Eq. ##EQU33## However, I1d* is the excitation current command value. Then, Φ2d and Φ2q are calculated using equation (33). First, by substituting equation (33) into equation (23), we get (
34) Equation is obtained. ##EQU34## Next, by substituting equation (33) into equation (20), equation (35) is obtained. ##EQU00003## Furthermore, equations (33) and (35) are transformed into (24)
), equation (36) is obtained. ##EQU36## Here, further considering the steady state, P(ωs
If the value of Φ2q) is assumed to be zero, then equations (34) and (35)
By substituting equation (36) into equation (32), we obtain (3
7) Equation is obtained. [Formula 37] [Formula 37] However, [Formula 38] [Formula 39] [Formula 39] By the way, in a steady state, PΦ2d
, PI1q is zero, so from equation (36), (3
It can be seen that the value of Ierr given by equation 9) also becomes zero. That is, if equation (33) holds true, Ie
The value of rr becomes zero. Also, from equation (37), I1d
It can be seen that if *, I1d, and I1q are obtained, V1d and V1q can be calculated without detecting Φ2d and Φ2q. Furthermore, the response characteristic of the control system when controlling the primary voltage of the induction motor according to equation (37) is the control gain K
It can be seen that it can be determined by adjusting the values of cd and Kcq. Next, the operation of the above embodiment will be explained with reference to FIGS. 2 to 5. First, as shown in Figure 2,
No-load voltage command V1q0* is output by multiplier 13. That is, after inputting the excitation current command I1d* output from the excitation current command setting device 4 via the input terminal 11 to the coefficient unit 12, the output of this coefficient unit 12 and the input terminal 10 from the frequency command generator 22 are input. By multiplying the primary frequency command ω1* input via the multiplier 13, the no-load voltage command V1q0* (=L1ω1*I1d*) corresponding to the second term on the right side of the equation for V1q in equation (37) is obtained. It is determined and output from the output terminal 14. Next, as shown in FIG. 3, the error current component I
err, d-axis and q-axis components I1d and I1q of the primary current are output from the error current component calculation circuit 6. That is, the primary currents I1u and I1v of the induction motor 1 detected by the current detector 2 are respectively input to the input terminal 3.
1 and 32, the coefficient multipliers 34 to 36 and the adder 37 calculate equation (9), and the coefficient multiplier 34
The adder 37 outputs α-axis and β-axis components I1α and I1β of the primary current, respectively. On the other hand, the analog primary frequency command ω1* outputted from the frequency command generator 22 is sent to the V/F converter 3 via the input terminal 33.
8, a pulse train signal whose frequency is proportional to the primary frequency command ω1* is obtained, and the counter 39 calculates a digital angle command θ1*, which is the time integral value of the primary frequency command ω1*, and sinθ1 The values of * and cos θ1* are input as addresses of the ROM 40 where they are stored. Then, sin θ1* from ROM40
and cos θ1* are output. Next, the α-axis and β-axis components I1α and I1β of the primary current output from the coefficient unit 34 and the adder 37, and the ROM
When the digital quantities of sin θ1* and cos θ1* outputted from 40 are input to multiplication type D/A converters 41 to 44 for multiplication and analog conversion, they are input to adder 45 and subtracter 46 to obtain the inverse of equation (13). It is an arithmetic expression (4
0) is performed to obtain the d-axis and q-axis components I1d and I1q of the primary current. [0083] [Formula 40] [0084] Next, these I1d and I1q
From the excitation current command I1d* input from the excitation current command setter 4 via the input terminal 30, the coefficient units 47 and 50, the multiplier 49, the divider 51, the adder 52, and the subtracter 53 performs the calculation of equation (39), and the error current component Ierr obtained as the output of the subtracter 53 is output from the output terminal 54. Further, I1d and I1q obtained as outputs of adder 45 and subtracter 46 are output from output terminals 55 and 56, respectively. Next, as shown in FIG. 4, d-axis and q-axis correction voltage components ΔV1d and ΔV1q are output from the correction voltage component calculation circuit 7. That is, the d-axis component I1d and the error current component I of the primary current are transmitted from the error current component calculation circuit 6 through the input terminals 60, 61, and 63, respectively.
err and the q-axis component I1q of the primary current are input. Then, the coefficient unit 64, the amplifier 65, and the adder 66 calculate the right side of the equation for V1d in equation (37),
It is output from the output terminal 73 as a d-axis correction voltage component ΔV1d. On the other hand, the error current component Ierr and 1 input from the frequency command generator 22 via the input terminal 62
From the next frequency command ω1*, the amplifier 67 and the coefficient unit 6
8, a multiplier 69, and an adder 70, the third term on the right side of the equation for V1q in equation (37) is calculated, and the coefficient multiplier 7
1, the first term on the right side of the equation (37) for V1q is calculated. Next, adder 70 and coefficient unit 7
When the outputs of 1 are added by the adder 72, V1 of equation (37) is obtained.
The voltage of the second term on the right side of the equation for q, that is, the voltage component excluding the no-load voltage, is output from the output terminal 74 as the q-axis corrected voltage component ΔV1q. Next, as shown in FIG. 5, the primary voltage command V
1u*, V1v* and V1w* are output from the primary voltage command calculation circuit 8. That is, d-axis and q-axis correction voltage components ΔV1d and ΔV1q are inputted from the correction voltage component calculation circuit 7 via input terminals 80 and 81, respectively. Here, as can be seen from equation (37), the d-axis component V1d of the primary voltage is zero at no load, so △
V1d can be regarded as the d-axis component command V1d* of the primary voltage. On the other hand, the adder 84 adds the no-load voltage command V1q0* inputted from the no-load voltage calculation circuit 5 via the input terminal 82 and the q-axis correction voltage component △V1q, and the equation (37) is The right side of the equation for V1q is calculated and output as a q-axis component command V1q* of the primary voltage. Next, when the primary frequency command ω1* is input from the frequency command generator 22 via the input terminal 83, the RO
Digital values of sin θ1* and cos θ1* are output from M87. Then, the d-axis component command V1d* of the primary voltage inputted via the input terminal 80, the q-axis component command V1q* of the primary voltage outputted from the adder 84, and the sinθ1* and cos
Multiplying digital quantity of θ1* D/A converter 88~
After inputting to 91, multiplication and analog conversion, subtracter 92
When input to the adder 93, the calculation of equation (12) is performed, and the α-axis component command V1α* and the β-axis component command V1β* of the primary voltage are obtained. Next, the coefficient unit 94, 9
7 to 99, the subtracter 95 and the adder 96, (
8) The calculation of the formula is performed, and the primary voltage commands V1u*, V1v* and V1w are obtained from the output terminals 100 to 102, respectively.
* is output. Next, these primary voltage commands V1
When u*, V1v* and V1w* are input to the variable frequency power conversion circuit 3, the actual values of the primary voltages applied to the induction motor 1 follow these primary voltage commands by the same operation as the conventional device. controlled to do so. Note that in the above embodiment, the primary resistance R1
The voltage drop due to the d-axis and q-axis components of the primary current I1
d, I1q in the correction voltage calculation circuit, but by changing the configurations of the correction voltage calculation circuit 7 and the primary voltage command calculation circuit 8 as shown in FIGS. 6 and 7, respectively, the above The voltage drop is detected by current detector 2.
It may be corrected using the next currents I1u and I1v. That is, the correction voltage calculation circuit 7 in the block diagram shown in FIG.
At a, only the voltage component related to the error current component Ierr in equation (37) is calculated and output as d-axis and q-axis correction voltage components ΔV1d0 and ΔV1q0. In other words, △
V1d0 and ΔV1q0 are given by equation (41). [Equation 41] [0089] Next, these correction voltage components △V1d
0, △V1q0 is provided to input terminals 80a and 81a, and the primary voltage command calculation circuit 8 of the block diagram shown in FIG.
When input to a, the primary voltage command V1 is obtained from the coefficient multipliers 97 to 99, ignoring the voltage drop due to the primary resistance R1.
u*, V1v* and V1w* are output. Next, when the u-phase primary current output from the current detector 2 via the input terminal 103 is input to the coefficient unit 107, the u-phase 1
Since the voltage drop VRu due to the next resistance R1 is obtained,
When added to V1u0* by the adder 110, the u-phase primary voltage command V1u* including the voltage drop due to the primary resistor R1 is output from the output terminal 100. Similarly, when the V-phase primary current outputted from the current detector 2 via the input terminal 104 is input to the coefficient unit 108, the V-phase primary current including the resistance drop due to the primary resistance R1 is The next voltage command V1v* is determined and output from the output terminal 101. Next, for the w phase, first, using the well-known equation (42), the primary currents I1u, I1 are
The W-phase primary current I1w is determined from v. Subsequently, the coefficient unit 109 and the adder 112 output the W-phase primary voltage command V including the voltage drop due to the primary resistance R1.
1w* is determined and output from the output terminal 102. [0090] Alternatively, as another embodiment, the primary resistance R1
The voltage drop caused by the α-axis and β-axis components of the primary current I
Using 1α, I1β, 1
The correction may be made in the next voltage command calculation circuit. By the way, since the error current Ierr expressed by equation (39) is not zero unless the actual value of the primary magnetic flux of the induction motor 1 matches the set value L1I1d*,
Alternatively, in the correction voltage calculation circuit shown in FIG. 6, if the gains Kcd and Kcq of amplifiers 65 and 67 are set sufficiently high, or if a PI calculation type amplifier is used, the voltage drop due to the primary resistor R1 can be Even without correction using the primary current as in the example described above, since the error current Ierr is controlled to be almost zero, the actual value of the primary magnetic flux almost matches the set value L1I1d*. do. Therefore, in this case, it is not necessary to correct the voltage drop caused by the primary resistor R1 using the primary current. Furthermore, in the no-load voltage calculation circuit, 1
The voltage drop caused by the secondary resistor R1 may be corrected in advance. As described above, in this invention, the actual value of the primary magnetic flux generated inside the induction motor is given by the product of the excitation current command value and the primary self-inductance of the induction motor. The system is configured to calculate an error current component that becomes zero when it matches a set value of the secondary magnetic flux from the primary current of the induction motor, and to correct the primary voltage command value so that the error current component approaches zero. Therefore, the primary magnetic flux is controlled so as to match the set value even during low-speed rotation, which solves the problems of conventional devices such as insufficient torque and overcurrent. In addition, since the primary magnetic flux of the induction motor is controlled to match the set value not only during low speed rotation but also in the entire speed range, it is possible to First, the rotational speed of the induction motor can always be controlled stably. Furthermore, since the configuration is configured to calculate the deviation between the actual value of the primary magnetic flux and the set value as a current error, there is no need to directly detect the actual value of the primary magnetic flux, so the control circuit configuration is simple and the control device This has the effect of making it cheaper.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing an entire embodiment of the present invention.

FIG. 2 is a block diagram showing the configuration of a no-load voltage calculation circuit according to an embodiment of the present invention.

FIG. 3 is a block diagram showing the configuration of an error current component calculation circuit according to an embodiment of the present invention.

FIG. 4 is a block diagram showing the configuration of a correction voltage calculation circuit according to an embodiment of the present invention.

FIG. 5 is a block diagram showing the configuration of a primary voltage command calculation circuit according to an embodiment of the present invention.

FIG. 6 is a block diagram showing the configuration of a correction voltage calculation circuit according to another embodiment of the invention.

FIG. 7 is a block diagram showing the configuration of a primary voltage command calculation circuit according to another embodiment of the present invention.

FIG. 8 is a block diagram showing the configuration of a conventional device.

FIG. 9 is a T-shaped equivalent circuit per phase of an induction motor.

FIG. 10 is a pattern diagram of a function generator of a conventional device.

[Explanation of symbols]

1 Induction motor 2 Current detector 3. Variable frequency power conversion circuit 4 Excitation current command setter 5 No-load voltage calculation circuit 6 Error current component calculation circuit 7 Correction voltage calculation circuit 8 Primary voltage command calculation circuit 22 Frequency command generator

Claims (1)

[Claims] 1. An induction motor, a current detector for detecting a primary current of the induction motor, a variable frequency power conversion circuit for driving the induction motor at a variable frequency, and a primary frequency command value and an excitation current. a no-load voltage calculation circuit that inputs a command value and outputs a no-load voltage command value for the induction motor; and a no-load voltage calculation circuit that inputs the primary current, the primary frequency command value, and the excitation current command value and outputs the no-load voltage command value of the induction motor. An error current component that becomes zero when the actual value of the internally generated primary magnetic flux matches the set value of the primary magnetic flux given by the product of the excitation current command value and the primary self-inductance of the induction motor. and a correction voltage calculation circuit that calculates a correction voltage that brings the value of the error current component closer to zero by inputting the primary frequency command value and the output of the error current component calculation circuit. a primary voltage that calculates a primary voltage command value of the induction motor by inputting the primary frequency command value, the no-load voltage command value, and the correction voltage, and outputs it to the variable frequency power conversion circuit; 1. A control device for an induction motor, comprising a command calculation circuit.