JPH04207801A - Digital fm signal demodulator - Google Patents

Abstract

PURPOSE:To attain the application of a low speed A/D converter by dividing an FM modulated wave into an in-phase component and an orthogonal component by a hybrid divider and setting a sampling period at an integer multiple of the period set in a non-modulation state of an input FM modulated wave when both in-phase and orthogonal components are converted into the digital signals. CONSTITUTION:The sampling cycles of A/D converters 321 and 322 are set at an integer multiple of the cycle set in a non-modulation state of an input FM modulated wave. Then the in-phase and orthogonal components are converted into the digital signals in the relevant sampling cycles respectively. The carrier wave component of the output of a differentiation circuit 34 is set at zero when the sampling cycle is set at an integer multiple of the carrier wave cycle. The carrier wave component is eliminated out of the output of the circuit 34 and therefore the digital FM demodulating operation is attained in a sampling frequency lower than the frequency of an input signal SIN.

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] (Industrial Application Field) The present invention relates to a digital FM signal demodulation device that demodulates FM modulated waves in a digital format.

(Prior Art) A typical example of a conventional FM signal demodulator is the configuration shown in FIG. FM modulated wave SI input to this device
N is divided into two systems by an IF distributor 11, and tuned circuits 12 . ．． It is guided to 12 degrees. Each tuning circuit 1'21, 122 is shown in FIG.
) has the frequency characteristics shown in fo is F FvI
This is the carrier frequency of the modulated wave SIN. Each tuning circuit 12□
.

The frequency signal obtained at 12° is passed through detection diodes 13, .
13□, the signals are detected with opposite polarities, and added by the adder 14. That is, this adder 14 is as shown in FIG.
) will have the character-8 demodulation characteristics shown in (). The output of the adder 14 is filtered by a low-pass filter 15 to remove unnecessary high frequency components, and is then converted into an FM demodulated signal S. Output as LIT. However, although this method has a simple circuit, it has the disadvantage that it is difficult to obtain good demodulation characteristics because it is affected by the balance of the tuning circuit and the nonlinearity of the detector.

On the other hand, as shown in FIG. 8, the FM modulated wave SIN is converted into a rectangular wave by the rectangular wave converter 21, and the monostable multivibrator 22 is operated at the edge of the FM modulated wave SIN, and the rectangular wave is converted into a rectangular wave with a constant width as shown in FIG. A method of generating a pulse train and extracting a DC component using a low-pass filter 23 has been put into practical use. Furthermore, Figure 9 (
(a) shows the waveform of the modulation signal, and (b) shows the carrier waveform (a).
) 1 to F when F LI modulation is performed using the modulation signal shown in the figure.
Change: Waveform of A wave s1, the same figure (c) is the square wave converter 21
The output waveform of monostable multivibrator 2 (d) is the output waveform of monostable multivibrator 2.
FIG. 2(e) shows the output waveform (SOUT) of the low-pass filter 2B. Although this method has a simple circuit configuration, it requires digital elements that operate at a considerably higher frequency than the frequency of the input signal SIN, and the circuit needs to be adjusted because the rectangular wave converter 21, monostable multivibrator 22, or analog operation is required. However, it has disadvantages such as being easily affected by temperature changes.

However, in recent years, with the increase in speed of A/D converters, the first
A digital FM signal demodulator as shown in FIG. 0 has been proposed. In this device, an input FM modulated wave SIN is guided to a hybrid distributor 31, where the in-phase component (oo
) and orthogonal component (90°), respectively.
D converter 32. ．． It is converted into a digital signal at 32 degrees and input to an arctangent circuit (phase angle calculator) 33. The arctangent circuit 33 takes the in-phase component at the time of sampling as sir+θ and the orthogonal component as eOsθ from the data after A/D conversion, and calculates the phase angle θ by calculating tan−′θ.
(1) is sought.

The output of this arctangent circuit 33 is output from the 1-sample delay circuit 34.
□ and a subtraction circuit 342 to the differentiation circuit 34 .

This differentiating circuit 34 delays the input phase angle data θ(To) by one sample in the delay circuit 34.
)), the values before and after the delay are subtracted by the subtraction circuit 342, and the change numerator (T,)(-θ(T,)-θ(T,-1))
This is what we seek. The output of the differentiating circuit 34 is filtered by a subtracting circuit 35 to remove frequency components specific to the carrier wave, and then the output is D
/A converter 36 converts the FM demodulated signal S into an analog signal. Output as U↑. This method is superior to the above two methods in terms of operational stability and demodulation characteristics because all signal processing is digital.

To explain more specifically, the input signal SIN is defined as follows.

S IN-sin ((IJ. ten Δω) t,'
-(1) However, ω0: Carrier frequency Δω: Frequency change due to modulation signal In this case, the output of the arctangent circuit 33 is given as (ω.1Δω)t−(2). Here, the A/D converter 32□.

322 sampling time is T. ,T.

Tn, and the interval is ΔT, the output of the arctangent circuit 33 is θ ("ro > , θ (T+), . . .
・, θ (T, ,)...(3) It is given by: However, θ(T) represents the phase angle of the input signal at time T. Therefore, the output of the differentiating circuit 34, Sa (T), is given by the following equation.

5B(Tfi)-θ(T,)-〇(T,,)-(ω0+
Δω)(T,, -1,000 ΔT)-(ω.+ΔωIT,.

One ω.・ΔT+Δω・ΔT...(4)(4)
The first term in the equation indicates the amount of phase angle change due to the carrier frequency component. ω0. ΔT is a constant value and becomes a constant. Therefore, the subtraction circuit 35 calculates ω. By subtracting ΔT, a component Δω·6 (ΔT is a constant) proportional to the modulation signal can be extracted.

However, in the digital FM signal demodulator according to the above method, in order to sample the input signal correctly, the frequency f of the sampling signal for A/D conversion needs to satisfy f 5 > 2 f o - (5). Yes (Shannon's sampling theorem). For example, for FM-TV signal, f0=140
MHz, and f5>2801vlHz. In this case, an extremely high-speed A/D converter is required, making the device expensive. Also, as shown in equation (4), a constant term ω is included in the difference calculation output.・Is ΔT included?
Usually ω. 〉〉Δω, there is a problem that the number of effective digits of the subtraction circuit 35 becomes larger than necessary.

(Problem to be solved by the invention) As mentioned above, in the conventional digital FM demodulator,
This requires an extremely high-speed A/D converter, making the device expensive. Another problem is that the number of effective digits in the subtraction circuit becomes larger than necessary.

This invention is the second invention to solve the above problem, and uses a relatively low-speed A/D converter to exhibit excellent demodulation characteristics, and easily removes unnecessary carrier-specific frequency components. It is an object of the present invention to provide a digital FM demodulation device that can be used in a digital manner and is economically advantageous.

[Configuration of the invention (means for solving the problem) In order to achieve the above object, the digital F according to this invention
The M demodulator includes a hybrid divider that inputs an FM modulated wave and divides it into an in-phase component and a quadrature component, and sets a sampling period to an integral multiple of the period when the input FM modulated wave is unmodulated, and performs the sampling. Analog/digital conversion means for periodically converting the in-phase component and quadrature component into digital signals, and phase angle calculation means for calculating the phase angle of the input signal at the time of sampling from the data after analog/digital conversion by this means; The apparatus is configured to include a difference calculating means for demodulating the modulated signal of the FM modulated wave by calculating the difference between the phase angles of the in-phase component and the quadrature component obtained by this means.

(Operation) In the digital FM demodulator having the above configuration, the hybrid distributor divides the FM modulated wave into in-phase components and quadrature components, converts these in-phase components and quadrature components into digital signals, and converts the converted data into digital signals. Calculate the phase angle of the input signal at the sampling time and find the difference, F
Demodulate the modulated signal of the M modulated wave. At this time, by setting the sampling period for converting to a digital signal to an integral multiple of the period when the input FM modulated wave is unmodulated, a slower A/D converter than conventional ones can be used. In addition, a subtraction circuit for removing the frequency component specific to the carrier wave, which was conventionally required, is no longer necessary, and excellent FM demodulation characteristics can be obtained while simplifying the circuit.

(Example) The present invention will be described in detail below with reference to FIGS. 1 to 5. However, in FIG. 1, the same parts as in FIG. 10 are designated by the same reference numerals, and the different parts will be mainly described here.

FIG. 1 shows an embodiment of the present invention, in which the subtraction circuit 35 of FIG. 10 is omitted.

In addition, the A/D converter 321. :322, its sampling period is set to an integral multiple of the period when the input FM modulated wave is unmodulated, and the in-phase component and quadrature component are each converted into a dental signal at the sampling period.

The operating principle of the above configuration will be explained below with reference to FIG. 2.

In FIG. 2, the sampling frequency f is 50 MHz and the input carrier frequency f is 50 MHz. (-ω./2π)-150MHz
The operation of each part is shown when the size is reduced to 1/3. Here, the spectrum spread due to FM modulation is ±15MH
Assume that it is less than or equal to z. The left side of FIG. 2 shows the spectrum of each part of the circuit, and the right side shows the time axis waveform. (a) to (d) show the case where the input signal is not modulated, and (e) to (i) show the case where the input signal is frequency modulated.

In the case of no modulation, the input signal SIN is assumed as follows.

S IN−sin tωot+φ0) −5in(2πfot+φO) −(e) However, φ0
indicates the input signal phase angle when 1-0. on the other hand,
The sampling period 1/f5 is the period 1/f of the unmodulated input signal.
Since it is assumed to be f003 times, the following equation holds true.

ΔT=1/fs=3/fo−(
7) Therefore, the sampling time tゎ-t. , tl.

t2+... is given as follows.

t, -N-ΔT = 3 N/ fo ・-(
8) (However, N simplified 0, 1, 2...) By substituting equation (8) into equation (8), sampling time t
The input signal phase angle θ(to) at ゎ is given by the following equation.

θ(t,,)−2πfo−t,+φ.

−2πfo(N’ΔT)+φ0 12yr fO” 3N/fo1φ.

=3Nx (2π) ↑φ.

・(9) On the other hand, the phase angle φ. is 2π, as shown in Figure 2(b).
Since it is a periodic function whose period is , the following equation holds true.

M.(2π)+φ. −φ. ...(10)(
However, from equations (9) and (10), θ(1,) is given by the following equation.

θ (t7)-φ0 (where n-0, 1, 2,...) Therefore, the output SB (t,) of the differentiating circuit 34 is
SB (tゎ) 1θ(1,,)−〇(t,−1) 1θ. −θ. -0...(11) That is, by selecting the sampling period to be an integral multiple of the carrier wave period, the carrier wave component of the output of the differentiating circuit 34 can be made zero.

Next, to explain the case where the input signal SI is modulated,
N is defined as follows.

S IN” 5in (ωot+ΔF-sinf,,,t+φ0)...
・(12) The second term in the box in parentheses in the above equation indicates the phase angle due to the frequency modulation component. In this case, the phase angle θ(to) is given by the following equation.

θ(1,) 13N・(2π)10φ.

+ΔF 5in(2πf−−N−ΔT)−φ. +ΔF
5in (2πf, ・N-ΔT) (13) Therefore, the output Sa (t,) of the differentiator circuit 34 is expressed by the following formula 1
Formula % Here, if we compare formula (4) and formula (14), we get
The carrier wave component is removed from the output of the differentiating circuit 34,
It can be seen that the calculation range of the differentiator circuit 34 is effectively used. Furthermore, as is clear from the explanation of equations (6) to (14), digital FM demodulation operation is possible at a sampling frequency lower than the frequency of the input signal SIN.

Therefore, the digital FM demodulator having the above configuration includes an A/D converter 32. ．． 32□ can be used at a lower speed than the conventional one, and the subtraction circuit (10th
35) in the figure becomes unnecessary, and excellent FM demodulation characteristics can be obtained while simplifying the circuit.

By the way, the digital FM signal demodulator of the above embodiment eliminates the phase error generated in the hybrid distributor 31 and the A/D
Distortion occurs in the demodulated output due to the influence of amplitude errors occurring in the converters 321 and 322. The relationship among the phase error, amplitude error, and demodulated output distortion will be explained below with reference to FIG.

Figure 3 shows the amplitude error of the in-phase component and quadrature component and the input signal SI.
The relationship between the phase angle of N and the calculation error is shown. In FIG. 3, coordinates (X, Y) indicate in-phase axis data and orthogonal axis data when there is no amplitude error. Here, regarding the relationship between the in-phase axis and the orthogonal axis, if an error occurs as follows: Δ θ −θ −θ′ −tan−′ (Y/ x) −tan−′ (Y′/
x')-tan""(tanθl-tan-'
(kΦtanθ) However, it becomes k-b/a. In other words, when the in-phase component is X1 and the orthogonal component is Y, if X-Y (15) holds, then the locus of (X + j
(indicated by the dotted line in the figure), and no phase angle error occurs. However, if the gains of the in-phase axis and the orthogonal axis are not equal,
The trajectory of (X+jY) becomes an ellipse (shown by the solid line in FIG. 3), and a phase error shown by the solid line in FIG. 3 occurs. Therefore, the phase error Δθ* (
nT) is given by the following formula.

Δθ, (nT) = θ (nT) −〇' (nT) -tan' fY (n T) /X (n T)
t-jan-'(Y'(nT)/X' (nT
) 1-tan-' (tanθ (nT)1-t
an-'fk-tan-'θ (nT)1 On the other hand, if the phase difference between the in-phase axis and the orthogonal axis is fi at 90°, the calculation error of the input signal phase angle and the difference from the orthogonal axis when the in-phase axis and the orthogonal axis are not orthogonal The relationship with the gap is shown in Figure 4.

In FIG. 4, coordinates (x, y) indicate in-phase and orthogonal data when the in-phase axes and orthogonal axes are orthogonal. Here, if the orthogonal axis is not perpendicular to the in-phase axis (X-axis) like the y' axis, but intersects at an angle of φ (φ<90° in the figure), x,
When a locus of (x, y) is drawn in the y' coordinate system, it becomes an ellipse as shown by the solid line in FIG. The data of the coordinates (X', Y'') corresponding to (X, Y) with this ellipse are given by the following equation.

X’ −X+Y cosφ y’ −y cosφ Therefore, the phase error Δθ is given by the following equation.

Δθ-θ-θ'-tan' (Y/X) -tan-'(Y'/
X')-tan-'(Y/X)-tan-' f
Y sinφ/(x+ycosφ)) In other words, when the angle φ between the in-phase axis and the orthogonal axis is 90°, the trajectory (X+jY) draws a circle as shown by the dotted line in the figure, but the trajectory when φ≠90° becomes an ellipse as shown by the solid line in the figure. Therefore, the phase error Δθ, (nT) at the sample point nT is given by the following equation.

80b (nT) = θ(nT)-θ' (nT) -tan-' (Y/X) -tan-' (Y stnφ/ (X + Y co
sφ))...(17) Also, since the demodulated output SB (nT) is given by equation (14), the error ΔSB (nT) in the demodulated output is given by the following equation.

Δ5s(nT) = (Δθ, (nT) + Δθb(nT)1−[Δθ, f(
n-1)T)+Δθゎ1(n-1)T)] ...(18) Figure 5 shows the configuration of a digital FM signal demodulator without demodulation distortion, which was made to correct the above drawbacks. It is something. However, in FIG. 5, the same parts as in FIG. 1 are given the same reference numerals, and their explanations will be omitted.

That is, this device has a data correction circuit 41 interposed between the arctangent circuit 33 and the differentiation circuit 34.

This data correction circuit 41 has a phase error Δθ, (nT) shown by equations (16) and (17).

It has a function of correcting Δθ, (nT). (113).

As is clear from equation (17), Δθ, (nT).

Δθ, (nT) is given as a function of the true value θ(nT) as shown in the following equation.

Δ θ (nT) = Δ θ , (nT) + Δ θ , (nT
)=f iθ (nT), φ)...(19
) In the above equation, since k2 φ takes a constant value regardless of the input data, it can be seen that Δθ(nT) is a function only of θ(nT).

The actual circuit 41 is a ROM or non-volatile RAM.
The output θ' of the arctangent circuit 33 may be measured by providing a known phase difference θ in advance using a memory according to the above, and the difference may be stored as correction data. This makes it possible to realize a distortion-free digital FM signal demodulation device.

[Effects of the Invention] As described above, according to the present invention, excellent demodulation characteristics can be achieved using a relatively low-speed A/D converter, and unnecessary frequency components specific to carrier waves can be easily removed. Therefore, it is possible to provide an economically advantageous digital FM demodulator.

[Brief explanation of the drawing]

Fig. 1 is a block circuit diagram showing an embodiment of the digital FM signal demodulation device according to the second invention, Fig. 2 is a waveform diagram for explaining the operation of the embodiment, and Fig. 3 is the same embodiment. Figure 4 shows the relationship between the amplitude error of the in-phase component and quadrature component and the calculation error of the phase angle of the input signal. 5 is a block circuit diagram showing another embodiment of the present invention, FIG. 6 is a block circuit diagram showing the configuration of a conventional FM signal demodulator, and FIG. is a characteristic diagram for explaining the operation of the device shown in FIG. 6, FIG. 8 is a block circuit diagram showing another configuration of the conventional device, and FIG. 9 is a waveform diagram for explaining the operation of the device shown in FIG. , FIG. 10 is a block circuit diagram showing the configuration of a conventional digital FM signal demodulator. SIN... Input FM modulated wave, 31... Hybrid distributor, 321.322... A/D converter, 33... Arctangent circuit, 34... Differential circuit, 34 l... Usample delay circuit, 34 □・・・Subtraction circuit, 35, Subtraction circuit, 3
6...D/A converter, Sou...FM demodulation signal, 4-digit data correction circuit. Applicant's agent Patent attorney Takehiko Suzue Figure 6 (b) Figure 7 Figure 8 Figure 9

Claims (3)

[Claims] (1) A hybrid distributor that inputs an FM modulated wave and divides it into an in-phase component and a quadrature component, and a sampling period that is set to an integral multiple of the period when the input FM modulated wave is unmodulated; Analog/digital conversion means for converting the in-phase component and quadrature component into digital signals, phase angle calculation means for calculating the phase angle of the input signal at the time of sampling from data after analog/digital conversion by this means, and this means A digital FM signal demodulation device comprising: a difference calculation means for demodulating the modulated signal of the FM modulated wave by calculating the difference between the phase angles of the in-phase component and the quadrature component obtained in . (2) Further, the device is interposed between the phase angle calculation means and the difference calculation means to calculate the unbalance of each amplitude value of the in-phase component and the quadrature component obtained by the phase angle calculation means and the phase error of each component. 2. The digital FM signal demodulation device according to claim 1, further comprising a correction means for correction. (3) The correction means is characterized in that it includes a memory, gives a known phase difference to the memory in advance, measures the output of the phase angle calculation means, and stores the difference as correction data. The digital FM signal demodulator according to claim 2.