JPH04197100A - Current control circuit for semiconductor power switch and generator control method employing the circuit - Google Patents

Abstract

PURPOSE:To obtain a current control circuit for continuously controlling the load current of an inductive load regardless of the conducting/nonconducting state of a semiconductor power switch without requiring a special external component by providing means for holding a current detection value under conducting state of the semiconductor power switch and means for correcting thus held current value to a value under nonconducting state. CONSTITUTION:When current control is performed, e.g. a load current if is controlled continuously to a constant level, flywheel current if (off) at the time of turn OFF of a semiconductor power switch 1 is detected. In other words, an analog switch 9 is turned ON when a PWM output e0 is turned ON and a current detection voltage value VKK(on) is outputted, as it is, as a sample and hold voltage Vff(on) whereas when the PWM output e0 is turned OFF, the analog switch 9 is turned OFF and the final value of the current detection voltage value VKK(on) is sampled and held in a capacitor 10 and then it is outputted as a sample and hold voltage Vff(off). The sample and hold voltage Vff(off) is then reduced through a discharge circuit 17 thus producing an output waveform 106 equivalent to the load current if(off).

Description

[Detailed Description of the Invention] [Industrial Application Field] The present invention relates to a system for controlling the current of an inductive load,
The present invention relates to a current control circuit for a semiconductor power switch suitable for, for example, a vehicle generator control device, an inverter control device, and the like.

[Prior Art] Conventionally, in a device for controlling the current of an inductive load, as described in Japanese Patent Application Laid-Open No. 58-500046, the load current is detected, compared with a reference voltage, and output is determined when the semiconductor power switch is in a conductive state. Methods for limiting current are known. Further, a similar system is described in Japanese Patent Application Laid-open No. 104500/1983.

[Problem to be solved by the invention]

In all of the above conventional techniques, the load current is detected through a semiconductor power switch element, and the current value is intermittently limited only when it reaches a certain level compared with a reference voltage. In order to stably control the current so that fluctuations in the load current are reduced even due to ambient temperature or disturbances, it is desirable to control the current continuously. In the prior art, it is possible to continuously control the current by providing a detection resistor in series with the load or by using a current detection element such as a Hall element. However, there were problems such as an increase in the number of components in the system, making it difficult to integrate and implement the control circuit.

An object of the present invention is to provide a current control circuit that can continuously control the load current of an inductive load regardless of whether the semiconductor power switch is in conduction or non-conduction without adding any special external parts. .

Another object of the present invention is to provide a current control circuit for a semiconductor power switch that is suitable for integration.

Another object of the present invention is to provide a current control circuit for a semiconductor power switch that continuously controls the load current of a vehicle generator control device and has less 8-force fluctuations due to external disturbances.

[Means to solve the problem]

To achieve the above object, in order to use the load current value for current control even when the semiconductor power switch is in the non-conducting state, there is provided means for holding the detected current value in the conducting state, and correcting the held current value to the non-conducting state. A correction means is provided.

In order to achieve the other objects mentioned above, means is provided for artificially limiting the discharge time constant by the pulse width of the electronic switch.

[Effect]

If the detected value of the load current can be obtained as a continuous value regardless of the conduction or non-conduction state of the semiconductor power switch, the control loop will be stabilized because the current feedback value will not be intermittent, and malfunctions due to errors in the average current will be eliminated. I can do it.

〔Example〕

Hereinafter, five embodiments of the present invention will be explained with reference to FIG.

The semiconductor power switch 1 is connected in series with the inductive load 3 and the current detection resistor 6e to the + side of the DC power supply 4. In this embodiment, the inductive load 3 is connected to the high potential side of the semiconductor power switch 1, and the current detection resistor 6e is connected to the positive side of the semiconductor power switch 1. is connected to the low potential side. In particular, a flywheel diode 2 is connected in parallel with the inductive load 3. The connection of the load is not particularly limited, whether it is on the high potential side or the low potential side. In this embodiment, the semiconductor power switch 1 is an N-channel power MO3F.
Although the description will be made using an ET, it may be a P-channel power MO5FET, a bipolar element, or an IGBT that is a composite element of a MOSFET and a bipolar element. The inductive load 3 electrically consists of an L component and an R component, and has a time constant of L/R.

The current detection circuit 8 uses a low resistance of 0.1 ohm or less for the current detection resistor 6e, and the amplifier 7 and the network resistor 6a.
, 6b, 6e, and 6c, the element current ich flowing through the semiconductor power switch 1 is detected as the output voltage value vkk of the amplifier 7. The output of the amplifier 7 is fed back to the drive circuit 5 of the semiconductor power switch 1 via the sample and hold circuit 13. Sample hold circuit 13
are an analog switch 9 driven by the PWM output eo of the drive circuit 5, a capacitor 1o that stores the output voltage value vkk of the amplifier 7, and an FET input buffer that feeds back the sample and hold voltage V□ to the drive circuit 5 as an output. An amplifier 11 is used. The voltage value of the sample hold voltage Via is adjusted by the discharge circuit 17.

The PWM power eo that drives the semiconductor power switch 1 is
The drive circuit 5 calculates the sample hold voltage V x t as a current feedback value, and the result is output as an on/off pulse waveform. In this embodiment, the discharge circuit 1
7 is a discharge resistor 1 with a high resistance value of several kilograms to several megaohms.
2, the time constant of the inductive load 3 and the time constant of the CR of the capacitor 1o and the discharge resistor 12 may be matched.

Next, the detailed operation of the present invention will be explained with reference to FIG. P.W.
The M output eo is output as an on/off pulse waveform 101 at a constant frequency or variable frequency. Like the pulse waveform 101, the element current ich flowing through the semiconductor power switch 1 is zero when off, but because of the inductive load 3, when on, it becomes a current waveform 102 that increases at a constant time constant. The output waveform 103 of the voltage value V h k converted by the current detection circuit 8 is also similar to the current waveform 102 . When the flywheel diode 2 is connected as in this embodiment, the load current i flowing through the inductive load 3 is equal to the load current i flowing through the semiconductor power switch 1.
When on, the current 1z(on) increases with a constant time constant like the element current ich, and when off, the current 1f(of) decreases with a constant time constant as a flywheel current.
f). As a result, the load current it becomes a triangular waveform 104 that repeatedly increases and decreases.

When performing current control such as continuously controlling the load current j to a constant value, it is necessary to detect the flywheel current 1i (off) when the semiconductor power switch 1 is off. In this embodiment, when the PWM output eo is on, the analog switch 9 is turned on, and the current detection voltage value Vkk (on
) is output as it is as the sample-hold voltage Vzf (on), and when it is off, the analog switch 9 is turned off, the final value of the current detection voltage value vkk (on) is sampled, and after being held in the capacitor 10, the sample-and-hold voltage Vzi
(off). In order to reduce the size of the capacitor 10 and reduce leakage current during hold, it is desirable to use an FET input amplifier for the buffer amplifier 11, and a CMO8 configuration may be used.

The output waveform 105 of the sample hold voltage Vff has a detection error of 201.2 because the hold voltage is held for a long time.
02 and 203 occur, and characteristic 4: detection error 203 tends to increase in region A where the off time T (off) of PWM output eo is long. In the present invention, especially the sample and hold voltage vi
z(off) is reduced by the discharge circuit 17 to realize an output waveform 106 equivalent to the load current 1t(off).

According to this embodiment, there is an effect that a continuous current detection waveform with less detection error of load current can be obtained.

Next, a second embodiment of the present invention will be explained with reference to FIG.

This semiconductor power switch 1, which is an application of the present invention to an integrated circuit, has a current detection circuit 8. Sample hold circuit 13. The discharge circuit 17 and the drive circuit 5 are integrated on the same board as a power ICl3. In integrated circuits, it is particularly difficult to obtain large capacitance capacitors of several hundred microfarads or more and high resistances of several hundred ohms or more. For example, if the time constant of the inductive load 3 is 50m5 and the value of the capacitor 10 is 50pf, normally the discharge resistance 1
2 requires 1000M ohm and cannot be integrated.

An analog switch 9 is provided on the high potential side of the discharge resistor 12,
If the oscillator 14 discharges the charge stored in the capacitor 10 intermittently at the frequency fkx, a pseudo high resistance time constant of 1000M ohms can be realized with a resistance value of several hundred ohms.

Details will be explained with reference to FIG. The frequency fk1 generated by the oscillator 14 is expressed as a duty ratio, which is the ratio of the discharge time T+ to the holding time T2. It is a frequency with a constant duty ratio. It is desirable that the sample and hold voltage Vii be equal to 'ip even when the PWM output eo is off. At this time, the sample-and-hold voltage V i i decreases by V due to the discharge resistor 12 during T1. After Vr decreases, a pseudo-long discharge time constant equal to 11 can be obtained by holding the voltage again for a period of T2. This discharge time constant is dominated only by the duty ratio and the discharge resistance value, and is not affected by the deviation of the frequency fhs. A long time constant can be easily obtained by shortening the discharge time T1 and lengthening the holding time T2. By using a clock frequency for the discharge time T1 and a frequency obtained by dividing the clock frequency by a flip-flop for the holding time T2, a frequency with a stable duty ratio can be easily realized. By changing the branch bit and changing the duty ratio, it is possible to match the time constant of the inductive load 3 without using adjustment means such as changing the discharge resistance value.

In the embodiment illustrated in FIG. 3, the discharge resistor 12 can also be realized by a constant current source. FIG. 5 shows an embodiment in which the discharge resistor 12 is realized using a constant current source.

Resistance values in the hundreds of ohms in integrated circuits are easier to obtain by using an active load with a constant current source. In this embodiment, an example using a current mirror in which the area ratio of MOS transistors is changed will be described, but bipolar transistors may also be used. A reference current I is applied to an nMOS transistor 21 diode-connected in series with a resistor 20.
ret is playing. This is typically tens to hundreds of microamperes of current. By changing the area ratio of the transistors, the nMOS transistor 22 can be
A minute current 1/N of that of the S transistor 21 can be caused to flow, resulting in an equivalently larger resistance value. According to this embodiment, the constant current source is used together with the analog switch 15.
If it is constructed using S transistors, there is an advantage that the circuit area can be particularly reduced during integration.

Next, a third embodiment of the present invention will be described with reference to FIG.

This is regardless of analog correction methods.

The data is digitized and corrected using logical operations. After outputting the current detection voltage value vkk as a sample and hold voltage Vit in the sample and hold circuit 13,
It is digitized by an A/D converter 51, and the digital amount is corrected by an arithmetic circuit 52 to obtain a continuous current detection value. In this embodiment, if the time constant of the inductive load 3 is prepared as a table in the arithmetic circuit 52, there is an advantage that even if the time constant of the inductive load 3 changes, it can be easily handled with few changes to the circuit.

Next, a fourth embodiment of the present invention will be described with reference to FIG.

In this embodiment, the current detection circuit 8 is an example in which the semiconductor power switch 1 is provided with a mirror current detection terminal. The mirror current j is the element current i of the semiconductor power switch 1
The current value corresponds to the cell ratio of the channel. Current detection resistor 6e
is connected to the mirror current i, so the element current 1
No power loss occurs when used in series with a channel. The mirror current i is converted into a current detection voltage value V h h by the current detection resistor 6e and used in the same manner as in the previous embodiments.

Further, in this embodiment, the current detection resistor 6e and the capacitor 10. A reference voltage V r e i generated by a constant voltage source 60 on the low potential side of the constant current source 16 is set as a virtual ground potential. The constant voltage source 60 is well known as a constant voltage source, for example. Resistors 61, 62, 63 and transistors 64, 65
．． A bandgap reference circuit consisting of 66 circuits or the like is sufficient. Details will be explained with reference to FIG. When accurately controlling the current of the semiconductor power switch l even in a small load current range, the dead zone of the buffer amplifier 11 of the sample and hold circuit 13 in the low input voltage range becomes a problem. This is limited by the power supply voltage of the buffer amplifier 11, and occurs because input/output is performed with the ground voltage E as a reference. Input voltage■. , the reference voltage V r
If only e t has an offset voltage, the effect of the dead zone in the low input voltage range of the buffer amplifier 11 can be eliminated. Reference voltage V due to increase/decrease in mirror current i
In order to eliminate the voltage fluctuation of r e i, the current source 67
Therefore, it is sufficient to flow the bias current ■1. According to this embodiment, a current control circuit with particularly excellent aspect ratio accuracy can be realized.

Next, a fifth embodiment of the present invention will be described with reference to FIG.

This is an example of an IC regulator 80 having a function of controlling load current of an alternator, which is a generator control device for a vehicle. By having a current control loop within the original voltage control control loop, it is possible to limit the load current and achieve voltage control with little output fluctuation. Further, according to this embodiment, the sample hold circuit 13. Discharge circuit 17
Since the load current when the semiconductor power switch 1 is turned off can be detected using the above, there is an advantage that the detection error of the sample-and-hold voltage V i i is small and the current control loop operates stably.

Next, a sixth embodiment of the present invention will be described with reference to FIG. This shows an example of a layout in which the present invention is integrated as a power ICl3 on the same substrate. In order to reduce the thermal influence on the semiconductor power switch 1, the capacitor 1o of the sample and hold circuit is arranged in parallel at a distance of at least 100 microns or more, and the discharge circuit 17 is placed on the opposite side of the semiconductor power switch 1. Place.

According to this embodiment, there is an advantage that the error in the discharge current due to the thermal influence of the semiconductor power switch 1 can be reduced.

Next, a technique for making the discharge time constant τ of the sample-and-hold capacitor match the attenuation time constant τ0 of the current flowing through the inductive load will be explained.

The sample-hold capacitor discharge time constant τ is τ=□
...(1)(α・xe) It is expressed as follows.

Here, C: Capacitance α of the sample hole capacitor: On-duty of the oscillator output signal T I + T z E: Drawing current (constant current) Now, the decay time constant τ0 of the current flowing through the inductive load is 50m5
Suppose that

C = 50 pF, Ie = 2, 05 m
If A, a = □, then the following can be obtained.

If the time constant of the inductive load is different and is 100m5, then
Inside the IC, adjustment can be made by changing ■8 or α.

In the case of Ie (Ie = 1.02 μA; half of the previous value.

) It is particularly desirable to change the duty α, which is easy to modify, inside the IC circuit. If α is generated by dividing the clock, the frequency division bits may be shifted. Normally, the aluminum pattern (wiring) should have 8 bit selections.

You can also connect the terminal to the outside and select from the outside.

Finally, an example in which the present invention is applied to the control of an automobile generator, and in particular an example in which the present invention is applied to an intelligent IC regulator, will be explained in detail.

As shown in Fig. 11, a generator for an automobile has various loads, such as a battery 4 that provides direct current, and a direct current load that uses this battery as a power source, or an alternating current load that uses the alternating current output of the generator as a direct power source. It is connected. Naturally, the battery itself is also one of the loads of the generator 7a.

The generator 7a is driven by an automobile engine and outputs three-phase AC power. This AC power is rectified by a rectifier 70a and supplied to the battery 4. A plurality of DC load groups are connected to the battery 4 via switch groups. Loads include car air conditioners, lighting equipment, sound and video equipment, electromagnetic equipment for fuel control, and daylight gas.

In addition, a load that uses the AC output of the generator as a direct power source may be connected. For example, there are quick clear glass systems that quickly melt ice from windows.

The generator 1 has a field winding 3, and by controlling the current flowing through the field winding, the output voltage (current) of the generator is sufficient to maintain the voltage of the battery 4 at a predetermined value. Control the generator as you get.

Note that 2 is a flywheel diode.

Control of field winding current will be explained below.

The voltage of the battery 4 is detected by the voltage detection circuit 130. The signal Vaa corresponding to the detected voltage is the battery setting voltage (
14,6±0, 25 V) Vac, and the deviation is amplified by the deviation amplifier 120 to output the voltage deviation signal E2.

The voltage-current command value conversion circuit 110 outputs a current command value Izi corresponding to the field current (target field current) required to maintain the battery voltage at the set voltage according to the voltage deviation signal ε2.

The switching circuit 170 receives a current command value 1t2. from the initial excitation circuit 140, which will be described later. Current command value from load response control circuit, 3. Which current command value from the current command value If+ from the temperature detection circuit 160 is to be output as the target current command value Izo is selected and switched.

The deviation amplification circuit 100 compares the target current command value Izo and the actual current value signal Iff from the field current detection circuit 8, which will be described later, amplifies the deviation, and outputs a current deviation signal E1 as the final current command value. .

The current supply circuit 70 is, for example, a PWM (Pulse Wi
dth Modulation) control circuit and, for example, an FET (field effect transistor) driven by this output, chopper-controls the field winding current icH with a duty according to the current deviation signal εl.

The current detection circuit 90 detects the current flowing there from the terminal voltage of the current detection resistor R connected in series with the field winding circuit, and outputs a constant current signal Itz in accordance with the detected current.

There are two types of current sources for the field current: a DC current rectified by the rectifier 70a and a DC power source from a battery.During normal operation, the field current is self-excited by the output current of the rectifier 70a.

When the rotational speed NG of the generator is low, such as when starting the engine, sufficient generated current cannot be obtained, so current is supplied from the battery 4 at this time.

The initial excitation circuit 140 sets the field current to the minimum necessary value as shown in FIG. 14 when the engine speed is lower than the predetermined value NGO and the driving torque of the generator places a burden on the engine. It has a function of setting the current command value Ifm to Izi to make the current command value Ifm.

The load response circuit 150 detects the application of a load by a sudden change in battery voltage, and when the engine speed is low such as idling speed, the current command value is changed over a period of 2 to 3 seconds as shown in FIG. A ramp-shaped current command value Izg that gradually increases to a target current command value 11& is output.

The temperature detection circuit 160 detects the temperature of the semiconductor switching element for the chopper, and when this temperature exceeds a predetermined value Ta, the current command value I is set as shown in FIG. Outputs the command value No. 5 that decreases according to the temperature.

The basic idea of the present invention will be explained based on the embodiments described above.

That is, the field winding current command value generating means A generates a signal Iz corresponding to the voltage deviation E2 between the battery voltage and a predetermined set voltage.
5. Signal equipment from o and field current signal generating means B. A field current command value E1 is generated based on this current command value El.
Based on this, a predetermined current is supplied from the field winding current supply means C to the field winding.

With this configuration, when the load connected to the battery is turned on and the battery voltage drops, the current command value ε1 increases commensurately, and the field winding current icH
increases. As a result, the output voltage (current) of the generator increases and the battery is charged to a predetermined voltage.

In this state, if the temperature of the field winding increases and the resistance value increases due to the influence of temperature, the field current will stop flowing and the current will drop unexpectedly.

However, when the current is about to decrease, the current command value increases and the supply amount is automatically increased, so the output of the generator does not change even if the resistance value of the field winding increases, and the load (battery) It is possible to maintain output according to the demands of

The specific circuit diagram shown in FIG. 16 will be explained below.

The same reference numerals indicate corresponding parts throughout the drawings. 71 is a chopper consisting of a switching element such as a power transistor or FET that controls the switching of the current flowing through the field winding 2; 170 is a constant voltage power supply device that supplies power supply voltage VCC to each of the above control circuits; and 180 is a DC load. . The other configurations are the same as in FIG. 1. In the voltage-current command value conversion circuit 110, Rz and R2 are voltage dividing resistors, which divide the output power supply voltage Vcc of the constant voltage power supply circuit 170 and supply the voltage to the battery 4.
Outputs the set value Vac of the charging voltage.

Rs and R4 are input voltage dividing resistors that feed back the battery voltage VB. AX is an operational amplifier, and input resistors R4 to R
6 and a feedback resistor R7, forming a deviation amplifier. In the current control circuit 100, A2 is an operational amplifier with input resistors Rs and Re.

RIO and a feedback resistor Rzz, and receives the current command Ill from the voltage control circuit 110 or the command value I f2 from the auxiliary circuit. I f8. The command value Izo selected from Ixi and the field current detection circuit output Il
This is an operational amplifier that calculates the deviation from l. P! In the t1M control circuit 70, A8 is an operational amplifier with an input resistance Rs.
x+ Ria, R14 and feedback capacitor C] constitute an integrator, perform integral operation on the input voltage, and
Addition and subtraction are performed between the input signal ε1 input through the input resistor Rsa and the voltage eo input through another input resistor Rzz. A4 in the latter stage is also an operational amplifier, which inputs the output ei of the integrator to the positive terminal via the input resistor R13, and also feeds back the output eo to the positive terminal via the feedback resistor Rs6 to provide hysteresis. Configure the comparator. The operating level of this comparator A4 is determined by dividing the power supply voltage Vcc by voltage dividing resistors R17 and Rls and applying it to the negative terminal via an input resistor R19. A combination of an integrator and a comparator having the above circuit configuration operates as a self-excited oscillator that outputs a square wave when the output eo of the comparator is fed back to the input of the integrator. In other words, PW whose duty changes in proportion to the input voltage E1
Functions as an M control circuit.

Next, 71 is a chopper, and a chopper circuit is constituted by a power transistor TI as a switching element, a driver transistor T2, a flywheel diode Di, a shunt resistor 8 for current detection of the power transistor T1, etc., and a current flowing through the field winding 2. switching of if is controlled by the output signal eo of the PWM control circuit. In addition to the chopper elements mentioned above, there are switching elements such as FETs.

Any means may be used.

9o is a current detection circuit. A5 is an operational amplifier, which is composed of input resistors R20-R22+feedback resistor R28. 9
1 is an analog switch and is driven by the PWM control output eo of 7 via a buffer 92. C2 is an output voltage holding capacitor.

Next, the operation of each part in the above configuration will be explained.

First, the operation of the field current detection circuit 90 will be described below.

FIG. 17 is a block diagram of the current detection circuit 90, and FIG. 18 shows operating waveforms of each part. The current detection by the current detection circuit detects the current icH of the power element which is an intermittent current as shown in FIG.

That is, the chopper current ic+-+ is detected by the shunt resistor R and amplified by the operational amplifier A5 to produce a vcH signal.

The chopper detection signal VCH is sampled and held by the analog switch 91 and hold capacitor C2 circuit to generate a simulated field current signal ■. is converted to

To explain in more detail, P of the output of the PWM control circuit 70
Analog switch 91 is set to 0F in synchronization with WM signal eo.
The chopper current icH during the OFF period of FL and the chopper is held at the current value immediately before the chopper is turned OFF, and the detection signal at this time is set as the V i 5 signal. Further, while the chopper is ON, the analog switch 91 is turned on so that the detection signal VC)I of the ONL chopper current icH is directly used as the V x i signal. Note that the ON/OFF operation of the analog switch 91 is performed via the buffer 92 in accordance with the above-mentioned PWM control signal eo.

Resistor 6e, oscillator 14. The operation of the discharge circuit of the capacitor C2, which is composed of the switching element 15, the constant current source 16, and the capacitor C2, is as explained in detail in FIGS. 7 and 9.

With the above operation, as shown in FIG. 18, the waveform of the simulated field current detection voltage Vtz obtained from the chopper current icH is not interrupted and becomes an operating waveform close to the field current j. As a result, the static characteristics of the field current detection circuit are
As shown in Fig. 9, characteristics with good linearity can be obtained, and a wide range of field currents from small to large field currents can be detected.

Furthermore, since an insulated type detector is not required, the current detector can be constructed at low cost.

Next, the current control operation will be explained. Returning to FIG. 16, the PWM control circuit 70 is for PWM control of the chopper 71, and includes an integrator consisting of an amplifier As, an integrating capacitor Cz, an integrating resistor Rzz, etc., and a resistor that connects the output of the amplifier A4. It is composed of a comparator with positive feedback and hysteresis using RIB. Then, the output e of comparator A4
By feeding back O to the integral input resistor R12,
This becomes a PWM control circuit capable of duty control. Above P
The WM control circuit has a function of proportionally controlling the conduction duty (conduction rate) of the output signal eo with respect to the input signal (voltage) ε1.

The PWM input signal ε1 is provided by a 100 deviation amplifier. That is, in the deviation amplifier 100, the difference between the signal 10 from the voltage control circuit and the field current detection signal Izz is multiplied by a gain (G=Rss/Rs=R1o/
Ra) and output as the input signal E1 of the PWM control circuit 70.

Therefore, the current control includes 100 deviation amplifiers, 70 PWM control circuits, and 90 field current detection circuits.

This is carried out using a circuit consisting of 71 chopper circuits, 2 field windings, etc.

Now, when the field current command Iio is given, the deviation amplifier 10
0, a deviation signal E1 obtained from the current feedback signal 1it is generated and applied to the PWM signal circuit 70.

The PWM control circuit 70 operates the chopper 71 using the output PWM signal eO to perform feedback control so that the field current ii matches the command value.

Therefore, the field current can be set arbitrarily by changing the current command value Io as shown in FIG.

Note that the PWM circuit shown in the figure is configured as a variable frequency PWM circuit.

Such a PWM control circuit is controlled so that the frequency reaches its maximum when eO is 50%, and the frequency decreases from that point regardless of whether eO is large or small, depending on eo, which indicates the conduction rate. The pulsating current rate can be suppressed within a certain narrow range.

Further, as shown in FIG. 21, even when the current command 1xo is suddenly changed, the field current it follows the command value. Therefore, when the present invention is used, for example, as shown in FIG. 22, in the case of conventional conductivity control, the driving torque of the generator increases due to temperature changes in the field winding resistance, and increases at low temperatures. It is a characteristic that shows a change that decreases over time.

As a result, the capacity of the field winding of the generator and the capacity of the chopper element had to be designed to withstand cold temperatures, resulting in overspec. However, by using the current control of the present invention, As shown in the figure, even if there is a difference in field winding resistance between cold and hot, the current can be controlled to the target value, so the influence of the difference in cold and hot temperatures does not appear.

Furthermore, it is not affected by current fluctuations caused by changes in power supply voltage, etc. Therefore, there is no need to design over-spec switching elements such as field windings and choppers of the alternator, and power can be increased. That is,
By amplifying the maximum operating value under normal conditions to the characteristics at low temperatures, the capacity can be increased by that amount, and the alternator can achieve high output. The amplification rate is several tens of percent, and the effect is great.

The operation of the voltage control circuit using the above-described current control circuit is as follows. Returning to FIG. 16, voltage control circuit 110
Now, feedback control is performed so that the actual battery voltage (generator output voltage) VB matches the battery charging voltage value Vsc. That is, the deviation amplifier A1 generates a deviation signal Iz between the battery set voltage Vsc and the battery voltage Vs.
o (current command) is output and given to the current control circuit 100. Then, as described above, the output signal El of the current control circuit 100 is generated. The PWM control circuit 70 generates eight ON/OFF PWM control (pulse width control) pulses eo in response to the output signal ε1, and intermittently applies the pulses to the field winding 3 of the generator 7a via the chopper 71. Apply voltage vi, field current j! control. In the above control operation, the field current i is detected by the shunt resistor R as described above and fed back to the current control circuit 100 via the current detection circuit 9 to perform current control. As a result, the armature winding output voltage of the generator 1 is controlled to charge the battery 4 and supply current to the load via the three-phase rectifier 3. The output voltage Va of the generator 7a is controlled by the voltage control circuit 110.
Feedback is performed and the output voltage is set to the battery setting voltage V.
Feedback control is performed to match ac.

Next, the peripheral technology of this embodiment will be explained based on FIG. 12.

1.? D: Bow Milk Path This circuit generates a basic clock of I MHz and a clock signal obtained by dividing the basic clock.

CL l drives a charge pump circuit with a basic clock of I MHz and charges a high voltage to the gate of FET1.

CL2 to CLso are clock signals obtained by dividing the frequency of CL l and supply clock signals for each timer circuit.

2. Mouth Ll This circuit detects the rotation speed of the generator and outputs a rotation speed signal for switching circuit operation.

The number of circuits is detected by the frequency f of the P terminal (one phase of the armature winding).
Since p is expressed as 6o・2 (N is the rotation speed of the generator (r, p, m); q is the number of poles of the generator; 2 is the constant during full-wave rectification), this frequency fp This is done by comparing the frequencies of the clock pulses CLθ and CLIO.

The N1 output becomes "1" when the generator is 500r, p, m or more, and becomes "O" when it is less than 500r, p, m.

N2 output is when the generator is 100 Or, p, +a or more.
1", and when it is less than 0, it becomes 0.

N3 output is "1" when the generator is 250 Or, p, m or more.
'', and when it is less than rOJ.

3. Electrification The role of this circuit is to prevent the field winding and armature winding from breaking.
To prevent the battery from being charged and eventually stalling when the FETI is opened and destroyed.1
When power generation is stopped (including when the engine is not rotating), the charge lamp lights up to notify you.

Its operation is to turn on the charge lamp when the generator is less than 100 Or, p, +u. When reaching 1000r, p, m, the charge lamp is turned off. When the engine speed drops again to below 50 Or, p, m, the charge lamp will turn on again.

The idle speed of the engine is 700 r, p, m.

If the pulley ratio between the crank pulley and the generator pulley is 2, the generator rotation speed at idle is 1400 r, p, m.
It is. Therefore, if the generator is normal, charging
The lamp goes out.

Note that when power is not being generated, the rotation speed is 0 and the charge lamp is lit.

The important point is that hysteresis is provided between N1 and N2. This has the effect that the lamp does not flash during cranking, etc., and does not make the driver feel anxious (Fig. 29).

(See Figure 30).

4.8 Terminal Open The role of this circuit is to prevent the generator from going uncontrolled when the S terminal (battery voltage detection terminal becomes open due to disconnection of the wiring, etc.).

■ It flashes the charging lamp and gives a warning to the driver6.

Its operation is as follows: (1) Normally, voltage control is performed by comparing the voltage at the S terminal with a reference voltage. When the S terminal becomes open, the battery voltage Vc decreases, and when it is below a certain value (7V), the terminal is switched from S to B by the S-S terminal voltage switching circuit.

■ At the same time, the flashing of the charge lamp turns the charge lamp on and off at 1 second intervals (second
(See Figure 1).

5.8 Open B The role of this circuit is that the S terminal (generator output cable)
■ Prevents the generator from going uncontrolled when it becomes open due to disconnected wiring, etc.

■ Flashes the charge lamp and gives a warning to the driver.

At the point.

If you continue to drive the car with the S terminal open, the battery will discharge because it is not charged, and eventually the engine will stall.

The operation is ■ If the S terminal is disconnected, the battery will not be charged, so the voltage at the S terminal will drop (approximately 11 to 12 V compared to 14.5 V under normal conditions), and as a result, the field current command value will increase. The S terminal voltage increases. As a result, when VB exceeds a certain value (18V), the voltage detection terminal is switched from the S terminal to the S terminal by the S-S terminal voltage switching circuit. This controls VB to 14.5V.

■ At the same time, the charge lamp flashes.

■ The operations from ■ to ■ are reset at fixed time intervals (1 minute). This is to restore the battery voltage (=Vs) to normal when the S terminal open state returns to normal state (see FIG. 22).

6. Electric port The role of this circuit is to issue an alarm if voltage control becomes uncontrollable for some reason.

Possible cases in which voltage control becomes uncontrollable in the unit:■ When the FETI is short-circuited and destroyed.■ When the S terminal and F terminal are shorted externally (when a piece of metal is caught between the terminals).

If operation continues without voltage control, (i) the battery may become overcharged, hydrogen gas may fill the engine compartment, and there is a risk of explosion.

(ii) Overvoltage occurs at high speeds, damaging on-vehicle electrical loads such as lamps and electronic devices.

Such problems may occur, but this can be prevented by notifying through this circuit.

When the operation is in the above mode, the field current command value becomes O, and the gate voltage of FETI becomes Ov continuously.
If the gate voltage remains Ov for a certain period of time (3 seconds) or more, it is determined that it is in overvoltage mode and the charge lamp blinks.

The blinking cycle is 0.25 seconds on and 0.25 seconds off (see Figure 33).

The role of this circuit is to flash the charge lamp when the S terminal is open, the S terminal is open, and the overvoltage 4 power generation is stopped.
The point is to issue the warning.

The operation is performed by calculating the logical sum (OR) of the above four signals and driving the gate of FET2.

What is important here is that the blinking period is set to an integer multiple for each event. This allows you to diagnose what is wrong by looking at the flashing pattern of the charge lamp. Furthermore, it is also possible to prioritize the lamp displays in descending order of importance. For example, the frequency may be lowered in the following order: ∎ Stopping power generation, ∎ Overvoltage, ∎ B terminal open, and ∎ S terminal open (see Fig. 24).

The role of this circuit is to prevent overcurrent from flowing through the FETI and destroying it when the field winding is short-circuited.

Its operation locks the gate of FETI when the voltage at the F terminal remains low even though eo is high.

The role of this circuit is to detect a state in which self-excited power generation is not possible when the rotational speed NG of the generator is low, such as the rotational speed N 1 (= 50 Or, p, m). Output the current command value Iz2 so that the chopper conductivity becomes about 30%,
Based on this, the target current command value Izo is output from the switching circuit.

10. S-B Electric Port The role of this circuit is to constantly feed back the S terminal voltage (voltage taken directly from the battery terminal) through the filter circuit, and when performing voltage control, if the S terminal is disconnected, The S terminal voltage (voltage extracted from the intermediate wiring between the generator and battery) is input and voltage control is continuously performed to prevent the generator from becoming uncharged to the battery.

The operation is performed by constantly inputting the S terminal voltage and the S terminal voltage.

When a signal from the S terminal open alarm circuit is generated, the detection terminal is switched from the S terminal to the S terminal. Also, S
When a signal is generated from the terminal open alarm circuit, the voltage signal is switched from the S terminal to the S terminal, and the S terminal voltage is output to the filter circuit.

The role of this circuit is to smooth out the rectified ripple voltage of the generator included in the S-S terminal voltage and stabilize the voltage feedback control.

Its operation uses a Miller integral low-pass filter to remove ripple voltage and output the average voltage of the battery.
Feedback the battery voltage to the voltage-current command value conversion circuit. As a result, the average value of the battery voltage can be detected with high accuracy, and the current command value 1tz can be used as a control signal that is not affected by ripples.

H constant i Gokai Hagi Converts battery voltage to a constant voltage of a predetermined value.

After that, it is supplied as a current to each control circuit.

13. - Electric' 8 B The role of this circuit is to generate a current command value Izx for controlling the field current of the alternator so that the terminal voltage of the battery becomes a constant value in accordance with the set value VBC of the battery voltage.

Its operation is based on the voltage command value VBC' from the set value switching circuit.
The deviation between the output VBC of the filter circuit and the output VBC of the filter circuit is taken and amplified by a gain factor to generate a current command value Ifil.

14. The role of this circuit is to set the internal reference value for setting the target voltage of the battery, that is, to lower the set value Vac and cut power generation when a signal from the power generation cut control circuit is generated. It is in.

Normally, the voltage setting value VBC is output as a voltage command value to the voltage-current command value conversion circuit, but when a signal from the power generation cut control circuit is generated, the voltage command value VBC is lowered than normal and power generation is performed. Make sure not to.

15. Power cut The role of this circuit is to reduce the drive torque of the generator (stop generation) when the load increases, such as when the vehicle accelerates.

Aim to improve acceleration.

Specifically, if the throttle opening detection switch SWI, which is connected in series with the charge lamp, opens at full throttle, power generation is cut for a period of time (for example, 10 seconds) until acceleration ends. .

As for the operation, since the drain voltage of FET 2 for lamp lighting is used as the voltage detection terminal for power generation cut detection, lamp lighting and power generation cut detection are shared.

That is, the power generation cut control circuit inputs the drain-source voltage VDS of FET2 and the current 1os flowing through the detection resistor Rgz of FET2. now.

When SWI opens, Vos of FET2 decreases, and if no current flows through FET2, the set value is changed to the set value switching circuit in order to cut power generation during the acceleration time of the vehicle (approximately 10 seconds). It outputs a switching signal and also generates a chopper gate lock signal to the gate lock circuit.

16. Measures゛ The role of this circuit is to control the maximum power generation of the generator with the signal from the external controller and suppress the generated torque.
The aim is to improve vehicle acceleration, improve fuel efficiency, and prevent engine stalling.

In its operation, a duty signal is input to the output current control circuit through the external controller C terminal, and the operation and stop of the PWM control circuit are controlled by the output signal from the control circuit.

If the duty of the load signal input to the C terminal is changed linearly according to the amount of external load (vehicle load) as shown in FIG. 35, the output current of the generator - Voltage characteristics can be controlled.

FIG. 36 shows representative examples of a case where the duty is 100% and a case where the duty is 50%.

180. In this embodiment, a load response control function is provided to reduce fluctuations in engine speed due to sudden changes in electrical load and vibrations caused thereby.

The operating principle is shown in FIGS. 24(a) and 24(b).

When a load is applied without normal load response control, the control voltage (battery terminal voltage) drops, but the control system's feedback operation causes the current command value to respond in a stepwise manner to rapidly charge the battery.

At this time, since the generator becomes a load on the engine, the engine speed decreases (FIG. 24(a)).

This is particularly a problem near idle operation where the engine speed is low, and if the engine speed fluctuates rapidly before idle correction, there is a risk of engine stalling.

On the other hand, in load response control, the generator is controlled so that it is unlikely to become a load on the engine before the idle correction. Even if the control voltage drops due to load application, if the current command value is controlled to increase slowly in a fixed pattern, the recovery of the control voltage will be delayed, but the amount of fluctuation in engine speed can be reduced (Figure 24). (b)). For this reason, a delay circuit with a constant time constant is provided in the control loop to change the current command value output for voltage control in accordance with the rotation speed. The pattern of the current command value is shown in FIG. 25, which shows the change in the command value pattern depending on the presence or absence of load response control. The current command value that changes stepwise in the case of no control is switched to the reference value itself when the command value exceeds the reference value v1 in the case of control, and is fixed for a certain period of time. After that, if it is determined whether the fire reference value v2 is exceeded and the command value is sequentially switched to the next reference value, the current command value will rise slowly and stepwise. After finally fixing to the last reference value, it becomes the same value as without control.

If the current command value drops, no switching operation with the reference value is performed and the value remains the same as without control.

Therefore, no matter what current command value is before the load is applied, the command value is fixed only when it exceeds the reference value, so overcharging and overdischarging can be prevented.

Load response control shall be performed near the idle speed,
It is now possible to operate at an alternator rotation speed of 250 Or/win or less. FIG. 16 shows a circuit block for generating an actual command value pattern. An analog switch is used to change the command value, a comparator is used to compare the reference voltage and the command value, and a timer is used to control control operations.

It is constructed using a digital logic circuit including a latch.

Considering the integration of IC, the number of comparison stages with the reference value is set to 3 to prevent the circuit scale from increasing.

A simulation was conducted to verify the effectiveness of the load response control described above. FIG. 17 shows a model of the external torque-engine speed characteristic using the bypass air amount by idle control as a parameter. FIG. 18 shows the step response of the idle rotation speed when an electric load (equivalent to 20 A) is applied using this model. It was confirmed that by performing load response control, the amount of decrease in rotational speed could be reduced from 100 r/ll1in to 25 r/Ein or less.

Although the number of comparison stages is three in this embodiment, the number of comparison stages is not limited to this, and may be set without stages.

Next, a control mode when controlling by a microcomputer installed in a vehicle will be explained below.

The principle will be explained using the functional block diagram shown in FIG.

The deviation between the set value VBC and the actual value S of battery electricity is amplified by a voltage deviation amplifier and output to the limiter.

The limiter outputs a current command value Izo according to the input from the voltage deviation amplifier. In determining the current command value Izo, the magnitude of the load current supplied to the electrical load and the load information or environmental information for the vehicle engine are input into the microcomputer, and the maximum value of the optimal current command value at that time is determined. The current command value ■io is determined within the range according to the voltage deviation and output.

Next, the deviation between the current command value Izo and the actual current value If is detected, the deviation is amplified by an amplifier, and the pulse width modulation circuit (
PWM) drive signal is output.

PWM drives the chopper of the field winding drive circuit with a duty that corresponds to the drive signal, and the field winding current works! control.

This allows the output generated in the armature winding of the generator to properly charge the battery.

Next, the relationship with engine control will be explained with reference to the block circuit diagram shown in FIG. 38 and the control flow chart shown in FIG. 39.

The microcomputer, whose registers have been initialized in step 200, passes step 20 through the A-D converter.
1, input signals such as engine speed, manifold intake pressure, knock signal, throttle opening signal, and battery load current are detected and input to the random access memory RAM.

In addition, the load current depends on the load application state, whether the switch is ON or OFF.
It is also possible to detect it using F and take it in via an input register.

In step 202, an ignition system control signal, a fuel system control signal, and an exhaust system control signal are calculated and output based on the input signal according to the calculation flow stored in the only memory ROM.

The next step 203 is to detect the magnitude of the engine load using the intake pressure. If the intake pressure is determined to be lower than a predetermined pressure Pa (negative pressure), the field current is set so that the generator does not reach the load torque of the engine. The maximum value of the command value is set to 0 so that it becomes zero.

If it is determined that the intake pressure is higher than the predetermined value P&, it is determined that the engine is operating under normal load and the process proceeds to the next step.

In step 204, it is detected whether the throttle opening is fully open or not, and when it is determined that the throttle opening is fully open, it is determined that the acceleration state is in progress.
At this time as well, the maximum value of the current command value ■i.

is set to O to prevent the generator from becoming a load on the engine.

If the throttle is not fully opened, it is determined that the vehicle is running normally and the process proceeds to the next step.

In step 205, it is determined from the knock signal whether or not there is a heavy knock state, and if it is determined that the knock state is a heavy knock state, the maximum value Izo of the current command value is set to O so that the generator does not become a load on the engine.

If you are not in a heavy knock state, proceed to the next Stella play.

In step 206, it is determined from the knock signal whether or not there is a light knock state, and if it is determined that the knock signal is a light knock state, the maximum value Iio of the current command value is set to 2A, and the power generation capacity is suppressed to a low level. reduce the load torque.

If it is not a light knock, it is determined that there is no knock and proceeds to the next step.

In step 207, the engine rotation speed is 150°r, p.
, m or less, and if it is determined to be less than that, calculate the amount of variation in the electrical load based on the load current or the number of ON times of the load switch, etc., and set the optimal current command value based on it.

Calculate and output aX.

If the rotation speed is 150 Or, p, m or more, the maximum value IffimaX of the current command value is set to the maximum allowable current value of 4.5 A, and control is performed so that the maximum output is obtained.

The maximum value Ifmax of the current command value thus determined is inputted to the limiter of the generator control circuit shown in FIG. 27 via the DA converter.

It is also possible to output the maximum value DIzmax of the field current command value as a duty signal from the output register of the microcomputer. In this case, control can be achieved by inputting the PWM output eo and DIzmax of the generator control circuit to the field winding drive circuit via an AND gate.

According to this embodiment described above, 1. The field current is cut and controlled according to the intake pressure of the engine, so when a sudden load is applied to the engine, such as when climbing a hill, the generator is turned off. Since the load on the engine can be prevented, engine stalling can be prevented.

In addition, since the power generation cut control is performed even when the slot is fully open, the engine output can be used sufficiently for acceleration during acceleration, and acceleration performance can be improved.

In addition, the power generation capacity of the generator is controlled according to the knock state of the engine, so when the ignition timing is delayed and the engine output is reduced, such as when a knock occurs, the driving torque for the generator is controlled. This can prevent problems such as engine stalling due to reduced output and redundant knocking conditions.

Furthermore, when the engine speed is low, optimum field current control can be performed depending on the load current, that is, the state of the electrical load, so that engine stalling due to a drop in the engine speed at low engine speeds can be prevented.

According to this embodiment, by controlling the field current of the automobile charging generator, it is possible to prevent fluctuations in the field current due to the difference in temperature and cold of the field winding resistance. Therefore, conventionally, the alternator (charging generator) was designed with a margin to account for the current fluctuation due to the difference in temperature and temperature, but since there is no need to take into account the fluctuation, the output power can be increased if the alternator has the same size. Alternatively, if the output is the same, the size can be reduced. It is also possible to reduce the capacity of the semiconductor element of the field current control chopper. Further, when the load suddenly changes, by controlling the rising operation of the field current using an external signal, it is possible to prevent sudden changes in the load on the automobile engine. That is, it is possible to arbitrarily and continuously vary the field current value from the minimum value to the maximum value in accordance with an external signal. Therefore, in response to an external request, for example, reduction or stop of the alternator's power generation from the engine control can be easily realized.

Furthermore, when the alternator rotates at low speed and the amount of power generated is small, the field current is reduced to the minimum necessary.

It is also possible to reduce the amount of discharge of the battery and to suppress field loss by setting it in a so-called initial excitation state.

Furthermore, if the current detection method of this embodiment is used, the intermittent current flowing through the chopper element can be detected without directly detecting the field current.
Since continuous field current can be detected equivalently, an expensive insulated current detector or the like is not required. Further, there are effects such as being able to continuously detect the field current from the minimum value to the maximum value.

Further, according to this embodiment, the command value of the current to be supplied to the field winding is determined from the signal corresponding to the deviation voltage signal and the signal corresponding to the actual current flowing through the field winding, and this command value is obtained. Since the current is supplied to the field winding based on the current, it is possible to control the output of the generator to the optimum output over a wide range according to the demands of the load, while preventing internal fluctuations in the field current. is possible,
We were able to achieve optimal control for output control of generators with large load fluctuations.

Further, in the invention related to the detection of field current, since the need to use a current transformer is eliminated, it is possible to obtain a generator control device and method that are low in cost and suitable for IC implementation.

Furthermore, in the invention of load response control, by organically combining it with current feedback control, it was possible to obtain a control device and a control method in which the torque fluctuation of the generator is small and the rotation of the prime mover is not adversely affected.

〔Effect of the invention〕

According to the present invention, the load current can be detected without using any external components other than the control circuit, which facilitates implementation. Furthermore, integration reduces the number of parts, increasing the reliability of the control circuit and reducing costs. Further, in integration, the circuit area can be reduced, so the cost of the IC chip can be reduced. Furthermore, loss caused by the detection resistor of the semiconductor power switch can be reduced. Furthermore, since the detected current can be obtained continuously with a small error, control accuracy is improved and control is stabilized. In addition, in the vehicle generator control device, the output can be increased by current limiting with the same main body as the conventional one. Furthermore, the control circuit can be easily adjusted even when the loads are different.

[Brief explanation of the drawing]

FIG. 1 is a circuit diagram used in the first embodiment of the present invention, and FIG. 2 is an explanatory diagram of its operating waveforms. Figure 3 shows
This is an example of a circuit converted into an IC in the second embodiment of the present invention,
FIG. 4 is an explanatory diagram of its operating waveforms, and FIG. 5 is its detailed circuit diagram. FIG. 6 is a block diagram of a circuit used in a third embodiment of the present invention. FIG. 7 is a circuit diagram used in the fourth embodiment of the present invention, and FIG. 8 is an explanatory diagram of its operating waveforms. FIG. 9 is a circuit diagram of a vehicle generator control device used in a fifth embodiment of the present invention. Fig. 10 is a layout diagram on an IC chip of the present invention, Fig. 11 is a block diagram of main parts showing the circuit configuration of a control device of a charging generator for a car according to an embodiment of the present invention, and Fig. 12 is a diagram of the main part of the control device circuit configuration of a charging generator for a car according to an embodiment of the invention. 13 to 15 are operation diagrams showing an example of the control operation, and FIG. 16 is a detailed circuit diagram of the embodiment shown in FIG. 11. FIG. 17 is a detailed diagram of the current detection circuit of this embodiment,
Figures 18 to 21 are operation and characteristic diagrams of each part of this embodiment, Figures 22 and 23 are explanatory diagrams explaining the effects of this embodiment, and Figures 24 (a) and (b) are A principle diagram for explaining the operating principle of the load response control circuit of the embodiment, Fig. 25 is an explanatory diagram of the control operation, Fig. 26 is a specific circuit diagram of the circuit, and Fig. 27 shows the bypass air amount as a parameter. Figure 28 is a diagram to explain the effect of the load response control circuit, and Figures 29 and 30 are diagrams showing the relationship between alternator drive torque and engine speed when A drawing showing the relationship with the lighting state of S.
A diagram showing the S-B terminal switching state and the charge lamp blinking state with respect to the terminal voltage, FIG. 32 is a diagram showing the terminal switching state, gate lock state, and charge lamp blinking state when the voltage is lower than the B terminal voltage, and FIG. 33 shows the gate A drawing showing the blinking state of the charge lamp with respect to voltage, FIG. 34 is a drawing showing the lighting and blinking state of the charge lamp in each abnormal state,
Fig. 35 is a drawing showing the signal input to the C input terminal as an external signal, Fig. 36 is a drawing showing the capacity control state of the generator, and Fig. 37 is control of the vehicle generator using a microcomputer. The device is shown ■Function''.Tsuku diagram, Figure 38 is the open control circuit''' Warming diagram, Figure 39
The figure is a control flowchart. 1... Semiconductor power switch, 2... Flywheel diode, 3... Inductive load, 4... Battery, 5
... Drive circuit, 6e... Current detection resistor, 8... Current detection circuit, 9... Analog switch, 1o... Capacitor, 11... Buffer amplifier, 12... Discharge resistor, 13 ...sample/hold circuit, 14..oscillator, 17..discharge circuit, 18..power IC180
...IC regulation 1Q figure 13 figure 14 figure 15 figure 17 figure V. Fig. 18 Fig. 19 Vzf Fig. 20 Fig. 21 Electrical Fig. 22 Fig. 23 Generator rotation speed 24 (a) Load application time t (b) Load application time t Time 26 Figure Time Figure 27 Figure 28 Time (SeC) Figure 29 (N+) (N2) Figure 30 (NI) Figure 31 Figure 32 Figure 33 Figure 34 Figure 36 (1) 100% 1□ Figure 36

Claims (1)

[Scope of Claims] 1. A current control circuit for a semiconductor power switch, comprising means for storing the conduction current of the semiconductor power switch, and means for correcting the memory according to changes in load current. 2. The device according to claim 1, further comprising a semiconductor power switch that controls the conduction current of the inductive load, a drive circuit thereof, a current detection circuit that detects the conduction current of the semiconductor power switch as a voltage, and a current detection voltage value that is stored. 1. A current control circuit for a semiconductor power switch, comprising a sample-and-hold circuit and a means for correcting a sample-and-hold voltage according to a time constant of an inductive load when the semiconductor power switch is non-conducting. 3. The device according to claim 2, wherein the sample and hold circuit includes a capacitor for holding voltage, an analog switch for switching the input voltage, and an FET input amplifier for outputting the charging voltage of the capacitor. 1. A current control circuit for a semiconductor power switch, characterized in that a discharge circuit in which a resistor is connected between a high potential side and a low potential side of a capacitor of the sample and hold circuit is used as a correction means. 4. The device according to claim 2, wherein the sample-and-hold circuit includes a capacitor for holding voltage, an analog switch for switching the input voltage, and an FET input amplifier for outputting the charging voltage of the capacitor. 1. A current control circuit for a semiconductor power switch, characterized in that the sample-and-hold voltage is digitized by an A/D converter, and the digital amount is corrected by an arithmetic circuit. 5. The semiconductor power switch according to claim 3, wherein the semiconductor power switch, the drive circuit, the current detection circuit, the sample and hold circuit, and the correction means are integrated on the same substrate. Switch current control circuit. 6. The device according to claim 3, further comprising a discharging resistor for discharging the charging voltage of the capacitor of the sample and hold circuit into the discharging circuit, an electronic switch for limiting the discharging time of the discharging resistor, and driving the electronic switch. 1. A current control circuit for a semiconductor power switch, characterized in that an internal oscillator is used to control the current, and a discharge resistor and an electronic switch are connected in series between a high potential side and a low potential side of a capacitor. 7. The device according to claim 3, wherein the discharge circuit includes a constant current source for discharging the charging voltage of the capacitor of the sample and hold circuit, an electronic switch for limiting the discharge time of the constant current source, and an electronic switch. 1. A current control circuit for a semiconductor power switch, characterized in that an internal oscillator is used to drive the circuit, and a constant current source and an electronic switch are connected in series between a high potential side and a low potential side of a capacitor. 8. A current control circuit for a semiconductor power switch according to claims 6 and 7, characterized in that an internal oscillator for driving the electronic switch is provided with frequency and duty ratio variable means. 9. A current control circuit for a semiconductor power switch according to claims 6 and 7, wherein the low potential side of the capacitor is connected to a virtual ground potential higher than the ground voltage of the circuit. 10. The current control circuit for a semiconductor power switch according to claim 7, wherein an active load of a transistor is used as a constant current source of the discharge circuit, and discharge is performed with a minute current. 11. A current control circuit for a semiconductor power switch according to claim 2, characterized in that it is used for current control of a generator control device for a vehicle. 12. The device according to claim 5, wherein the capacitor of the sample and hold circuit is arranged in parallel with the semiconductor power switch at a constant distance, and the discharge circuit of the correcting means is adjacent to the capacitor and on the opposite side of the semiconductor power switch. A current control circuit for a semiconductor power switch characterized by being arranged and integrated. 13. In order to maintain the voltage of the battery power source at a predetermined voltage setting value, a field current flows through a field winding of a generator that charges the battery according to a voltage deviation between the battery voltage and a predetermined voltage setting value. field current signal generating means, comprising a field current control means for controlling a current, the field current control means generating a signal according to the current flowing through the field winding; and the field current signal generation means. a current command value that provides a field winding current command value such that the voltage of the battery has a magnitude necessary to maintain the voltage of the battery at the predetermined voltage setting value based on a signal from the means and a signal corresponding to the deviation current; the field winding current supply means for supplying a predetermined current to the field winding based on the current command value from the field winding current command value generating means; The line current supply means is turned on in response to the command value from the field winding current command value generation means.
pulse signal generating means for generating a pulse signal with varying OFF duty; semiconductor switching means for periodically turning on and off in accordance with the pulse signal from the pulse signal generating means; and when the switching means is in the ON state, It is composed of a DC power supply connected to the field winding, and
The current signal generating means of the field winding current command value generating means includes a storage means for storing a field winding current value immediately before the semiconductor switching means is turned off, and the field winding current command value generating means the semiconductor switching means is OFF
During the period of the state, the current value stored in the storage means is regarded as the actual field winding current value, and a signal corresponding to the current value stored in the storage means and a signal corresponding to the deviation voltage are generated. A generator comprising a current command value correction means for generating the current command value based on the signal and correcting the current command value according to a change in the load current flowing through the field winding. control device. 14. In a device that controls the output of the generator by chopper-controlling the current flowing through the field winding of the generator, a target current of the field winding is determined according to the operating state of the generator, while the detecting the actual current flowing in the field winding, determining the conduction rate of the chopper based on this actual current and the target current, and immediately before the chopper is turned off during the off period of the chopper; A method for controlling a generator, characterized in that the conductivity of the chopper is determined based on a stored current value, and the conductivity is corrected according to a change in the current flowing through the chopper. 15. The method of controlling a generator according to claim 14, wherein the target current of the field winding is determined according to at least one of a load condition and a rotational speed of the generator. 16. Control of the generator according to claim 15, wherein one of the loads of the generator is a battery, and the field current is controlled so as to maintain the voltage of the battery at a predetermined value. Method. 17. Chopper means for chopper controlling the current flowing through the field winding of the generator; current detection means for detecting and storing the current flowing through the field winding when the chopper means is turned on; A control device for a generator, comprising: memory current value correction means for correcting according to attenuation characteristics of a field winding current at a given time. 18. The device according to claim 17, wherein the current detection means is constituted by a voltage holding means that holds a voltage according to the detected current, and the stored current value correction means is held by the voltage holding means. A control device for a generator, comprising a voltage attenuation means that attenuates a voltage according to an attenuation characteristic of the field winding current. 19. The generator according to claim 18, wherein the voltage holding means comprises a sample and hold capacitor, and the voltage attenuation means comprises a discharge circuit of the capacitor including a resistor connected to the sample and hold capacitor. Control device. 20. In claim 19, the discharge circuit comprises a constant current drawing circuit, a switching means for intermittent drawing current, and a duty for switching the switching means at a duty corresponding to the attenuation characteristic of the field winding current. A generator control device consisting of a signal source. 21. The generator control device according to claim 20, wherein the duty signal source is an oscillator.