JPH0347021B2 - - Google Patents

Description

[Detailed Description of the Invention] [Field of Industrial Application] The present invention is directed to detecting the amount of co-frequency interference in a wireless communication method, and particularly relates to a method capable of detecting the amount of interference with high accuracy. It is something.

[Conventional technology]

In wireless communications, in order to use frequencies efficiently, a small number of channels are commonly assigned to multiple devices for use. Therefore, if a receiving device causes interference on the same channel due to radio waves from another transmitting device, avoid using that channel, and if the channel in use is subject to interference from another device, avoid using that channel. It is necessary to automatically detect same-channel interference and reset to another channel for use.

As a method for detecting the amount of co-frequency interference used for this purpose, a method of detecting an envelope beat that occurs when a desired wave and an interfering wave are simultaneously received is conventionally known.

FIG. 1 is a block diagram showing an example of the configuration of a conventional co-frequency interference detection circuit using beat detection (Japanese Patent Application No. 58-18102 (Japanese Patent Publication No. 1-20817), 58-
No. 68428 (see Special Publication No. 1-19779)).

In the figure, 1 is an antenna, 2 is a receiver, 3 is a detector, 4 and 5 are analog-to-digital converters (A-D converters), 6 is an arithmetic unit, 7 is a delay circuit, and 8
is the interference amount output terminal.

In the figure, a composite wave e of a desired wave and an interfering wave is received by a receiver 2 through an antenna 1 and input to a detector 3.

Now, assuming that the desired wave (hereinafter referred to as D wave) and the interfering wave (hereinafter referred to as U wave) are FM modulated and expressed by equations (1) and (2), the composite wave e is expressed by equation (3).

The D wave is e ₁ = E ₁ sin (ω ₁ t + Δω ₁ /P ₁ sinP ₁ t) ...(1) The U wave is e ₂ = E ₂ sin (ω ₂ t + Δω ₂ /P ₂ sin (P ₂ t + θ) + φ ) ...(2) Composite wave e = e ₁ + e ₂ = (E ₁ ² + E ₂ ² + 2E ₁ E ₂ cosψ) ^1/2 Xsin (ω ₁ t + Δω ₁ /P ₁ sinP ₁ t + tan ^-1 Λsinψ (t) /1+Λcosψ(t))…
...(3) However, Λ=E ₂ /E ₁ , E ₁ , and E ₂ are the amplitudes of the D and U waves and vary depending on the propagation conditions, and since the propagation paths of the D and U waves are generally different, their fluctuations are mutually exclusive. It is independent.

ω ₁ and ω ₂ are the carrier wave angular frequencies of the D wave and the U wave, and φ is the phase difference. P ₁ and P ₂ are the angular frequencies of the D-wave and U-wave modulation signals. Δω ₁ and Δω ₂ are the angular frequency deviations of the D wave and the U wave.

Also, φ(t) is the instantaneous beat phase, ψ(t)=(ω ₂ −ω ₁ )t+φ+Δω ₂ /P ₂ sin(P ₂ t+θ)−Δω ₁ /P ₁ sinP ₁ t……(4) becomes.

When the composite wave e is square-law detected by the detector 3, its envelope R(t) becomes R(t)=E ₁ ² +E ₂ ² +2E ₁ E ₂ cosψ(t) (5).

E ₁ and E ₂ are the amplitudes of the D wave and the U wave, which are generally subjected to fading and fluctuate around their average value. FIG. 2 is a diagram showing the envelope of the composite wave, and an example of the waveform of R(t) in this case is shown in (1) (detector output (envelope)).

R(t) has a waveform in which a high frequency component is superimposed on a low frequency component E ₁ ² +E ₂ ² (broken line 2 in the figure) which slowly fluctuates due to fading.

As can be seen from equation (4), the high frequency component (2E ₁ E ₂ cosψ(t)) is a high frequency component with a beat frequency determined by (ω ₂ - ω ₁ )/2π, modulation degree, Doppler, etc.
It changes at a frequency that is orders of magnitude higher than the low frequency component.

R(t) having such two frequency components is sampled by A-D converters 4 and 5, and then
By processing, the low frequency component and the high frequency component are separated and their average values X and Y are determined.

First, the A-D converter 4 converts R(t) into period T
The average value X of the low frequency components is determined by sampling each time and taking the collective average 〈〉.

X=1/N _N 〓 ⁱ⁼¹ (E _1i ² +E _2i ² +2E _1i E _2i cosψ _i =〈E ₁ ² +E ₂ ² 〉=〈E ₁ ² 〉+〈E ₂ ² 〉 ……(6) This At the same time, the A-D converter 5 generates a sample value R delayed by a time Δt (Δt≪T) with respect to R(t).
(t+Δt) and remove the low frequency component by averaging [R(t)-R(t+Δt)] ² .
The root mean square Y of the high frequency components is determined.

Y=〈[R(t)−R(t+Δt)] ² 〉 =4〈E ₁ ² E ₂ ² (cosψ(t) −cosψ(t+Δt)) ² 〉 ……(7) In the modification of the above formula, E _i ² (t)≒E _i ² (t+
The relationship Δt) (i=1, 2) is used. ψ(t)
When is not a periodic signal and can be regarded as random for N samples of NT seconds, equation (7) becomes as follows.

Y=4〈E ₁ ² 〉〈E ₂ ² 〉...(8) Now, let Γ be the square ratio of the amplitudes of D wave and U wave (D/U ratio), Γ=〈E ₁ ² 〉/〈E ₂ ² 〉 ...(9) Then, from equations (6), (8), and (9), the D/U ratio Γ is It can be seen that the amount of interference (D/U ratio) can be quantitatively detected from X and Y.

The above calculations are performed by the computer 6 and Γ is output to the output terminal 8.

[Problem that the invention seeks to solve]

As is clear from the above description, in order to accurately measure the amount of interference, the average value X of the low frequency components and the root mean square value Y of the high frequency components must be accurate. It is generally known that X can be determined with high accuracy by averaging a sufficient number of sample values.

On the other hand, there are various factors that can cause errors in Y, but one of the factors that poses a problem is due to the carrier frequency difference (ω ₂ −ω ₁ )/2π between the D wave and the U wave. As is clear from equation (7), the value of Y depends on the instantaneous phase ψ(t) expressed by equation (4).

In particular, when the angular frequency deviations Δω ₁ and Δω ₂ of the angular modulation of the D and U waves are small, equation (4) becomes ψ(t)≒(ω ₂ −ω ₁ )t+φ ……(11) , (ω ₂ −ω ₁ ) becomes smaller, the correlation between ψ(t) and ψ(t+Δt) becomes extremely strong. Therefore, cosψ(t)≒cosψ(t+Δt), and from equation (7), Y=4〈E ₁ ² E ₂ ² (cosψ(t) −cosψ(t+Δt)) ² 〉≒0 ...(12) Therefore, there was a problem that interference detection became impossible.

Also, even when (ω ₂ −ω ₁ )≒0 does not hold, Y
Since the value of depends on (ω ₂ −ω ₁ ), there was a problem in that a large error occurred in detecting the amount of interference.

In view of the above-mentioned conventional problems, the present invention provides D-wave and U-wave
It is an object of the present invention to provide a highly accurate method for detecting the amount of same-frequency interference, which does not make interference detection impossible or cause large detection errors due to differences in carrier wave frequencies.

[Means for solving problems]

According to the present invention, the above object is achieved by the means described in the claims. In the present invention, the maximum frequency deviation f _d of the desired wave and the interference wave is 0.85×n/4Δt≦f _d ≦1.2×n/4Δt (n is any natural number) (13) The most important feature is that angle modulation is always applied, and with conventional technology,
The maximum frequency deviation of the angular modulation of the desired wave and the interference wave is not limited at all, and specific examples thereof (for example, literature: Kozono, Sakamoto "Measurement of co-channel interference in mobile communications" Institute of Electronics and Communication Engineers) Journal (B), January 1985 issue p109-116), the relationship between Δt and f _d does not satisfy equation (13).

In the present invention, it has been found that when the relationship of equation (13) is established between f _d and Δt, a detection error due to the difference in carrier frequency between D wave and U wave does not occur, and this phenomenon is utilized. Therefore, the present invention differs from the conventional technology in that the D wave and the U wave are always subjected to angular modulation that satisfies equation (13) to suppress detection errors.

〔Example〕

FIG. 3 is a block diagram of one embodiment of the present invention, in which 1 to 8 are the same as those in FIG. 1, 9 is a radio device that transmits D waves, and 9-1 is a radio device that transmits D waves. Audio band signal generator for always applying FM modulation with maximum frequency deviation f _d , 9-2 is an adder, 9-3 is FM
Modulator 9-4 represents a transmitting antenna.
Further, 10 is a microphone for the radio device 9, 11 is a radio device that transmits U waves, and 11-1 is a signal generator outside the audio band for constantly applying FM modulation to the U waves with a maximum frequency deviation f _d . 11-2 is an adder, 11-3
is an FM modulator, 11-4 is a transmitting antenna, and 12 is a microphone for the radio device 11.

The configuration of the receiving side in this embodiment is exactly the same as that in FIG. 1, and its interference detection operation is also exactly the same.

On the other hand, on the transmitting side, out-of-band signal generators 9-1 and 11-1 are provided for D waves and U waves, respectively. The D wave is constantly FM modulated using the information from 9-1 so that the maximum frequency deviation f _d satisfies the condition of equation (13). Also, for U waves, 11-
FM modulation with the maximum frequency deviation f _d (the same amount of deviation as the D wave) is always performed using the information obtained from 1 and independent of the D wave. The information output from the out-of-band signal generator may be any signal that does not interfere with the audio band. For example, a tone signal may be inserted using the lower audio band of 300 Hz or less.

By band-limiting random digital signals,
It may also be one that can transmit in a band of 300Hz or less. However, when using a tone signal, it is better to make the tone signal frequencies different between the D wave and the U wave.

The reason for this is that if the tone signal frequencies of the D wave and the U wave match, the root mean square value Y of the high frequency component changes due to the phase difference between the tone signals, resulting in a detection error.

The effect of performing constant FM modulation on the transmitting side in this way will be explained below.

FIG. 4 is a diagram showing the relationship between the instantaneous frequency locus of a modulated wave and the modulation spectrum.

Now, according to the information from the out-of-band signal generators 9-1 and 11-1, the D wave and U wave are respectively FM.
Modulated, the instantaneous frequency trajectory g _D as shown in a and b in the figure
(t) and g _U (t). The out-of-band signal is a random signal, and the frequency component of the out-of-band signal is sufficiently small compared to f _d , so the modulation spectra of the D wave and U wave are based on the carrier frequencies f 1 and f ₁ as shown in d and e, respectively. It has a flat shape with f ₂ as the center and a flat spread of ±f _d above and below.

When receiving such D waves and U waves in combination, the formula
The instantaneous frequency (beat frequency) of the high-frequency component of the envelope R(t) shown in (5) {ψ〓(t)/2π} is centered around |f ₂ −f ₁ |, as shown in c in the figure. and varies randomly within a range of ±2f _d .

Therefore, the power spectrum B(f) of the high frequency component {2E ₁ E ₂ cosψ(t)} is as shown in f in the figure, |
With f ₂ - f ₁ | as the center, the upper side is up to +2f _d , and the lower side is a shape in which components below zero frequency are folded back. In this way, the power spectrum of high frequency components is divided into D wave and U wave.
The shape changes depending on the carrier frequency of the wave, but
As is clear from equation (5), the total power P _B is: P _B =〈{2E ₁ E ₂ cosψ(t)} ² 〉 = 2〈E ₁ ² 〉〈E ₂ ² 〉 ... (14) and is constant regardless of (f ₂ − f ₁ ).

Next, consider the output (differential sampling output) when a high frequency component having such a power spectrum B(f) is sampled at a sampling interval Δt and the difference is taken.

The differential sampling operation can be considered using a model as shown in FIG. That is, in FIG. 5, 13 represents sampling, 14 represents delay (Δt), and if a cosine wave is considered as input signal b(t), output b(t) is b(t)=cos(2πf _b t) ... (15) (t) = cos (2πf _b t) - cos {2πf _b (t + Δt)} = 2 {sinπf _b Δt) ・cos (2πf _b t + Q) ... (16) However, Q = tan ^{- 1} (2tanπf _b Δt)+π/2...(17) Therefore, the power transfer function |H(f)| ² of the system shown in FIG. 5 becomes |H(f)| ² =4|sin(πfΔt)| ² (18).

The characteristics of the power transfer function (|H(f)| ² ) in FIG. 5 are shown in FIG.
When ^inputting _a ^high frequency ^component _{with )} ｜H(f)｜ ²・σdf……
(19) (f2−f1−2fd) σ=2〈E ₁ ² 〉〈E ₂ ² 〉/4fd...(20) However, B(f) is the band [f ₂ −f ₁ −2f _d , f ₂
−f ₁ +2f _d ] and have a uniform spectral density σ.

An example of calculating the relationship between |f ₂ −f ₁ | and Y by substituting equation (18) into the above equation and using f _d as a parameter is a diagram showing the relationship between carrier frequency difference and high frequency component power detection value. As shown in Fig. 7.

In the figure, a is f _d = 0, b is f _d = 1/
For 4Δt, c is for f _d =1.5/4Δt.

In the case of b, Y is constant regardless of the carrier frequency difference |f ₂ -f ₁ |, and its value Y ₀ is Y ₀ = 2〈E ₁ ² 〉〈E ₂ ² 〉=P _B ... (21) , equal to the input power.

As can be seen from FIG. 7, the maximum error in Y depends on f _d . Therefore, the difference between the maximum value Y _nax and the minimum value Y _nio of Y is defined as E, and the relationship between the interference detection deviation E and the frequency deviation f _d is shown in FIG.

From this figure, it can be seen that E becomes zero when f _d =n/4ΔT (n is any natural number) (22).

Furthermore, in order to suppress E to 1.5 dB or less, f _d should satisfy the condition of equation (13).

In this way, if f _d is set to satisfy equation (22) or condition (13), the interference amount detection deviation due to a change in carrier frequency difference can be suppressed to zero or to a very small amount.

Therefore, it is possible to detect interference with higher accuracy than with conventional techniques.

〔Effect of the invention〕

As explained above, according to the present invention, D waves and U
This makes it possible to measure the amount of co-frequency interference with high accuracy, which is not affected by changes in the carrier frequency difference between waves.For example, when the situation of co-frequency interference changes from moment to moment, such as in mobile communications, the difference in carrier frequency between D and U waves can be measured. This is effective when the value is not constant.

Then, if the interference detection method according to the present invention is used,
Since the amount of interference can be quantitatively detected during a call, if the amount of interference exceeds a certain set value, it is possible to continue high-quality calls without deteriorating call quality by switching to another channel. It has the advantage of being able to form a system.

Additionally, in systems that aim to improve frequency utilization by setting the radio zone to a small zone of 3 to 5 km, such as the car telephone system, this system can further reduce the frequency repetition distance, thereby improving frequency utilization. This can contribute to an increase in subscriber capacity.

[Brief explanation of drawings]

FIG. 1 is a block diagram showing an example of the configuration of a conventional co-frequency interference amount detection circuit, FIG. 2 is a diagram showing an envelope of a composite wave, FIG. 3 is a block diagram of an embodiment of the present invention, and FIG. The figure shows the relationship between the instantaneous frequency trajectory of the modulated wave and the modulation spectrum, Figure 5 shows the differential sampling model, Figure 6 shows the power transfer characteristics of the system in Figure 5, and Figure 7 8 is a diagram showing the relationship between the carrier wave frequency difference and the high frequency component power detection value, and FIG. 8 is a diagram showing the relationship between the frequency shift amount and the interference detection deviation. 1... Antenna, 2... Receiver, 3... Detector, 4, 5... A-D converter, 6... Arithmetic unit, 7
...Delay circuit, 8...Interference amount output terminal, 9...D
Radio device for transmitting waves, 9-1...Audio band signal generator, 9-2...Adder, 9-3...FM modulator, 9-4...Transmission antenna, 10...Microphone,
11...Radio device that transmits U waves, 11-1...Audio band signal generator, 11-2...Adder, 11
-3...FM modulator, 11-4...Transmission antenna, 12...Microphone, 13...Sampling, 14...
Delay (Δt).

Claims (1)

[Claims] 1. In a wireless communication system in which when a desired wave and an interference wave are simultaneously received, the envelope fluctuates at a frequency higher than the fading frequency by angle modulation or carrier frequency offset. ,
By sampling the envelope square law detection output of the received wave and taking the average value, a low frequency component having a frequency component almost equal to the fading frequency is obtained, and the envelope square law detection output is calculated at time t and t+Δt
Sampling is performed with a delay time Δt such that the values sampled at are considered to be the same value for fading, and the product of the two sampling values is considered to be zero for high frequency components that fluctuate at a frequency higher than the fading frequency, A high frequency component is obtained by taking the square mean of the difference between the sampling values of t and t+Δt, and the amount of interference of the same frequency is detected by processing these low frequency components and high frequency components to detect the amount of interference. In this method, angular modulation is always applied to the desired wave and the interfering wave so that the maximum frequency deviation f _d is 0.85×n/4Δt≦f _d ≦1.2×n/4Δt (n is any natural number). A co-frequency interference detection method characterized by: