JPH02290185A - Vector control method and device for induction motor - Google Patents

Abstract

PURPOSE:To prevent phase deviation between a voltage command and a synchronous carrier wave and prevent the generation of current beat by a method wherein the differential value of the internal phase difference angle of an induction motor is added to a frequency command value while the frequency of the synchronous carrier wave is controlled by the sum. CONSTITUTION:The speed of an induction motor 1 is controlled variably by vector control effected by an inverter circuit 2 while the primary current I1 of the induction motor 1 is detected by a current transformer CT and is inputted into a torque current component detecting means 3. A frequency command in regard to the determination of the frequency of a PWM carrier wave is corrected based on the differential value of the internal phase difference angle command of the induction motor 1. In this case, the differential value is subtracted and/or added only while the phase difference angle command is being changed and, therefore, the phase of the carrier wave is advanced or delayed in appearance in synchronism with the change of the phase difference angle. According to this method, the phase deviation between a voltage command and the carrier wave may be prevented and the frequency of them are retained, whereby the generation of current beat phenomenon may be prevented.

Description

DETAILED DESCRIPTION OF THE INVENTION (Field of Industrial Application) The present invention relates to a vector control method and apparatus for an induction motor, and specifically to a vector control method and apparatus using a PWM control inverter.

[Conventional technology]

As an example of a variable speed control method for an induction motor, for example, the National Conference of the Institute of Electrical Engineers of Japan, Industrial Applications Division, National Conference Lecture Paper "Simple Speed Sensorless Vector Control Method" (August 1987, &128)
The proposed method is known. This method uses a current detector instead of a conventional speed detector to detect the primary current of the induction motor, and based on this detected primary current, the output frequency, output voltage, and rotating magnetic flux of the inverter are determined. This control enables high-performance variable speed control using vector control. Additionally, in general, PWM control uses an asynchronous method in which the carrier wave frequency fc is constant in the low frequency operation range, and a synchronous method in which the carrier wave frequency fc is changed according to the inverter output frequency fe in the high frequency operation range. ．． The reason for this is that in the high-frequency operation range, the relative ratio fc/f with the carrier wave frequency becomes smaller, so the number of PWM pulses in a half cycle or one cycle decreases, and the one-blow voltage command waveform and carrier wave become periodic. This is because when a phase shift occurs, a current beat phenomenon occurs.

Note that the synchronization method is generally a method in which synchronization is achieved by setting the frequency fc of the carrier wave to an integral number (p) times the frequency command f1 of the inverter. Here, the basic operation of the vector control described above will be explained with reference to the vector diagram shown in FIG. The figure is represented by a ct-q axis orthogonal coordinate system rotating at a synchronous speed ω1. In the figure, the torque current component.
q is the detected instantaneous value of the primary current of the induction motor it(
iu''iw), it is determined by the following equation (1).

? is calculated, and this ωS is added to the speed command ωRS to obtain the primary frequency command ω. of the induction motor. S is determined, and the inverter output frequency is controlled based on this ω1s. On the other hand, the primary voltage vector v1 of the induction motor is given by the vector sum of the induced electromotive force E■ and the motor's internal impedance drop Z-I■, and these v1 and E1 are the internal phase difference angle δ according to Z It becomes a relationship that has the following. So, what is the detected torque current detection value? 1q and excitation current commands I, d
s, the primary frequency command ω, S, and the motor constant (for example,
Based on the impedance element), the primary voltage
and the command values VS and δS of the phase difference angle δ, respectively.

Next, based on the command values vis, δS and the magnetic flux phase reference signal θS obtained from this, the instantaneous voltage command Uus for the three phases is calculated.
=Uws is determined by calculation. For example, Uus is expressed by the following formula (
2).

Uus=V1s sin(e s+δs)
-(2) Here, es=ω, st The voltage command Uus''Uws obtained in this way is compared with the carrier wave ec to generate a pulse width modulation signal, and the output voltage of the inverter is thereby controlled by PWM. Therefore, the inverter output voltage is its command value U us ” U
Since this command value U us =U ws is determined by the command values V1s and δS of the primary voltage and phase difference angle, the induced electromotive force E1 is constant regardless of the primary current 1. As a result, vector control that keeps the rotating magnetic flux φ4 constant at the desired value can be realized. By the way, with the development of microcomputers, the generation of the above-mentioned PWM signal has come to be performed by software, and one known example is the one proposed in Japanese Patent Application Laid-open No. 124773/1983. According to the publication, as shown in FIG. 7, PWM
It generates a signal. In microcomputer processing, the period T for each timing of the peak of the triangular wave is
Start interrupt processing at c, and use the following equation (3). More sine wave U
The intersection with S and S is calculated.

Here, Tc: interrupt period (= 1 / fc = 1
/p-fz)VyuS: Peak value of voltage command T1 and T2 obtained from this are determined as the rise and fall times of the PWM signal pulse. ? Problems to be Solved by the Invention] However, when performing synchronous PWM modulation, the above (
As shown in equation 2), since the phase of the voltage command U is the sum of the magnetic flux phase reference signal θS and the phase difference angle δS, the phase of the carrier wave ec is adjusted in consideration of the phase difference angle δS, and the voltage command It is necessary to synchronize U1 and carrier wave ec.

However, in the method disclosed in the above publication, the interrupt period Tc is determined from the primary frequency commands ω and S, and the phase difference angle δS is not considered.
The phase of G will be shifted. Therefore, completely synchronous PWM control cannot be performed, and if the internal impedance changes depending on the load or speed, the phase difference angle δ changes, so the current beat may increase, and vector control in the high-speed range There is a problem that there is a problem with

An object of the present invention is to provide an induction motor vector capable of performing stable and high-performance altitude control over a wide range from low speed to high speed by correcting the phase shift between the output voltage command and the PWM carrier wave caused by the internal phase difference angle. The purpose is to provide a control device.

[Means to solve the problem]

In order to achieve the above object, the present invention provides an output voltage command of an inverter and a P having a frequency that is an integral multiple of the frequency of the command.
A PWM signal is generated by comparing the WM carrier wave, and the PWM signal is
The method of vector-controlling an induction motor by controlling an inverter using a signal is characterized in that a frequency command related to determining the frequency of the PWM carrier wave is corrected based on a differential value of an internal phase difference angle command of the induction motor.

[Effect]

With this configuration, the frequency of the synchronous carrier wave is increased or decreased in accordance with the phase change of the voltage command that changes according to the internal phase difference angle. That is, since the differential value is increased/decreased and added only while the phase difference angle command is changing, the phase of the carrier wave appears to advance or lag in accordance with the change in the phase difference angle. As a result, the phase shift between the voltage command and the carrier wave is prevented and their period is maintained, resulting in a current beat phenomenon? Occurrence can be prevented.

〔Example〕

The present invention will be explained below based on examples. FIG. 1 shows an overall configuration diagram of an induction motor vector control device according to an embodiment of the present invention.

First, the basic configuration will be explained.

The induction motor 1 is controlled at variable speed by vector control using an inverter circuit 2. The primary current I8 of the induction motor 1 is detected by a current transformer CT, and is input to the torque current component detection means 3. Torque current component detection means 3 determines torque current component Lq according to the procedure explained in FIG. 6, and outputs it to frequency calculation means 4 and voltage command calculation means 5. The frequency calculation means 4 calculates the torque current component I
1q and speed command ωRS are taken in to obtain the primary frequency command ω1s of the inverter. The voltage command calculation means 5 takes in the given excitation current commands I, ds, the torque current component Lq and the primary frequency command ω■S, and based on these, calculates the primary voltage command VYS and the phase difference angle command δ.
It is designed to find S. The three-phase voltage command generation means 6 takes in the primary voltage command V1s and the phase data (θS+δS), and generates the three-phase voltage command US (Uus”Uws) as exemplified in the above equation (2). and outputs it to the PWM modulation means 7.The PWM modulation means 7 includes an asynchronous carrier generation means 9 which generates a triangular wave carrier signal of a constant frequency and an asynchronous carrier wave generation means 9 which generates a triangular wave carrier signal of a variable frequency via a switching means 8. A carrier wave signal ec is inputted from the synchronous carrier wave generating means 10.

Note that the primary frequency command ω, S obtained by the frequency calculation means 4 is input to the integrator 12 via the adder 11,
Here, it is converted into a magnetic flux phase command IJs. This magnetic flux phase command θS is input as a phase reference signal to the torque current component detection means 3 via an adder 13. Further, the magnetic flux phase command OS is determined by the phase difference angle command δS given from the voltage command calculating means 5 via the adder 14.
5 and then input to the three-phase voltage command generating means 6.

Next, the features of this embodiment will be explained.

The mode discrimination means 16 takes in the speed command ωRB, and synchronizes and
The switching signal DF is compared with a predetermined speed ωL set as the asynchronous mode switching speed, and a switching signal DF is output to the switching means 8 and 17. This switching signal DF is ωRS≦ω
When it is L, it becomes “H” level, and when ωRS>ωL, it becomes 11L.
+j level is set.

The switching means 8 and 17 each have a switching signal DF of ゛' I{
The A side of the switching means 17 is grounded, and the B side of the switching means 17 is set to be switched to the A side when the level is "L", and to the B side when the level is "L".
S, and the common end C is connected to the input of delay element 18. The delay element 18 is composed of a first-order delay chestnut or the like, and outputs a phase difference angle command δ! that is obtained by integrating changes in the input phase difference angle command δS. Outputs S. This phase difference angle command δIs is input to the input ends of the adders 13 and 14, and is also input to the differentiator 19. Is this differentiator 19 an input phase difference? The command δ[S is differentiated (dδ+s/dt) L, and a correction signal ωδ corresponding to a change in the phase difference angle command δS is output to the input terminal of the adder 11. The output of this adder 11 is input to the integrator 12 mentioned above, and is also input to the synchronous carrier generation means 10 through the coefficient multiplier 20. The coefficient multiplier 20 multiplies the input primary frequency command ωos by an integer P, and obtains the frequency f of the synchronous carrier wave.
The operation of the embodiment configured as described above, which is designed to control c, will be explained.

First, before time t■ shown in FIG. 2, the speed command ωR
Assuming that S is maintained below the predetermined value ωL, the output of the mode determining means 16 is at the 14 H It level, and both the switching means 8 and 17 are maintained on the A side. At this time, since the A side of the switching means 17 is grounded, the output δ of the delay element 18! Since S is held at "o", the correction system of the differentiator 19 does not function. Therefore, the vector control device shown in FIG. 1 operates in the same asynchronous manner as the conventional one.
In other words, the phase of the voltage command U 1g is the primary frequency command ω
, integrate S? It is controlled by the added value of the magnetic flux phase command OS and the phase difference angle command δS. Moreover, the torque current component L'l is calculated based on the above-mentioned magnetic flux phase command OS (1).
It is calculated based on the formula and fed back. on the other hand,
The PWM modulation means 7 is connected to the asynchronous carrier generation means 9 by the switching means 8, and compares the given carrier wave EC consisting of a triangular wave with a constant frequency with the three-phase voltage commands U and s to generate a PWM signal. The ignition of the inverter switch element of the inverter circuit 2 is controlled. Since the PWM control at this time is a so-called asynchronous method, the phase relationship between the voltage command U■S and the carrier wave eC is not constant and varies according to changes in the phase difference angle δVS. However, since the frequency fc of the carrier wave ec with respect to the primary frequency command ω1s is sufficiently high, problems such as current beat do not occur even if there is a slight phase shift.

Next, at time t in FIG. 2, when the speed command ωRS exceeds the predetermined value ωL, the output signal DF of the mode discrimination means becomes 1j
This causes the switching means 8 and 17 to switch to the B side.As a result, the delay element 18
The phase difference angle command δS at that time is input to the output δI.
s gradually increases with the time constant of the delay element 18, and becomes a signal that reaches the input value δS after a time ΔT. This delayed phase difference angle command δIs is differentiated by a differentiator 19 and output as a correction frequency ωδ having the changes shown in FIG. This correction frequency ωδ is determined by the adder 11 as the primary frequency command ω, S
A primary frequency command ωas (=ω, S+ωδ) corrected according to the internal phase difference angle is obtained. As is clear from the change in ωδ shown in Fig. 2, this primary frequency command ωos is corrected by ωδ once after ΔT time.
1. only during the ΔT time when the phase difference angle δS changes. The next frequency command ωos is increased or decreased. Primary frequency command ω corrected by this change in phase angle
as is multiplied by P by the coefficient multiplier 20 and input to the synchronous carrier generation means 10. As a result, the synchronous carrier wave generation means 1o generates a carrier wave ec of a frequency fc corresponding to P·ωas, and outputs it to the PWM modulation means 7 via the switching means 8. In other words, transportation? The frequency fc of eC is the correction frequency ωδ
It will be increased or decreased according to the increase or decrease in the output only during the period that is being output.

As a result, the carrier wave ec changes according to the change in the phase difference angle command δ3.
The phase of the carrier wave ec is apparently advanced or retarded, and the phase of the carrier wave ec is adjusted to follow the phase change of the voltage command U S accompanying the change of the phase difference angle command δS. Prevented. Therefore, according to this embodiment,
Stable and high-performance speed control can be performed over a wide range from low speeds to high speeds.

Note that the adder 13 subtracts δts from the magnetic flux phase command θ! input to the torque current component detection means 3. This is because S needs to be the OS with the influence of the phase difference angle δS removed.

Similarly, in the adder 14, the phase difference angle command δS is changed to δ! The reason why S is subtracted is to cancel out δIs added to the magnetic flux phase command OS by the differentiator 19 and the integrator 12.

Note that the above-mentioned operation has been explained for synchronous switching when the speed command ωRS increases, but the operation when the speed command ωR8 decreases and changes below ωL and switches to asynchronous is shown by the dotted line in Fig. 2. This is a similar action. Further, even when the phase difference angle command δS increases or decreases during the synchronous mode, a similar operation prevents a phase shift between the voltage command U1g and the carrier wave ec.

Furthermore, the function of the delay element 18 in the embodiment of FIG. 1 is to slowly start the correction based on the phase difference angle δS upon switching from asynchronous to synchronous, and to suppress sudden changes in the correction due to sudden changes in the phase difference angle δS. It's about doing. ,: This allows smooth correction of phase shifts caused by synchronization switching, etc.

It goes without saying that the above-described embodiments can be realized by software using a microcomputer. In this case, the sampling period (interrupt period) Tc explained in equation (3) above is changed to ωo corresponding to the input of the integrator 12 in FIG.
Change it according to s=ωis+dδts/dt. f!
The t-pressure command U1S and the carrier wave ec are synchronized. That is,
One cycle of voltage command U1s? Tc is changed so that the number of pulses of the PWM signal becomes an integer number. FIG. 3 shows the phase relationship and sampling timing between the voltage commands U and S and the carrier wave ec at this time. The upper line diagram of the same figure is ωo above.
The magnetic flux phase command OS, which is the integral value of s, is shown, and Δt in the figure corresponds to the integral step width of the integrator 12. The bottom diagram is U
(2) This diagram schematically shows the relative phase relationship between S and ec, and the frequency fc of eC is actually higher. Note that although the figure shows an example in which the zero-crossing point of the voltage command UES and the sampling timing coincide, the meaning of synchronization in the present invention is not limited to such synchronization. In short,
Regardless of the change in the internal phase difference angle δS, if the frequency ratio of the voltage command U1S and the carrier wave ec is an integral multiple, the pulse of the PWM signal for every half cycle or one cycle of the waveform of the voltage command U, s This makes it possible to prevent current beats from occurring.

Here, for comparison, FIG. 4 shows the phase relationship of the conventional example corresponding to FIG. 3. As shown in the upper diagram of the figure, is the phase OS of the voltage command Uis the primary frequency? While the phase difference angle command δS is added to the integral value of the order ωXs, the frequency fc of the carrier wave ec is determined by the primary frequency command ω1S, so there is a phase difference corresponding to the phase difference angle δS. Therefore, the relative phase changes in accordance with the change in the phase difference angle δS, and the number of pulses of the PWM signal for a half cycle or one cycle of the voltage command US changes, so the occurrence of current beats cannot be avoided.

FIG. 5 shows another embodiment of the present invention. This example is the first
The difference from the illustrated embodiment is that the switching means 17, delay element 18, and adders 14 and 15 are omitted. In other words, the presence or absence of correction based on the phase difference angle δS is not switched depending on the speed, and the differential value ωδ of the phase difference angle δS is always used as the primary frequency command ω,S.
The reason is that it is added to . Therefore, the phase of the voltage command UES is controlled by the integral value OS of the added value ωos. Furthermore, the frequency and phase of the carrier wave ec during synchronization are controlled based on the added value ωos.

In this way, according to this embodiment, the phase difference angle δS is taken into consideration in the phases of the voltage commands U and S and the carrier wave ec.
Even if PWM control is switched asynchronously and in the same period, changes in phase shift can be reduced, and changes in the phase difference angle δS are reflected gently in the voltage command, reducing the occurrence of current beats and overcurrents associated with switching. This can be prevented and the configuration can be simplified compared to the embodiment shown in FIG.

Note that since the output of the differentiator 19 generally has a finite value, according to this embodiment, the phase difference angle δ8 is not reflected in the magnetic flux phase command OS, which may cause an unstable phenomenon, especially when starting the electric motor. . In order to prevent this, the initial value OOS of the integrator 12 is set to δS at the time of startup, as shown by the dotted line in FIG.

〔Effect of the invention〕

As explained above, according to the present invention, the differential value of the internal phase difference angle of the induction motor is added to the frequency command value, and the frequency of the synchronous carrier wave is controlled by this added value. The frequency of the synchronous carrier wave is increased or decreased in accordance with the phase change of the voltage command, which changes according to the voltage command. As a result, it is possible to prevent a phase shift between the voltage command and the synchronous carrier wave, and it is possible to prevent the occurrence of current beats.

[Brief explanation of the drawing]

Fig. 1 is a block diagram of an embodiment of the present invention, Fig. 2 is an explanatory diagram of the operation, Fig. 3 is an explanatory diagram of the operation of an embodiment configured using a microcomputer, and Fig. 4 corresponds to Fig. 3. FIG. 5 is a block diagram of another embodiment of the present invention, and FIGS. 6 and 7 are diagrams for explaining the background art of the present invention. 1...Induction motor, 2...Inverter circuit, 3...
Torque current component detection means, 4...frequency calculation means, 5
...Voltage command calculation means, 6.. Phase voltage command generation means, 7.. PWM modulation means, 8.. Switching means, 9..
Asynchronous carrier wave generation means, 10...Synchronized carrier wave generation means, 12...Integrator, 16...Mode discrimination means, 17
. . . switching means, 18 . . . delay element, 19 . . . differentiator, 20 . . . coefficient unit.

Claims (1)

[Claims] 1. A PWM signal is generated by comparing the output voltage command of the inverter with a PWM carrier wave having a frequency that is an integral multiple of the frequency of the command, and the inverter is controlled by the PWM signal to operate the induction motor. A vector control method for an induction motor, comprising correcting a frequency command related to frequency determination of the PWM carrier wave based on a differential value of an internal phase difference angle command of the induction motor. 2. Determine the phase and frequency of the inverter's output voltage command based on the induction motor's magnetic flux phase command and internal phase difference angle command, and also determine the frequency that is an integral multiple of the inverter's output frequency command determined based on the induction motor's speed command. A vector control method in which a PWM carrier wave having a frequency of A vector control method for an induction motor, characterized in that the vector control method is determined based on a value obtained by adding a differential value of the internal phase difference angle to a frequency command. 3. Let a microcomputer perform the generation process of the PWM signal, set a PWM carrier wave function whose cycle matches the interrupt cycle related to the execution of the processing operation for generating the PWM signal, and set the PWM carrier wave function and the voltage command. P based on
3. The vector control method for an induction motor according to claim 1, wherein the pulse width of the WM signal is calculated and the interrupt period is corrected based on a differential value of the internal phase difference angle command. 4. Integrating means for integrating the output frequency command of the inverter determined based on the speed command of the induction motor and outputting a magnetic flux phase command; voltage command generation means for generating an inverter output voltage command having a frequency; and frequency correction means for differentiating the internal phase difference angle command and adding it to the inverter output frequency command input to the integrating means; carrier wave generating means for generating a PWM carrier wave having a frequency that is an integral multiple of the output frequency command of the inverter; PWM modulating means for generating a PWM signal by comparing the carrier wave and the output voltage command and outputting the PWM signal to the inverter; A vector control device for an induction motor, comprising: 5. Frequency calculation means for calculating and outputting an output frequency command of the inverter based on the detected torque current component and the given speed command; voltage command calculating means for calculating and outputting an inverter output voltage magnitude command and internal phase difference angle command; integrating means for calculating and outputting a magnetic flux phase command by integrating the output frequency command; a voltage command generating means for calculating and outputting an output voltage command having a phase, frequency, and magnitude of an output voltage based on the sum of the phase command and the internal phase difference angle command and the output voltage magnitude command; asynchronous carrier wave generating means for generating and outputting a PWM carrier wave of a frequency based on an input frequency command; and a synchronous carrier wave generating means for generating and outputting a PWM carrier wave having a frequency based on an input frequency command; P that compares the carrier wave output from the P and the voltage command to generate a PWM signal and output it to the inverter.
A WM modulation means compares the speed command with a predetermined value, and when the speed command is less than the predetermined value, switches the switching means to the asynchronous carrier generation means side, and when the speed command exceeds the predetermined value, switches the switching means to the side of the asynchronous carrier generation means. mode discrimination means for outputting a mode discrimination signal for switching to the synchronous carrier generation means side; and a differentiation means for taking in the internal phase difference angle command in response to the synchronous mode signal of the mode discrimination means and outputting a differential value obtained by differentially processing the command. A vector control device for an induction motor, comprising: means for controlling the frequency of a PWM carrier wave generated by the synchronous carrier wave generation means based on a value obtained by adding the differential value and the output frequency command. . 6. The vector control device for an induction motor according to claim 5, wherein the differentiating means takes in the internal phase difference angle command via a delay element.