JPH0226420B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an antenna input circuit in a radio receiver, and particularly to an antenna input circuit suitable for application to a vehicle-mounted AM radio receiver.

The antenna used for car radio receivers is FM.
It has an appropriate length for the band, and when it resonates, it appears to be almost pure resistance. However, the antenna
Regarding the AM band, it is too short. For this reason, antennas in AM radio receivers generally appear to be capacitive. On the other hand, radio receivers in recent years are electronically tuned, and are tuned by making the C component of the LC resonant circuit variable using, for example, a variable cap (variable capacitance diode). However, when changing the capacitance value of the varicap, a problem arises because the capacitance component of the antenna described above is added to the capacitance of the varicap. That is, since a fixed antenna capacity is always attached, it is inconvenient that the resonant frequency, especially in a high frequency region, cannot be covered. In order to solve this problem, two types of antenna input circuits have already been proposed. The first type is to be referred to as a non-tunable antenna input circuit (described later), and the second type is to be referred to as a tunable antenna input circuit (described later). The first type receives the signal from the antenna side with a FET and then provides a resonant circuit, and the second type receives the signal from the antenna side with a resonant circuit and then provides an amplifier.

Comparing these first and second types of antenna input circuits, each has advantages and disadvantages.
First, in the first type, since the FET receives inputs of all frequencies equally, it has the disadvantage that intermodulation occurs due to interference waves near the desired frequency. However, since the resonant circuit is connected through the high impedance on the FET drain side, the resonant circuit can be constructed relatively easily and has the advantage that the load on the varicap is light. On the other hand, the second type has the advantage that since it first enters the resonant circuit from the antenna side, there is no need to worry about cross-modulation due to the above-mentioned interference waves. However, if the capacitance on the antenna side is not set to a specific value, mismatching will occur, and if the Q (sharpness) of the resonant circuit is not set high, the S/N will be poor.
However, there are disadvantages such as the signal-to-noise ratio (signal-to-noise ratio) being deteriorated and the variable cap having a relatively wide range of capacitance change required.

Therefore, an object of the present invention is to take into account the advantages and disadvantages of the first and second types of antenna input circuits, and basically to create a system that causes cross modulation based on the first type of antenna input circuit. Difficult and low Q
However, we have developed an antenna input circuit that does not significantly degrade the S/N ratio, can match antennas of any capacity, and has various advantages such as a narrow capacitance variation range of the variable cap. It is to provide.

In accordance with the above object, the present invention provides an antenna input circuit of the first type.
Focusing on the fact that the g _n (mutual conductance ₎ of the FET has nonlinear characteristics, it induces waveform distortion, which causes intermodulation of the interference wave. further introduce a transistor or other FET into the FET, and connect the drain of the FET and the base of the transistor or the other FET.
The gate of the FET, the source of the FET, and the collector of the transistor or the drain of the other FET are each connected in an alternating current manner.

The present invention will be explained below with reference to the drawings.

FIG. 1 is a circuit diagram showing a conventional non-tunable antenna input circuit that forms the basis of the present invention, and FIG. 2 is a circuit diagram showing another type of conventional tunable antenna input circuit. In Figure 1, the left side of the center
ANT represents the antenna side, and RD on the right side of the center represents the radio side. E _a represents the antenna input signal voltage, C _a is the capacitance component of the antenna in the AM band as described above, and C _b is the capacitance component of the cable from the antenna to the radio side. When entering the radio side, the antenna input signal first goes to FET1.
1 and guided to the LC resonant circuit 12 via the FET 11. Circuit 12 includes a variable cap 13 and a tuning transformer 114. On the other hand, the second
In the figures, the same reference numbers and symbols as in FIG. 1 indicate the same components.
The configuration of the resonant circuit 22 is slightly different from the resonant circuit 12 of FIG. 1, as shown. This resonant circuit 22
After that, the antenna input signal is amplified by an amplifier 25.

By the way, the present invention is based on the first type of antenna input circuit shown in FIG. This has the serious drawback that the sound enters the audio output. Therefore, we will investigate the cause of this poor cross-modulation characteristic. Focusing on the FET 11 provided in the first stage of the radio side RD in Fig. 1, the FET 1
The equivalent circuit of No. 1 is as shown in FIG. 3A, and its V _{GS -ID} curve is as shown in the graph of FIG. 3B. In FIG. 3A, G is the gate, D is the drain, and S is the source, and when receiving the voltage E, the FET 11 functions as a current source of g _n E. g _n is the transconductance and the gate
If we plot the V _GS - ID curve as shown in the graph of Figure 3B, with the source voltage V _GS on the horizontal axis and _{the drain current ID on} the vertical axis, the mutual conductance g _n
is expressed as g _n =ΔI _D /ΔV _GS . That is, g _n is expressed by the differential value of the curve 31. If we pay attention to this curve 31, we can see that it is not a straight line but a quadratic curve. Therefore, in the case of small amplitude input (∆V _GS is small), g _n can be regarded as constant to some extent, but when the amplitude becomes large (∆V _GS is large), curve 3
Due to the curvilinearity of 1, g _n cannot be considered constant.
That is, g _n has nonlinearity. This nonlinearity causes distortion in the output signal, resulting in cross-modulation due to interference waves. In the end, the amplification factor of FET11 is
The problem of cross modulation lies in the dependence on the nonlinearity g _n .

First, we will try to make this g _n invisible. Figure 4 shows the radio side RD in Figure 1.
FIG. In this figure,
For FET11, E _i is the input voltage, E _p is the output voltage,
Z _L is the impedance of the load element (resonant circuit 12) 41. There is a relationship between E _i , E _p and Z _L as shown in equation (1) using the above g _n .

E _p = E _i g _n Z _L (1) As seen in equation (1), in the circuit shown in Figure 1, g _n is visible as it is, so nonlinear distortion appears significantly. Therefore, as shown in Figure 5,
A passive element 51 having an impedance Z _S is added to the source S side of the FET 11. This constitutes the basic circuit of the present invention. In this circuit, the relationship of equation (2) holds between E _i , E _p , Z _L and Z _S using the g _n described above.

E _p = E _i g _n Z _L /1 + g _n Z _S ………(2) Here, if Z _S in equation (2) is chosen to be large,
If g _n Z _S ≫1 and the numerator and denominator are divided by g _n Z _S , E _p E _i Z _L /Z _S (3) can be obtained approximately. As a result, the cause of nonlinear distortion is
g _n no longer appears in the input/output relationship between E _p and E _i , and the cross-modulation characteristics are improved. Thus, FET
The principle of the present invention is to provide a passive element (Z _S ) 51 at , and to select g _n Z _S ≫1.

In equation (3) expressing the above principle, FET11
Since the gain of is proportional to Z _L /Z _S , choosing a large Z _S results in a decrease in gain. Therefore, it is more preferable if a measure is taken to prevent this gain from decreasing. Therefore, as shown in FIG. 6, the present invention connects a passive element 62 corresponding to the impedance Z _S to a series circuit of LC, that is, as shown in FIG.
Coil 63 with reactance L _p and capacitance C _p
variable capacitor 64, e.g., a variable cap,
It consists of And at the desired frequency
The passive element 62 of the LC series circuit is made to resonate. In this way, the impedance Z _S becomes very small at the desired frequency, so the gain described above does not decrease. In this case, Z _S in equation (2) above is expressed as Z _S =1-ω ² L _p C _p /jωC _p (4), and at the desired frequency, 1-ω ² L _p C _p L _p and C _p are determined so that =0. In this case, Z _S
is substantially zero, which creates a contradiction in the above-mentioned g _n Z _S ≫1. However, since g _n Z _S ≫1 is not actually satisfied at the desired frequency, it has nothing to do with cross-modulation due to interference waves in the vicinity. In other words, in order to suppress cross-modulation, Z _S needs to be large at least at a point outside the desired frequency.
Z _S [=1-ω ² L _p C _p /jωC _p ] sufficiently satisfies these requirements. The circuit configuration shown in FIG. 6 is the premise of the present invention. In addition, in this circuit configuration, since the passive element 62 functions as a resonant circuit, the load element 61 of impedance Z _L functions as the resonant circuit 1 in FIG.
There is no need to form two circuits. The present invention
The circuit configuration shown in Figure 6 has been further improved to further reduce the effect of the nonlinear distortion factor mentioned above, that is, the effect of g _n , which is a factor of cross modulation, and to increase Q, improving the ability to eliminate interference waves. It is something. FIG. 7 is a circuit diagram showing an embodiment of the present invention. 8 is an equivalent circuit diagram of the circuit shown in FIG. 7. Note that portions corresponding to the gate, drain, source, base, emitter, and collector in FIG. 2 are shown with the same alphanumeric symbols. As shown in FIG. 7, a transistor 71 is newly introduced in this embodiment, and FET1 is connected to the base and collector of the transistor 71, respectively.
The drain and source of 1 are connected. If these transistor 71 and FET 11 are represented as current sources, the current flowing through each part in FIG.
g _n e(1+β), g _n e at the point, g _n eβ, and g _n e(1+β) at the input of the passive element 61. however,
As shown, e is the voltage between the gate and source, and β is the current amplification factor of the transistor 71. The semicircular loop at the other end means that the base current is controlling the current source. From FIG. 8, the following relationship can be obtained: E _i = _gn e(1+β)・Z _S +e (5). Then, e is calculated from this, e=E _i /1+g _n Z _S (1+β) (6). Also, the output voltage E _p is E _p = -g _n e (1 + β) Z _L ...... (7), so by substituting equation (6) into this, E _p = -E _i g _n (1 + β) Z _L /1+g _n (1+β)Z _S ...
…(8) is obtained. Comparing this equation (8) with the above equation (2), we get
It can be seen that g _n Z _S and g _n Z _L are each multiplied by (1+β). Since β can generally be around 100, the above-mentioned g _n Z _S ≫1 can be achieved without increasing the value of Z _S.
The following relationship is obtained. In the end, without increasing Z _S , the relationship E _p E _i Z _L /Z _S (9) similar to equation (3) above is satisfied, and since g _n is no longer visible, nonlinear distortion disappears. Even at frequencies other than the desired frequency, the cross-modulation characteristics are further improved without reducing the usability (increasing Z _S ).

Here, consideration must be given to new nonlinear distortion caused by introducing the transistor 71. That is, it should be noted that the current amplification factor β also has nonlinearity. However, this issue is not worth considering. This is because, as shown in equation (9) above, not only g _n but also β become invisible in the input/output relationship. Therefore, the introduction of the transistor 71 causes almost no adverse effects. On the contrary, we should pay attention to the fact that it has even more positive effects. This is due to the fact that in the case of the circuit configuration shown in FIG. 6, the Q cannot be made very high, but in the case of the embodiment of the present invention shown in FIG. 7, the Q can be made quite high. Here, an equivalent circuit of the resonant circuit of FIGS. 6 and 7 will be shown, and it will be as shown in FIG. 9. In this figure, r is the resistance of FET11 in Figure 6, and the resistance in Figure 7 is
The resistance of the FET 11 and the transistor 71 is represented by r=1/g _n in FIG. 6, and r=1/g _n (1+β) in FIG. This is because in Figure 6, I _S = E _i [1/1/g _n + Z _S ] (I _S is the source current of FET 11), and [1/g _n + Z _S ] is the impedance of the resonant circuit. ,
This 1/g _n corresponds to r. On the other hand, in Fig. 7, I _S = g _n e (1 + β) holds true from Fig. 8, and e = E _i /1 + g _n Z _S (1 + β) holds from the above equation (6), so I _S = g _n (1+β)E _i /1+g _n Z _S (1+β) is obtained, which is transformed to obtain I _S =E _i [1/1/g _n (1+β)+Z _S ]. Of these, [1/g _n (1+β)+Z _S becomes the impedance of the resonant circuit, so 1/g _n (1+β) corresponds to r.

Thus, it can be seen that r in the circuit configuration of FIG. 6 and the embodiment of the present invention in FIG. 7 is expressed by 1/g _n and 1/g _n (1+β), respectively. Since Q is determined by Q=ωL/r, the smaller r is, the higher Q can be obtained. In other words, since r is smaller in the present invention [1/g _n (1+β)<1/g _n , Q can be higher.

Finally, an example of how to take out the output from the antenna input circuit of the present invention will be shown. FIG. 10 is a principle circuit diagram showing an example of this. In this figure, the components 11, 71, 62, 63, and 64 are as already explained, and the secondary side of the transformer with the coil 63 as the primary winding. The output voltage E _p can be extracted from. Of course, transistor 71
It is also possible to extract the output voltage E _p from the emitter of .

FIG. 11 is a circuit diagram showing a specific example of the principle circuit shown in FIG. 10, and the components already described are designated with the same reference numbers or symbols. However, variable capacitor 64 is shown as a varicap. A resistor R5 is a bias resistor for applying a bias voltage, that is, a tuning voltage TV, to the variable cap 62. Resistors R1 and R2 are bias resistors for FET 11, capacitor C1 is a bypass capacitor, _L1 is a choke coil for making the source side of FET 11 high impedance in AC, capacitor C2 is a coupling capacitor, and C3 is a bypass capacitor.
Resistors R3, R4 and R6 are each bias resistors.

As explained above, according to the present invention, the cross-modulation characteristics are improved, the S/N ratio does not deteriorate even when the Q is low, the antenna can be matched to any capacity antenna, and the variation in capacitance of the varicap can be adjusted. An antenna input and a circuit can be realized which have various advantages such as a narrow range, a high Q value, improved selectivity, and an improved ability to reject interference waves.

[Brief explanation of drawings]

Fig. 1 is a circuit diagram showing a conventional non-tunable antenna input circuit which is the basis of the present invention, Fig. 2 is a circuit diagram showing another type of conventional tunable antenna input circuit, and Fig. 3A is an equivalent of FET11. Circuit, Figure 3B is a graph showing the V _GS -I _D curve of FET 11 in Figure 3A, Figure 4 is a generalized diagram of the radio side RD in Figure 1, and Figure 5 is the premise of the present invention. FIG. 6 is a diagram showing the principle configuration of the antenna input circuit. FIG. 6 is a circuit diagram showing the antenna input circuit which is the premise of the present invention.
The figure is a circuit diagram showing one embodiment of the antenna input circuit of the present invention, FIG. 8 is an equivalent circuit diagram of the circuit of FIG. 7, and FIG. 9 is an equivalent circuit of the resonant circuit in FIGS. 6 and 7. The circuit diagram, FIG. 10 is a principle circuit diagram showing an example of a method for extracting the output voltage E _p from the circuit according to the present invention, and FIG. 11 is a circuit diagram showing a specific example of the principle circuit shown in FIG. 10. . In the figure, 11 is a FET, 51 is a passive element,
62 is a resonant circuit, 63 is a coil, 64 is a capacitor, and 71 is a transistor.

Claims (1)

[Scope of Claims] 1. An antenna input circuit of a type in which an input signal from an antenna in a radio receiver is once received at the gate of an FET and then supplied to a resonant circuit, the source of the FET having a mutual conductance g _n In an antenna input circuit including a passive element with an impedance of _Zs connected to the side, a transistor with a current amplification factor of β is introduced into the FET, and the drain of the FET, the base of the transistor, and the source of the FET are connected to each other. The collectors of the transistors are connected in an alternating current manner, and the Z _S is set in a frequency range other than the reception frequency range.
An antenna input circuit characterized in that the antenna input circuit is set to a value that satisfies g _n (1+β)Z _S ≫1.