JPH0219660B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Technical Field of the Invention] The present invention relates to a spread spectrum receiving apparatus that demodulates a data signal from a spread spectrum received signal.

[Technical background of the invention and its problems] Spread spectrum (hereinafter abbreviated as "SS") communication system is also called code division multiplex communication system, and is used in, for example, satellite communication. Unlike time division multiplex communication systems and frequency division multiplex communication systems, this SS communication system does not require central control to control signal transmission and allows random access.

In SS communication method, PSK (Phase Shift Keying)
SS signals are often created using the direct spreading method and sent from the transmitting side to the receiving side. In this case, the SS signal is generated, for example, by adding an information signal to be transmitted with a high-speed pseudorandom (PN) code sequence signal, and then phase-modulating a carrier wave in accordance with the addition result.

On the receiving side, the SS signal is received by the SS receiving device, and the information signal is demodulated through, for example, the following steps or operations equivalent to these steps.

(a) Searching for the start position of the PN code sequence of the input SS signal (b) Generating a spread demodulation code sequence synchronized with the PN code sequence of the input SS signal (c) Searching for the start position of the PN code sequence of the input SS signal (c) Searching for the starting position of the PN code sequence of the input SS signal Multiplication by the code sequence (d) Extraction of the carrier wave from the PSK modulation signal obtained in (c) above (e) Multiplication of the carrier wave reproduced in (d) above and the PSK modulation signal These (a) to (e) By performing the following operations, it is possible to demodulate the information signal from the received SS signal.

However, in conventional SS receivers, these operations (a) to (e) are performed individually, which results in a complex demodulation section and a lack of stability in circuit operation. Furthermore, if the PN code sequence length is long, the above control operation (a) takes time, so there is a drawback that the synchronization establishment time in the above (b) becomes long.

On the other hand, there is a device shown in FIG. 2 as an SS receiving device that eliminates these drawbacks. That is, this second
In the SS receiver shown in the figure, the received SS signal is supplied to multipliers 21 and 22, and the multiplier 21 multiplies (mixes) the output signal of the local oscillator 23 and the multiplier 22 multiplies (mixes) the output signal of the phase shifter 24. be done. Phase shifter 2
4 is a 90° phase shifter that provides a 90° phase change to the output signal of the local oscillator 23.

Now, the angular frequency of the carrier wave is ω, the PN code (0 or π) is P(t), and the information (data) to be transmitted is
(0 or π) as θ(t), and the phase error as φ
Then, the received SS signal is cos{ωt+P(t)+θ
(t)+φ}. In addition, the local oscillator 23
The output signal of the phase shifter 24 is expressed as cosωt, and the output signal of the phase shifter 24 is expressed as sinωt.

Therefore, the multipliers 21 and 22 output the DC component, the frequency component of the carrier wave, and its harmonic components. Low-pass filters 25 and 26 receive signals cos{P from the outputs of multipliers 21 and 22, respectively.
(t)+θ(t)+φ}, sin{P(t)+θ(t)+
φ}
Extract. The output signals of the low-pass filters 25 and 26 are converted into digital signals by A/D converters 27 and 28, respectively, and then supplied to matched filters 29 and 30, respectively. The matched filters 29 and 30 are configured to filter the PN included in the input signal.
It detects the match between the code P(t) and a predetermined PN code unique to the receiving side. When both PN codes match, the data signals cosθ(t), sinθ
(t) respectively. That is, the matted filters 29 and 30 are
It is configured to correlate the PN code with an input digital signal, and is configured to sequentially obtain, for example, the sum of the products of individual values of the PN code and each input digital signal value obtained from time to time. If the receiving device
When the PN code and the PN code of the input digital signal match, the sum of the products becomes large, and the matched filters 29 and 30 output signals with large peaks to detect the match of the PN codes. If the PN codes do not match, the sum of the products described above will have no peak and the output will be nothing more than noise. Therefore, data signals cosθ(t) and sinθ are output from the matched filters 29 and 30 only when the PN codes match.
(t) is output, so there is no worry about interference between receiving devices or reception. The output signals of these matched filters 29 and 30 are squared by a squarer 3
1 and 32, and after being squared, the adder 3
3 is added. When the addition result is above a predetermined level, the filter 34 detects the beginning position of the PN code sequence signal in the received SS signal and outputs a signal.

Further, the output signals of the matched filters 29 and 30 are supplied to a multiplier 35, and the output of the multiplier 35 is supplied to a loop filter 36 that controls the local oscillator 23. As a result, the phase error component φ included in the initial state of reception is removed. That is, in the initial state, the multiplier 35 calculates cos {θ(t)+φ}×sin{θ
(t)+φ}=1/2sin{2θ(t)+2φ}=sinφ
An output of φ (θ (t) = 0, π) is obtained, which is passed through a loop filter 36, integrated, and based on this result, is sent to a local oscillator 23 (for example, a voltage controlled oscillator).
The phase of the output is controlled so that φ=0.

This control is performed only when the filter 34 detects the leading position of the received PN code.

Demodulated data is obtained by determining the output of the matched filter 29.

In the configuration shown in FIG. 2, since the filter 34 detects the beginning position of the received PN code sequence, the above-mentioned
The search operation in (a) is not necessary, and the synchronization establishment time is shortened.

Further, the signal processing after the output signals (baseband signals) of the low-pass filters 25 and 26 can be easily digitized, and stable operation can be expected.

However, with this configuration, if the number of quantization bits is increased in order to obtain high-quality demodulated data, the scale of the matched filter section increases, making it difficult to obtain a simple receiving device. On the other hand, if the number of quantization bits is reduced in order to simplify the configuration, the deterioration of the signal amplitude due to quantization increases accordingly, and the quality of demodulated data deteriorates.

[Purpose of the invention]

The present invention has been made in consideration of the above-mentioned circumstances, and an object of the present invention is to provide an SS receiving device that has a simple configuration and can suppress deterioration of demodulated data quality.

[Summary of the invention]

The SS receiving device according to the present invention is provided with a code generating means for outputting a pseudo-random code sequence signal synchronized with the pseudo-random code sequence component included in the SS signal, and a carrier frequency component is obtained from the output signal of the code generating means and the received SS signal. The data signal is extracted by multiplying by the baseband SS signal extracted by removing the .

[Embodiments of the invention]

An embodiment of the SS receiving device according to the present invention will be described below with reference to FIG.

FIG. 1 is a schematic configuration diagram illustrating an embodiment of an SS receiving device according to the present invention. In FIG. 1, the same components as in FIG. 2 are given the same numbers, and the different components will be mainly explained. That is, in FIG. 1, when the filter 34 detects a leading position in the PN code sequence included in the received SS signal, the output signal of the filter 34 is supplied.
A PN code generator 37 is provided. The PN code generator 34 generates a PN code sequence signal based on the detection signal of the leading position of the PN code sequence signal from the filter 34. The filter 34 can detect the start position of the PN code sequence signal in at least one cycle of the received SS signal, but it is better to detect the start position of the PN code sequence signal multiple times over multiple cycles when the reception condition is, for example, noisy. You can accurately determine the starting position. The PN code generator 37 outputs a PN code sequence signal in synchronization with the received PN code of, for example, the next cycle of the cycle in which the start position of the received PN code was determined. The output signal of this PN code generator 37 is supplied to a multiplier 38. The output signal of the low-pass filter 25 is also supplied to the multiplier 38, and the output of the low-pass filter 25 is multiplied by the output of the PN code generator 37. The output signal of the low-pass filter 25 is a baseband SS signal obtained by removing the carrier frequency component from the received SS signal.
Since a local oscillation signal that follows the phase of the received SS signal carrier wave is obtained from the local oscillator 23 in a closed loop, the output of the low-pass filter 25 does not include a phase error. Therefore, in the multiplier 38,
Multiplication can be performed by matching the leading positions of the PN code sequence included in the SS signal and the output PN code sequence of the PN code generator 37. As a result, multiplier 3
8 outputs a signal from which the PN code sequence component has been removed. This output signal is supplied to a filter 39, high frequency components are removed, and the data signal is demodulated. This demodulated data is passed through the matted filter 29
This is a signal obtained as a result of multiplication of the SS signal before being input to the SS signal and a PN code sequence signal synchronized with the PN code sequence included therein, and the data signal has not been degraded by the matched filter 29. That is,
When the matched filter 29 is constructed from a digital circuit, the input signal is quantized, so that the multiplication results and addition results within the matched filter are degraded by the quantization and the signal-to-noise ratio is lowered. However, in the configuration shown in FIG.
Since the SS signal before being input to the filter 9 is used, it is not degraded by the matched filter 29. For example, if the same demodulated data error rate is obtained and the signal input to the matched filter 29 is 1-bit quantized, the configuration shown in FIG.
The transmission power may be 2 dB lower than in Figure). That is, in the configuration of FIG. 1, since it is not affected by deterioration in the matched filter 29 caused by quantization,
The signal power to noise power ratio can be improved by 2 dB under the above conditions. Therefore, if the transmission power is the same between the configurations in Figures 1 and 2, the configuration in Figure 1 can reduce the number of quantization bits more to obtain the same demodulated data quality. This makes it possible to reduce the size of the combined filter section and construct a simpler SS receiving device.

Note that instead of the output of the low-pass filter 25, the output digital signal of the A/D converter 27 may be supplied to the multiplier 38 and multiplied by the PN code sequence.

〔Effect of the invention〕

As explained above, according to the SS receiving device according to the present invention, deterioration in demodulated data quality can be suppressed with a simple configuration.

[Brief explanation of drawings]

FIG. 1 is a schematic diagram illustrating an embodiment of a spread spectrum receiver according to the present invention, and FIG. 2 is a schematic diagram illustrating a conventional spread spectrum receiver. 21, 22, 35, 38...multiplier, 25, 26
...Low pass filter, 27, 28...A/D converter, 29, 30...Matched filter, 31, 32
... Squarer, 33... Adder, 34, 39... Filter, 37... PN code generator, 36... Loop filter, 23... Local oscillator, 24... Phase shifter.

Claims (1)

[Claims] 1. A first extraction means to which a spread spectrum signal including a carrier frequency component, a pseudorandom code sequence component, and a data component is supplied, and which removes the carrier frequency component and outputs a first signal including a pseudorandom code sequence component and a data component. and outputs a second signal containing the data component when the output signal of the first extraction means is supplied and the pseudo-random code sequence component is a predetermined pseudo-random code sequence component. a second extracting means; and a code generating means that is supplied with the output signal of the second extracting means and outputs the predetermined pseudo-random code sequence signal synchronized with the pseudo-random code sequence component included in the first signal. and a third extracting means that is supplied with the output signal of the code generating means and the first signal and multiplies them to extract the data component.