JPH0137709B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a radar device that accurately detects and tracks targets flying at low altitude.

Conventionally, this type of device is a monopulse radar (MN
OPULSE RADAR).
Figure 1a is a block diagram showing the principle of monopulse radar, in which 1 is a monopulse antenna, 2
is a hybrid circuit, 3 is an amplifier, 4 is an automatic gain control device, 5 is a phase detector, 6 is the central axis of monopulse antenna 1, 7 is a target, 8 is an image of target 7, 9 is a reflected wave from target 7 ( 10 is a reflected wave from the image 8, that is, a reflected wave from the target 7 due to the multipath effect (indirect wave), and 11 is the sea surface. FIG. 1b shows the monopulse antenna 1
FIG. 2 is an antenna pattern diagram. Monopulse radar simultaneously receives reflected waves from a target using two partially overlapping antenna patterns as shown in Figure 1b, and calculates the sum signal Σ and difference signal of the two received signals obtained from each antenna pattern. Δ is obtained by the hybrid circuit 2. As a result, the output signal of the hybrid circuit is completely equivalent to the signal received by the two antenna patterns as shown in FIG. In the figure, the horizontal axis shows the reception angle θ (the angle between the antenna center axis 6 and the received reflected wave), and the vertical axis shows the reception voltage V.
12 is a sum pattern, 3 is a difference pattern, 14 is a received voltage Σ _d according to the sum pattern 12 of reflected waves 9,
15 is a received voltage Δ _d based on the pattern 13 of differences in the reflected waves 9 , 16 is a received voltage Σ _i based on the pattern 12 of the sum of reflected waves 10 , and 17 is a received voltage Δ _i based on the pattern 13 of differences in reflected waves 10 . However, Σ, △,
Σ _d , Δ _d , Σ _i , Δ _i are complex signal voltages including amplitude and phase.

First, consider the case where image 8 does not exist. At this time, the monopulse antenna 1 receives only the reflected wave 9 from the target 7, so the received signal voltage Σ due to the sum pattern 12 and the received signal voltage △ due to the difference pattern are respectively Σ=Σ _d = Σ(θ _d ) (1) △=△ _d = △(θ _d ) (2). Here, θ _d is the receiving angle of the reflected wave 9.
The gain A of the amplifier 3 is controlled by the automatic gain control device 4 as shown in the following equation.

A=V _s /|Σ| (3) Here, V _s is a constant determined by the feedback amount of the automatic gain control device.

_{Therefore ,} the received signal voltages after passing _through the amplifier _{3 are} as follows, respectively _: V _s △ _d / | Σ _d | = V _s △ (θ _d ) / | Σ (θ _d ) | (5). The phase detector 4 is a detector that maintains the amplitude and phase difference of two input signals, and when the two complex input signals are I and Q, the output voltage E is E=|I| ² R _e [Q/I] (6) becomes. Here, R _e [ ] represents the real part. Therefore, the two signals expressed by equations (4) and (5) are detected by the phase detector 5.
, the output voltage E becomes E=V ² _{s Re} [Δ _d /Σ _d ]=V ² _{s Re} [Δ(θ _d ) /Σ(θ _d )] (7).

FIG. 3 is a diagram in which the relationship between the reception angle θ _d and the phase detector output voltage E is calculated based on equation (7) and the antenna pattern shown in FIG. As shown in FIG. 3, the output voltage _E and the reception angle θ _d have a proportional relationship in the region where the angle θ _d is not very large. The positional direction θ _d can be accurately known. Furthermore, by feeding back the output voltage E given by equation (7) to the drive section of the monopulse antenna 1 as an angular error signal, the target can be tracked with high accuracy.

On the other hand, when the image 8 exists, that is, when the reflected wave 10 due to the multipath effect and the reflected wave 9 from the target are simultaneously received by the monopulse antenna 1, the voltage Σ received in the sum pattern 12 and the difference pattern The voltage Δ received at 11 is a composite value of two voltages as shown below.

Σ=Σ _d + Σ _i = Σ (θ _d ) + Σ (θ _i ) (8) △=△ _d + △ _i = △ (θ _d ) + △ (θ _i ) (9) Here, θ _i is the reflected wave 10 is the receiving angle.

By substituting equations (8) and (9) into equation (7), the phase detection output voltage E when there is a multipath effect is similarly given by the following equation.

E=V ² _s R _e [△ _d + △ _i / Σ _d + Σ _i ] = V ² _s R _e [△ (θ _d ) / Σ (θ _d )・1 + △ (θ _i ) / △
(θ _d )/1+Σ(θ _i )/Σ(θ _d )] (10) In the absence of multipath effects, the phase detection output voltage E is equal to It was determined only by the angle, but in this case, due to the influence of the reflected wave from the image, in addition to the above parameters, the amplitude ratio and phase difference of the two reflected waves are determined as shown in equation (10). is also affected. This means that the reception angle of the reflected wave from the target cannot be determined from the phase detection output voltage alone, and essentially monopulse radar cannot accurately detect and track the target due to the influence of multipath effects. It becomes possible.

However, conventional radar equipment uses a monopulse radar to detect and track a target flying at low altitude. As a result, the target detection and tracking accuracy is very poor, and in extreme cases, the target may be lost.

In order to eliminate these drawbacks, this invention arranges a plurality of antennas in a straight line in the vertical direction,
The system processes received signals received independently by each antenna according to a predetermined method to remove multipath effects and improve target detection and tracking accuracy.The drawings will be described in detail below.

FIG. 4 shows an embodiment of the present invention, in which:
18 is an antenna, 19 is a transceiver, 20 is a ranging device, 21 is a signal processing device, 22 is a pulse radio wave, 2
3 is a synchronization signal generator. FIG. 5 is a detailed block diagram of one of the transceivers 19 in FIG. 4, in which 24 is a transmitter/receiver switch, 25 is a high output amplifier, 26 is a pulse modulator, and 27 is a reference signal generator. 28 is a low noise amplifier, 29 is a band pass filter, and 30 is a phase detector. FIG. 6 is a diagram showing a transmission signal waveform, in which 31 is a reference signal and 32 is a transmission signal.

Here, for convenience of explanation, the total number of antennas 18 and transceivers 19 will be N, and each antenna and transceiver will be referred to as #1, #2, ..., #N antennas and transceivers from the bottom. . 7th
The figure shows the relationship between the transmission frequency (f ₁ , f ₂ , ..., f _N ) of pulse radio waves generated by each transceiver and the passband of the bandpass filter 29 of each transceiver. , in the figure, 33 is the transmission frequency, 3
4 is the frequency characteristic of the bandpass filter.

As shown in FIG. 6, a transmission signal 32 is generated by a reference signal generating device 27, and a continuous sine wave reference signal 31 whose frequency is equal to the transmission frequency is transmitted to a pulse modulator 26.
It is generated by pulse modulation. The driving of the pulse modulator 26 is synchronized by a synchronization signal T, so that the N transceivers 19 simultaneously generate transmission signals having different frequencies.

Here, the #1 transceiver generates a transmission signal with a transmission frequency f ₁ , the #2 transceiver generates a transmission signal with a transmission frequency f ₂ , and the #N transceiver similarly generates a transmission signal with a transmission frequency f _N. shall be The intervals between these frequencies are selected in consideration of the Doppler frequency shift that accompanies the movement of the target, as will be described later.

The generated transmission signal 32 is amplified by a high-output amplifier 25, and then transferred to a transmission/reception switch 24 and an antenna 1.
The pulsed radio waves 22 are radiated to the outside space via the radio waves 8 and 8 as pulse radio waves 22. That is, pulse radio waves of frequencies f ₁ to f _N are transmitted simultaneously. The pulsed radio waves reflected by the target 7 and the image 8 of the pulsed power source 22 are received by the antenna 18, passed through the transmitter/receiver switch 24, amplified by the low noise amplifier 28, and then inputted to the bandpass filter 29. The bandpass filter 29 is for removing, from among the received signals, received signals caused by pulsed radio waves transmitted from other antennas. For example, with antenna #1, in addition to the received signal with frequency f ₁ + f _d1 ,
Reception signals with frequencies f ₂ + f _d2 , f ₃ + f _d3 , ... f _N + f _dN due to pulsed radio waves transmitted from antennas #2, ... #N are simultaneously received, and the reception signal has a frequency component of f ₁ + f _d1 . Only the signal is extracted by a bandpass filter. Here, f _di (i=1, 2, . . . , N) is the Doppler frequency due to the movement of the target 7, and when the target speed is υ, it is given by the following equation.

f _di =2υ/Cf _i (11) (i=1, 2,..., N) Here, C is the speed of light. However, the sign of υ is positive in the direction toward this device. The frequency characteristics of each bandpass filter in the transmitters/receivers #1 to #N are determined as follows in consideration of this Doppler frequency shift. First, the passband width W _i (i=1, 2,
..., N) is determined by the following formula, with the expected maximum speed of the target being υ _nax .

W _i =4υ _nax /Cf _i (12) (i=1, 2,..., N) Next, the transmission frequency f _i (i=1, 2, ..., N) are determined. The situation is shown in FIG. the result,
Transmitter/receiver #1 can extract only received signals with frequencies f ₁ +f _d1 , and similarly, transmitter/receivers #2 to #N can extract only received signals with frequencies f ₂ +f _d2 to f _N +f _dN .

The received signal that has passed through the bandpass filter 29 is input to a phase detector 30. The phase detector 30 phase-detects the input signal using the reference signal 31,
The signal processing device 21 converts the detected signal into a complex signal.
Enter. Due to the phase detection described above, only the phase corresponding to the propagation distance of the pulsed radio wave is stored in the complex signal, regardless of the phase of the transmitted pulsed radio wave.

The received signals received by the antennas #1 to #N are a composite value of the reflected wave 9 from the target 7 and the reflected wave 10 from the image 8, and therefore, the signal processing device 21
The complex signal S _i input to is expressed as S _i =a _i +b _i (13) (i=1, 2, . . . , N). Here, i indicates the number of the antenna from which the complex signal S _i was received, and a _i and b _i correspond to reflected waves 9 and 10, respectively.

Further, the received signal received by the #1 antenna is used to calculate the distance between the target 7 and the present radar device by a known distance measuring device 20, and is inputted to the signal processing device 21 as a distance signal R _p . The distance signal R _p measured here includes an error due to the influence of the reflected wave 10, but this error can be ignored if the altitude of the target 7 is low enough to cause a multipath effect.

FIG. 8 is a diagram showing the principle of the processing method performed by the signal processing device 21. In the figure, 35 is a virtual target for signal processing, 36 is the plane where the target 7 and image 8 exist, and 37 is the antenna 18. The arrayed plane 38 is an axis indicating the center of the antennas #1 to #N.

The signal processing device 21 performs arithmetic processing as shown in the following equation on the received signal given by equation 13.

V=| _N 〓 ⁱ⁼¹ S _i t _i | (14) Here, t _i (i=1, 2,..., N) is a phase correction coefficient, which is calculated by the signal processing device 21 as follows. Ru. First, the virtual target 35 is set at a position at an elevation angle θ _p (θ _p is the angle formed by the central axis 38 and the direction of the virtual target 35). Here, θ _p is set to a value at which target 7 is considered to exist. At this time, the phase coefficient t _i is calculated by the following formula.

t _i = KeXP [−j4π/λ _i R _i ] (15) Here, λ _i (i=1, 2,...N) is the transmission wavelength, and λ _i = C/f _i (16) There is a relationship . Further, K is an arbitrary constant that is not zero, and R _i (i=1, 2, . . . , N) is the distance between the virtual target 35 and each antenna during transmission and reception, and is given by the following equation.

R _i [(h _M + R _p tanθ _p −h _i ) ² + R ² _p ] ^1/2 (17) (i=1, 2,..., N) Here, h _i (i=1, 2,..., N) is the height of each antenna from the sea surface, and h _M is h _M = (h _N + h ₁ )/2 (18).

The signal processing device 24 performs the above calculations on a virtual target 35.
This is done by sequentially changing the position of , that is, the elevation angle θ _p .
At this time, let the elevation angles for target 7 and image 8 be θ _t and θ _i , respectively, and the elevation angle θ _p is |θ _p −θ _t |<△θ (19) or |θ _p −θ _i |<△θ (20). The calculation result of formula (14) when satisfied is V=
V _IN , and the calculation result of equation (14) when the elevation angle θ _p does not satisfy either equation (19) or (20) is V =
If V _OUT , then V _IN /V _OUT ≫1 (21) holds true.

Here, Δθ is an equivalent antenna beam width obtained by performing the above-described signal processing on the received signals obtained by antennas #1 to #N.

The phase correction coefficient shown by formula (15) is #1~
Phase difference of the received signal due to the difference in round-trip distance between the virtual target and each antenna, assuming that pulse radio waves of different frequencies are transmitted from each #N antenna toward the virtual target and the reflected waves are received. This is to correct and bring them into the same phase.

The situation is shown in FIG. In the figure, 39 is an equiphase plane from the virtual target. Now, assuming that the position of the virtual target 35 coincides with the position of the target 7, the calculation expressed by equation (14) is to arrange the antennas #1 to #N on the plane 39 and target the target 7 with the same frequency. The calculation result is 39.
This is completely equivalent to transmitting pulsed radio waves toward the target using a real aperture antenna with a mirror surface, and then receiving the reflected waves. Therefore, to sequentially change the elevation angle θ _p of the signals received by antennas #1 to #N and perform the arithmetic processing shown in equation (14), it is necessary to sequentially change the beam direction of the real aperture antenna and achieve the target. Or send pulsed radio waves towards the image,
This is equivalent to receiving those reflected waves, and if equations (19) and (20) hold, it corresponds to receiving the reflected waves from the target or image with the main beam of the real aperture antenna, and vice versa. In formula (19),
If (20) does not hold, this corresponds to receiving the reflected waves from the target and the image as side lobes, and the relationship shown in equation (21) holds. Also,
The beam width Δθ can also be considered in the same way as in the case of a real aperture antenna, and is given by the following equation from a known formula giving the two-way beamwidth of a real aperture antenna.

Δθ=λ _M /2(h _N −h ₁ ) (22) Here, λ _M is an equivalent transmission wavelength determined by the number of antennas, the spacing between antennas, etc., and satisfies the following equation.

min 1 (λ _i ) < λ _M < max 1 (λ _i ) (23) As described above, according to the radar device according to the present invention, radio waves are transmitted toward a target using a radar device having a real aperture antenna with a large aperture length in the vertical direction. It is possible to obtain the same angular resolution as that obtained when transmitting and receiving a beam, and the phase given by Equation (15) This has the advantage that it can be done easily and quickly by changing the correction coefficient. In addition, in this radar device, since the frequency of the pulse radio waves transmitted from each antenna element is changed and transmission and reception are performed at each antenna element simultaneously, there is no need to take into account the movement of the target when correcting the phase of the received signal. As a result, the error in phase correction becomes extremely small, and the accuracy of the calculation result of equation (14) becomes high. Furthermore, since this device transmits and receives pulse radio waves using the same antenna, compared to this type of radar device that is configured to transmit and receive pulse radio waves using separate antennas,
The angular resolution is approximately twice as high.

Therefore, by making the range of change in the elevation angle θ _p sufficiently small,
If the calculation of equation (14) is carried out, a calculation result as shown in FIG. 10 will be obtained.

FIG. 10 is a diagram showing the effect of the present invention, and 4
0 and 41 are calculation results corresponding to target 7 and image 8, respectively. As shown in Fig. 10, the target and image can be separated by applying the signal processing described above to the received data received by multiple antennas, and the target and image can be separated by detecting the peak value of the calculation result V. The presence of the target can be detected accurately by removing the influence of multipath, and the target can be accurately tracked by extracting the angle information of the peak position of the calculation result V.

Now, in the above explanation, it was assumed that the ranging data R _p gives the horizontal distance between the target 7 and the antenna element 18 as shown in FIG.
R _p is not the horizontal distance but target 7 and antenna element 1
It gives a slant range of 8. Even if R _p is not a horizontal distance but a slant range, such an error in R _p will be Things that do not affect the effects of the present invention will be explained below.

First, the error occurring in the phase correction coefficient t _i calculated using equations (15) and (17) will be explained.

In equation (17), h _M + R _p tan θ _p indicates the altitude of virtual target 35 from sea level 11, which is
Let h _p , h _p = h _M + R _p tanθ _p (24) Rewriting equation (17) using equation (24) yields the following equation.

R _i = {(h _p h _i ) ² + R _p ² } ^1/2 (25) When the radar device according to the present invention observes a target flying at a low altitude where multipath effects are a problem, the following equation is generally used. holds true.

R _p ≫ (h _p −h _i ) ² (26) Using equation (26), equation (25) can be approximated by the following equation.

R _i = R _p + 1/2R _p (h _p −h _i ) ² (27) On the right side of equation (27), the first term R _p is a constant with respect to the change in i, and R _p is the horizontal distance Even in the slant range, there is no phase correction error at all. Here, the horizontal distance is expressed as R _H , and the slant range R _p is expressed as the following formula: R _p = R _H + △R (28) △R is the difference between the slant range R _p and the horizontal distance R _H , and the target 7 The altitude from the sea level 11 is expressed as _ht by the following formula.

△R=√ _H ² + _t ² −R _H (29) In the radar device according to the present invention, R _H =10Km, h _t =
Since the purpose is to search and track a low-altitude flying target of about 100 meters, equation (29) can be approximated by the following equation.

△Rh _t ² /2R _H (30) From equation (30), it is clear that △R/R _H ≪1 (31). Using equation (31), the second term on the right side of equation (23) can be expressed as the following equation.

Second term on the right side of equation (27) = 1/2R _H (h _p −h _i ) ² −△R/2R _H ² (h _p −h _i ) ² (32) Second term on the right side of equation (32) provides phase fixing because the distance data is a slant range rather than a horizontal distance. The phase correction and the magnitude of the error depend on the transmission wavelength λ. That is, No. (32)
If the second term on the right side of the equation is sufficiently small with respect to λ, there is no problem that the distance data is in the slant range. In the radar device according to the present invention, a transmission wavelength of about λ=0.3m is often used, and
Since the purpose is to search and track low-altitude flying targets, R _H = 10 Km, h _t = 100 m, and h _p −h _i = 200 m at most. At this time, it is clear that the following formula holds true.

△R/2R _H ² (h _p −h _i ) ₂ ≪λ (33) Next, the altitude of the target measured in this way
The error when the target elevation angle θ _t is derived from h _t will be explained. Given the target altitude h _t , the eighth
From the geometrical relationship shown in the figure, the target elevation angle θ _t is expressed by the following equation.

θ _t = tan ^-1 h _t −h _M /R _p (33) Substituting equation (28) into equation (33),
Using equation (31), equation (33) can be expressed as the following equation.

θ _t = tan ^-1 ((h _t −h _M )/R _H −h _t ² (h _t −h _M )/2R _H ³ )
(34) Equation (34) is expressed as Equation (35) using h _t <<R _H and h _M <<R _H.

θ _t h _t −h _M ／R _t −h _t ² (h _t −h _M )／2R _H ³ (35) The second term in equation (35) indicates that the distance measurement data R _p is not the horizontal distance, but the slant distance. Although the error due to the range is shown, it is clear that the second term is sufficiently smaller than the first term.

In this way, the radar device according to the present invention can measure the elevation angle of the target by generating the phase correction coefficient with sufficient accuracy even when using the slant range instead of the horizontal distance, as shown in Fig. 10. effect can be obtained.

As described above, according to the present invention, it is possible to eliminate the influence of multipath effects on targets flying at low altitude and to accurately know the target position.
It is highly effective when used in radar equipment that detects and tracks targets at low elevation angles.

[Brief explanation of drawings]

Figure 1a is a block diagram showing the principle of monopulse radar, Figure 1b is a diagram showing the monopulse antenna pattern, Figure 2 is the antenna pattern diagram of the monopulse antenna, and Figure 3 is the reception angle and phase detector. 4 is a block diagram showing an embodiment of the present invention, FIG. 5 is a block diagram showing the configuration of a transmitter/receiver, FIG. 6 is a diagram showing the transmitted signal waveform, and FIG. 7 is a diagram showing the relationship between output voltages. Frequency characteristic diagram of a bandpass filter. Figure 8 is a diagram showing the principle of the signal processing method. Figure 9 is a diagram showing the principle of the signal processing method.
The figure is a diagram showing the equivalence between this invention device and a real aperture antenna, and FIG. 10 is a diagram showing the effect of this invention device. In the figure, 1 is a monopulse antenna, 2 is a hybrid circuit, 3 is an amplifier, 4 is an automatic gain control device,
5 is a phase detector, 6 is the central axis of the antenna, 7 is a target, 8 is an image, 9 and 10 are reflected waves, 11 is a sea surface, 12 is a sum pattern, 13 is a difference pattern,
14, 15, 16, 17 are received voltages, 18 is an antenna, 19 is a transmitter/receiver, 20 is a ranging device, 21 is a signal processing device, 22 is a pulse radio wave, 23 is a synchronization signal generator, 24 is a transmission/reception switch, 25 is a high output amplifier, 26 is a pulse modulator, 27 is a reference signal generator, 28 is a low noise amplifier, 29 is a band pass filter, 30 is a phase detector, 31 is a reference signal, 3
2 is the transmission signal, 33 is the transmission frequency, 34 is the frequency characteristic, 35 is the virtual target, 36 is the plane where the target and image exist, 37 is the plane where the antenna is arranged, 38 is the central axis of the antenna, 39 is etc. phase plane,
40 and 41 are calculation results. It should be noted that the same or corresponding parts in the figures are indicated by the same reference numerals.

Claims (1)

[Claims] 1. An antenna in which a plurality of antenna elements are arranged vertically in one row at predetermined intervals, and a radio wave with a different frequency is provided corresponding to each of the antenna elements to a target from each antenna element. A plurality of transceivers extract only specific frequency components from the radio waves transmitted toward the target and reflected back from the target, and any one of the plurality of transceivers mentioned above. A ranging device that measures the slant range between the antenna and the target using the received signal, where i is a number that distinguishes the antenna elements, C is the speed of light, N is the number of antenna elements, and f _i (i=1, 2, 3,
..., N) is the transmission frequency, h _i (i=1, 2, 3, ...,
N) is the height at which the antenna elements are arranged, R _p is the slant range measured by the above distance measuring device, and θ _p
When is a signal processing variable and K is an arbitrary constant that is not zero, t _i = K・eXP [−j4πf _i /C{(h ₁ +h _N /2+ R _p tanθ _p −h _i ) ² + R _p ² }1/2] is _calculated for each antenna element, and the signal obtained from each antenna element is calculated as S _i
When (i = 1, 2, ..., N), calculate V = | _N 〓 ^{i = 1} S _i t _i |, change the variable θ _p in the above signal processing, and calculate the peak value of the above calculation result V. A radar device characterized in that it tracks a target using a signal processing variable θ _p that gives the peak value of the calculation result V as angle information of the target position.