JP7531810B2 - Chopper Circuit - Google Patents

Description

The present invention relates to a chopper circuit that converts voltage between a first DC voltage between first external connection terminals and a second DC voltage between second external connection terminals.

In recent years, the application of battery power storage systems to DC electric railways has progressed. Because the overhead line voltage of DC electric railways and the operating voltage of the energy storage elements are different magnitudes, voltage conversion (power conversion) must be performed using a bidirectional chopper circuit. In recent years, a bidirectional chopper circuit using an auxiliary power converter composed of chopper cells has been proposed.

For example, a chopper circuit that converts voltage between a first DC voltage at a first external connection terminal and a second DC voltage at a second external connection terminal includes a first switch unit having a first external connection terminal, a second switch unit that is connected in series to the first switch unit so that the conduction direction when the first switch unit is on is aligned with that of the first switch unit and has a second external connection terminal on the opposite side to the side to which the first switch unit is connected, one or a plurality of semiconductor power converters that are cascaded together and are provided on a wiring branched from a wiring that connects the first switch unit and the second switch unit, and a first external connection terminal that is connected to the second switch unit and a second external connection terminal that is connected to the first switch unit and a second external connection terminal that is connected to the second switch unit and a first ... second switch unit and a second external connection terminal that is connected to the first switch unit and a second external connection terminal that is connected to the first switch unit and a second external connection terminal that is connected to the second switch unit and a first external connection terminal that is connected to the second switch unit and a second external connection terminal that is connected to the first switch unit and a second external connection terminal that is connected to the second switch unit and a first external connection terminal that is connected to the first switch unit and a second external connection terminal that is connected to the second switch unit and a first A chopper circuit is known that includes an inductor connected in series to a semiconductor power converter, a semiconductor power converter control unit that controls the power conversion operation of the semiconductor power converter so as to output a current having a DC component and an AC component with a predetermined cycle, and a switch control unit that controls one of the first switch unit and the second switch unit to ON and the other to OFF, and that switches the first switch unit and the second switch unit from ON to OFF and from OFF to ON when the value of the current output by the semiconductor power converter is controlled to a predetermined value or less by the semiconductor power converter control unit (see, for example, Patent Document 1).

International Publication No. 2020/085172

The chopper circuit described in Patent Document 1 controls the auxiliary power converter to pass an inductor current having a DC component and a single fundamental AC component through the inductor, and achieves so-called "soft switching operation" in which the switch section performs switching operation when this current is zero, thereby improving converter efficiency and reducing the size and weight of the inductor. However, because the inductor current contains a single fundamental AC component, the DC currents flowing on the first DC voltage side and the second DC voltage side are not sufficiently smoothed and contain a large amount of noise. For this reason, it is necessary to take measures such as connecting a large and heavy filter circuit to the chopper circuit in order to smooth the DC current.

In addition, in a chopper circuit system in which three or more chopper circuits described in Patent Document 1 are connected in parallel, the inductor current can be controlled as a direct current by applying a non-interference current control method based on dq coordinate transformation, which is commonly used in the power system field, so that no phase delay occurs between the command value for the inductor current and the actual current. Therefore, the "timing at which the inductor current becomes zero" required for soft switching operation can be easily created. On the other hand, when the chopper circuit described in Patent Document 1 is used alone, the non-interference current control method based on dq coordinate transformation cannot be applied, so it is possible to apply a general PI control method instead of the non-interference current control method based on dq coordinate transformation. However, under the PI control method, a phase delay occurs between the command value for the inductor current and the actual current, so it becomes difficult to create the "timing at which the inductor current becomes zero" required for soft switching operation. Therefore, the chopper circuit described in Patent Document 1 lacks versatility because it is difficult to use it alone. In addition, the non-interfering current control method based on dq coordinate transformation, which is applied to a chopper circuit system in which three or more chopper circuits are connected in parallel as described in Patent Document 1, requires more complex calculation processing than the PI control method.

Therefore, there is a need for a chopper circuit that does not require a filter circuit for smoothing DC current, is small and lightweight, has easy calculation processing, and is highly versatile.

According to a first aspect of the present disclosure, a chopper circuit that converts voltage between a first DC voltage at a first external connection terminal and a second DC voltage at a second external connection terminal includes a first switch unit having a first external connection terminal, a second switch unit that is connected in series to the first switch unit so that the conduction direction when on is aligned with that of the first switch unit and has a second external connection terminal on the opposite side to the side to which the first switch unit is connected, one or a plurality of semiconductor power converters that are cascaded together and are provided on a wiring branched from a wiring that connects the first switch unit and the second switch unit, and an inverter that is connected in series to the semiconductor power converter on a wiring branched from a wiring that connects the first switch unit and the second switch unit. The semiconductor power converter control unit performs current control to output an inductor current that follows a current command value consisting of a DC component command value and an AC component command value from the semiconductor power converter, and a switch control unit that controls one of the first switch unit and the second switch unit to ON and the other to OFF, and executes switching from ON to OFF and from OFF to ON for the first switch unit and the second switch unit when the value of the inductor current output by the semiconductor power converter is controlled to a predetermined value or less by the semiconductor power converter control unit. The semiconductor power converter control unit performs lead compensation control to compensate for the phase lag of the inductor current with respect to the current command value in the current control.

According to a modified example of the first aspect of the present disclosure, a chopper circuit for converting a voltage between a first DC voltage at a first external connection terminal and a second DC voltage at a second external connection terminal includes a first switch unit having a first external connection terminal, a second switch unit connected in series to the first switch unit so that the conduction direction when on is aligned with that of the first switch unit and having a second external connection terminal on the opposite side to the side to which the first switch unit is connected, one or a plurality of semiconductor power converters connected in cascade to each other, which are provided on a wiring branched from the wiring connecting the first switch unit and the second switch unit, an inductor connected in series to the semiconductor power converter on a wiring branched from the wiring connecting the first switch unit and the second switch unit, and an inductor connected in series to the semiconductor power converter and connected to a DC component command value. The semiconductor power converter control unit performs current control to output an inductor current from the semiconductor power converter that follows a current command value consisting of a flow command value and a current distribution command value; and a switch control unit that controls one of the first switch unit and the second switch unit to ON and the other to OFF, and executes switching from ON to OFF and OFF to ON for the first switch unit and the second switch unit when the value of the inductor current output by the semiconductor power converter is controlled to a predetermined value or less by the semiconductor power converter control unit. Two first switch units and two second switch units are provided, and the semiconductor power converter control unit performs lead compensation control to compensate for the phase lag of the inductor current with respect to the current command value in the current control.

According to a second aspect of the present disclosure, a chopper circuit for voltage conversion between a first DC voltage between a pair of first external connection terminals and a second DC voltage between a pair of second external connection terminals includes a first switch unit, a second switch unit, a third switch unit, and a fourth switch unit that are connected in series to each other so that the conduction directions when turned on are aligned; one or a plurality of semiconductor power converters connected in cascade to each other, which are provided on a wiring that connects a connection point between the first switch unit and the second switch unit and a connection point between the third switch unit and the fourth switch unit; an inductor that is connected in series to the semiconductor power converter on a wiring that connects the connection point between the first switch unit and the second switch unit and the connection point between the third switch unit and the fourth switch unit; a semiconductor power converter control unit that performs current control to output an inductor current that follows a current command value consisting of a DC component command value and an AC component command value from the semiconductor power converter; and a set of the first switch unit and the third switch unit and a second switch unit. A switch control unit controls one of the pairs of the first switch unit and the fourth switch unit to ON and the other pair to OFF, and when the value of the inductor current output by the semiconductor power converter is controlled to a predetermined value or less by the semiconductor power converter control unit, the switch control unit executes switching from ON to OFF and switching from OFF to ON for each pair of the pair of the first switch unit and the third switch unit and the pair of the second switch unit and the fourth switch unit. A terminal on the opposite side to the connection side of the first switch unit and the second switch unit and a terminal on the opposite side to the connection side of the third switch unit and the fourth switch unit are a pair of first external connection terminals, and a terminal on the connection side of the second switch unit and the third switch unit and a terminal of the fourth switch unit on the opposite side to the connection side to the third switch unit are a pair of second external connection terminals, and the semiconductor power converter control unit performs lead compensation control to compensate for the phase lag of the inductor current with respect to the current command value in current control.

In addition, in the chopper circuits according to the first aspect, the modified example of the first aspect, and the second aspect, the AC component command value may have a lead compensation component for compensating for the phase lag of the inductor current relative to the current command value.

The lead compensation component may also be generated based on at least the proportional gain in the current control and the inductance of the inductor.

The AC command value may also have a fundamental wave AC component having a fundamental frequency and a tertiary wave AC component having a frequency three times the fundamental frequency.

The AC command value may also have a trapezoidal AC component.

The semiconductor power converter may also be composed of a chopper cell that includes two semiconductor switches connected in series and a DC capacitor connected in parallel to the two semiconductor switches, with each terminal of one of the two semiconductor switches serving as an output terminal.

Each semiconductor switch may also have a semiconductor switching element that passes current in one direction when on, and a feedback diode connected inversely parallel to the semiconductor switching element.

According to the first aspect, the modified example of the first aspect, and the second aspect of the present disclosure, it is possible to realize a chopper circuit that does not require a filter circuit for smoothing DC current, is small and lightweight, is easy to perform calculations, and is highly versatile.

FIG. 2 is a circuit diagram showing a chopper circuit according to the first embodiment of the present disclosure. FIG. 2 is a circuit diagram illustrating a semiconductor power converter in a chopper circuit according to the first and second embodiments of the present disclosure. FIG. 2 is a circuit diagram showing a chopper circuit according to a first modified example of the first embodiment of the present disclosure. FIG. 11 is a circuit diagram showing a chopper circuit according to a second modified example of the first embodiment of the present disclosure. 1 is a circuit diagram showing an example of the arrangement of semiconductor power converters and inductors in a chopper circuit according to the first and second embodiments of the present disclosure. FIG. FIG. 11 shows ideal waveforms of each part when current control is performed based on a current command value including an AC component command value according to the first form in the chopper circuit according to the first embodiment of the present disclosure, in which (A) shows the relationship between a triangular wave and a modulated wave used in the switch control part, (B) shows a voltage appearing across both ends of a first switch part, (C) shows a voltage appearing across both ends of a second switch part, (D) shows an output voltage of an auxiliary power converter, (E) shows an inductor current, and (F) shows a current flowing through the first switch part and the second switch part. FIG. 11 shows ideal waveforms of each part when current control is performed based on a current command value including an AC component command value according to a second form in the chopper circuit according to the first embodiment of the present disclosure, in which (A) shows the relationship between a triangular wave and a modulated wave used in the switch control part, (B) shows a voltage appearing across both ends of a first switch part, (C) shows a voltage appearing across both ends of a second switch part, (D) shows an output voltage of an auxiliary power converter, (E) shows an inductor current, and (F) shows a current flowing through the first switch part and the second switch part. 1A is a block diagram showing a DC capacitor voltage control system in a control unit for a semiconductor power converter in a chopper circuit according to a first embodiment of the present disclosure, in which (A) shows a DC voltage central control system, (B) shows an inductor current control system, and (C) shows an individual balance control system. 3 is a block diagram showing a voltage command value for each semiconductor power converter (chopper cell) in the chopper circuit according to the first embodiment of the present disclosure. FIG. FIG. 4 is a diagram showing circuit constants used in a simulation of the chopper circuit according to the first embodiment of the present disclosure. FIG. 11 shows simulation waveforms when 152 kW of power is transmitted from a first DC voltage side to a second DC voltage side when current control is performed based on a current command value including an AC component command value according to the first form in the chopper circuit according to the first embodiment of the present disclosure, in which (A) shows an inductor current, (B) shows a voltage appearing at both ends of a first switch unit and a second switch unit, (C) shows a current flowing in the first switch unit, (D) shows a current flowing in the second switch unit, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). FIG. 11 shows simulation waveforms when 152 kW of power is transmitted from the second DC voltage side to the first DC voltage side when current control is performed based on a current command value including an AC component command value according to the second form in the chopper circuit according to the first embodiment of the present disclosure, in which (A) shows an inductor current, (B) shows a voltage appearing at both ends of a first switch unit and a second switch unit, (C) shows a current flowing in the first switch unit, (D) shows a current flowing in the second switch unit, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). FIG. 11 shows simulation waveforms in a case where a direction of power transmission is reversed between a first DC voltage side and a second DC voltage side over 2.2 [ms] when current control is performed based on a current command value including an AC component command value according to a second form in the chopper circuit according to the first embodiment of the present disclosure, in which (A) shows an inductor current, (B) shows a voltage appearing at both ends of a first switch unit and a second switch unit, (C) shows a current flowing through the first switch unit, (D) shows a current flowing through the second switch unit, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). FIG. 4 is a circuit diagram showing a chopper circuit according to a second embodiment of the present disclosure. 11A and 11B are diagrams showing ideal waveforms of each part when current control is performed based on a current command value including an AC component command value according to the first form in a chopper circuit according to a second embodiment of the present disclosure, in which (A) shows the relationship between a triangular wave and a modulated wave used in a switch control part, (B) shows a voltage appearing across both ends of a first switch part and a voltage appearing across both ends of a second switch part, (C) shows an output voltage of an auxiliary power converter, (D) shows an inductor current, (E) shows a current flowing on the first DC voltage side, and (F) shows a current flowing on the second DC voltage side. 11A and 11B are diagrams showing ideal waveforms of each part when current control is performed based on a current command value including an AC component command value according to a second form in a chopper circuit according to a second embodiment of the present disclosure, in which (A) shows the relationship between a triangular wave and a modulated wave used in a switch control part, (B) shows a voltage appearing across both ends of a first switch part and a voltage appearing across both ends of a second switch part, (C) shows an output voltage of an auxiliary power converter, (D) shows an inductor current, (E) shows a current flowing on the first DC voltage side, and (F) shows a current flowing on the second DC voltage side. FIG. 11 is a diagram showing circuit constants used in a simulation of a chopper circuit according to a second embodiment of the present disclosure. FIG. 13 shows simulation waveforms when 200 kW of power is transmitted from the second DC voltage side to the first DC voltage side when current control is performed based on a current command value including an AC component command value according to the first form in the chopper circuit according to the second embodiment of the present disclosure, in which (A) shows an inductor current, (B) shows voltages appearing at both ends of the first switch unit and the second switch unit, (C) shows a current flowing on the first DC voltage side, (D) shows a current flowing on the second DC voltage side, and (E) shows a DC capacitor voltage of a capacitor in the semiconductor power converter (chopper cell). FIG. 13 shows simulated waveforms when 200 kW of power is transmitted from a first DC voltage side to a second DC voltage side when current control is performed based on a current command value including an AC component command value according to the second form in a chopper circuit according to a second embodiment of the present disclosure, in which (A) shows an inductor current, (B) shows a voltage appearing at both ends of a first switch unit and a second switch unit, (C) shows a current flowing on the first DC voltage side, (D) shows a current flowing on the second DC voltage side, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). FIG. 13 shows simulation waveforms in a case where a direction of power transmission is reversed between a first DC voltage side and a second DC voltage side over 2.2 [ms] when current control is performed based on a current command value including an AC component command value according to a second form in a chopper circuit according to a second embodiment of the present disclosure, in which (A) shows an inductor current, (B) shows voltages appearing at both ends of a first switch unit and a second switch unit, (C) shows a current flowing on the first DC voltage side, (D) shows a current flowing on the second DC voltage side, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell).

The chopper circuit will be described below with reference to the drawings. The scale of these drawings has been appropriately changed to facilitate understanding. The form shown in the drawings is one example for implementation, and is not limited to the illustrated embodiment.

Figure 1 is a circuit diagram showing a chopper circuit according to a first embodiment of the present disclosure. Figure 2 is a circuit diagram illustrating a semiconductor power converter in a chopper circuit according to the first and second embodiments of the present disclosure.

The chopper circuit 1 according to the first embodiment of the present disclosure performs bidirectional voltage conversion between a first DC voltage _vdc1 between a pair of first external connection terminals _T1 and _G1 and a second DC voltage _vdc2 between a pair of second external connection terminals _T2 and _G2 . A DC power supply is connected to one of the first external connection terminals _T1 and _G1 and the second external connection terminals _T2 and _G2 , and a load or another DC power supply is connected to the other.

For example, when a DC power supply is connected to the first external connection terminals _T1 and _G1 and a load is connected to the second external connection terminals _T2 and _G2 , the chopper circuit 1 operates as a step-down chopper. In this case, the voltage output by the DC power supply is the first DC voltage _vdc1 , and the voltage applied to the load is the second DC voltage _vdc2 .

For example, when a load is connected to the first external connection terminals _T1 and _G1 and a DC power supply is connected to the second external connection terminals _T2 and _G2 , the chopper circuit 1 operates as a boost chopper. In this case, the voltage applied to the load is the first DC voltage _vdc1 , and the voltage output by the DC power supply is the second DC voltage _vdc2 .

Also, for example, a DC power supply may be connected to the first external connection terminals _T1 and _G1 , and another DC power supply may be connected to the second external connection terminals _T2 and _G2 .

The chopper circuit 1 includes a first switch section 11, a second switch section 12, a semiconductor power converter 13, and an inductor 14. The chopper circuit 1 also includes a semiconductor power converter control section 15 and a switch control section 16 as its control system.

The first switch unit 11 is a semiconductor valve device capable of unidirectional current interruption. The first switch unit 11 is composed of a semiconductor switching element that conducts in one direction when on and a feedback diode connected in anti-parallel to the semiconductor switching element. Examples of semiconductor switching elements include IGBTs, SiC (Silicon Carbide)-MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors), thyristors, GTOs (Gate Turn-OFF Thyristors), and transistors, but the type of semiconductor switching element itself does not limit the present invention and may be other semiconductor elements.

The first switch section 11 has a first external connection terminal _T1 . The connection point between the first switch section 11 and the second switch section 12 is represented as _P1 . That is, the connection point _P1 is located on the side of the first switch section 11 opposite to the side where the first external connection terminal _T1 is provided. Here, the forward voltage of the first switch section 11 (i.e., the potential difference between the first external connection terminal _T1 and the connection point _P1 ) is represented as _vS1 .

The second switch section 12 is a semiconductor valve device capable of unidirectional current interruption. The second switch section 12 is composed of a semiconductor switching element that conducts in one direction when on and a feedback diode connected in anti-parallel to the semiconductor switching element. Examples of semiconductor switching elements include IGBTs, SiC (Silicon Carbide)-MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors), thyristors, GTOs (Gate Turn-OFF Thyristors), and transistors, but the type of semiconductor switching element itself does not limit the present invention and may be other semiconductor elements.

The second switch section 12 is connected in series to the first switch section 11 at a connection point _P1 so that the conduction direction when on is the same as that of the first switch section 11. The second switch section 12 has a second external connection terminal _T2 on the side opposite to the side (connection point _P1 ) to which the first switch section 11 is connected. Here, the forward voltage of the second switch section 12 (i.e., the potential difference between the connection point _P1 and the first external connection terminal _T2 ) is represented as _vS2 .

In this specification, a set of power converters consisting of a first switch unit 11 and a second switch unit 12 is referred to as a main power converter 10. As described below, while one of the first switch unit 11 and the second switch unit 12 is controlled to be on, the other is controlled to be off.

A variable control voltage source using a semiconductor power converter 13 and an inductor 14 are provided on a wiring branched from a connection point P ₁ between the first switch section 11 and the second switch section 12 .

The semiconductor power converter 13 is provided on a wiring branched from a connection point _P1 between the first switch section 11 and the second switch section 12, either singly or in a state in which a plurality of semiconductor power converters 13 are cascaded to each other. In this specification, the single semiconductor power converter 13 or a plurality of semiconductor power converters 13 is referred to as an auxiliary power converter 19. In this specification, when there is one semiconductor power converter 13, the side to which an inductor 14 described later is connected is referred to as a "first DC side", and when a plurality of semiconductor power converters 13 are cascaded to each other, the side to which another semiconductor power converter 13 different from the semiconductor power converter 13 is connected is also referred to as a "first DC side". In addition, the DC side opposite to the "first DC side" is referred to as a "second DC side". As an example, FIG. 1 shows a case in which a plurality of semiconductor power converters 13 (N, where N is an integer of 2 or more) are cascaded to each other on the first DC side. Hereinafter, the number of cascades of semiconductor power converters 13 is represented by j (where j is a natural number from 1 to N). By simply adjusting the number of cascade-connected semiconductor power converters 13 as appropriate, the chopper circuit 1 can easily be made to withstand high voltage.

The semiconductor power converter 13 is configured as a bidirectional chopper cell having a DC-DC converter 131 and a capacitor 132. That is, the semiconductor power converter 13 is composed of a chopper cell including two semiconductor switches connected in series and a DC capacitor connected in parallel to the two semiconductor switches, with each terminal of one of the two semiconductor switches serving as an output end. That is, the DC-DC converter 131 is composed of two semiconductor switching elements S connected in series to each other, and a feedback diode D connected in anti-parallel to each of the semiconductor switching elements S. Examples of the semiconductor switching element S include an IGBT, a SiC (Silicon Carbide)-MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), a thyristor, a GTO (Gate Turn-OFF Thyristor), a transistor, etc., but the type of the semiconductor switching element itself does not limit the present invention, and other semiconductor elements may be used. The capacitor 132 is provided on the second DC side of the semiconductor power converter 13. When the chopper circuit 1 is operated, the DCDC converter 131 is operated to initially charge the capacitor 132. Here, the voltage of the DC capacitor of each semiconductor power converter 13 is v _Cj , and the voltage on the first DC side of the auxiliary power converter 19 is v. Although details will be described later, by controlling the inductor current i using the auxiliary power converter 19, the inductor 14 and the auxiliary power converter 19 operate as a controlled current source. Note that, although the auxiliary power converter is realized by cascading a plurality of semiconductor power converters 13 (chopper cells) in Fig. 1, any semiconductor power converter having a similar function can be used instead.

The inductor 14 is connected in series to the semiconductor power converter 13 on a wiring branched from a connection point _P1 on the wiring connecting the first switch section 11 and the second switch section 12. Also, the inductor current flowing through the inductor 14 provided between the connection points _P1 and _P2 is denoted as i.

Therefore, the semiconductor power converter 13 and the inductor 14 are provided on the same wiring branched from a connection point P1 on the wiring connecting the first switch section 11 and the second switch section 12 in the main power converter 10. In the example shown in Fig. 1, the inductor ₁₄ is arranged between the connection point _P1 and the semiconductor power converter 13, and the second external connection terminals _T2 and _G2 are arranged at a connection point _P2 on the side of the auxiliary power converter 19 opposite to the side to which the inductor 14 is connected (i.e., on the side opposite to the side to which the inductor 14 is connected of the set of multiple semiconductor power converters 13), but the arrangement order of these semiconductor power converters 13 and inductors 14 can be designed arbitrarily on the wiring between the connection point _P1 and the connection point _P2 . Other examples of the arrangement of the semiconductor power converters and inductors in the chopper circuit will be described later.

The semiconductor power converter control unit 15 controls the power conversion operation of the semiconductor power converter 13 so as to output a current having a DC component and an AC component. In particular, the semiconductor power converter control unit 15 performs current control to cause the semiconductor power converter 13 to output an inductor current that follows a current command value consisting of a DC component command value and an AC component command value. In this current control, the semiconductor power converter control unit 15 performs lead compensation control to compensate for the phase delay of the inductor current relative to the current command value. For this reason, the AC component command value of the current command value has a lead compensation component to compensate for the phase delay of the inductor current relative to the current command value. This lead compensation component is generated based on at least the proportional gain in the current control and the inductance of the inductor 14. Details of the lead compensation control in the current control by the semiconductor power converter control unit 15 will be described later.

The switch control unit 16 controls either the first switch unit 11 or the second switch unit 12 to ON and controls the other to OFF. In addition, when the semiconductor power converter control unit 15 controls the value of the current output by the semiconductor power converter 13 to a predetermined value or less, the switch control unit 16 executes switching from ON to OFF and from OFF to ON for the first switch unit 11 and the second switch unit 12. Here, the predetermined value is a value sufficiently smaller than the rated current of the semiconductor power converter 13. As an example, the predetermined value is a value of, for example, about 0% to 10% of the rated current of the semiconductor power converter 13, but depending on the application environment of the chopper circuit 1, it may be a value exceeding 10% of the rated current of the semiconductor power converter 13.

The arithmetic processing device (not shown) has a semiconductor power converter control unit 15 and a switch control unit 16. The semiconductor power converter control unit 15 and the switch control unit 16 are functional modules realized by, for example, a computer program executed on the arithmetic processing device. For example, when the semiconductor power converter control unit 15 and the switch control unit 16 are constructed in the form of a computer program, the functions of the semiconductor power converter control unit 15 and the switch control unit 16 can be realized by operating the arithmetic processing device according to the computer program. The computer program for executing the processing of the semiconductor power converter control unit 15 and the switch control unit 16 may be provided in a form recorded on a computer-readable recording medium such as a semiconductor memory, a magnetic recording medium, or an optical recording medium. Alternatively, the semiconductor power converter control unit 15 and the switch control unit 16 may be realized as a semiconductor integrated circuit in which a computer program that realizes the functions of each unit is written. The operation of the semiconductor power converter control unit 15 and the switch control unit 16 will be described in detail later.

The chopper circuit 1 according to the first embodiment can perform bidirectional voltage conversion between a first DC voltage _vdc1 between a pair of first external connection terminals _T1 and _G1 and a second DC voltage _vdc2 between a pair of second external connection terminals _T2 and _G2 . However, in the chopper circuit 1 having two semiconductor valve devices, i.e., a first switch section 11 and a second switch section 12, as the main power converter 10 as shown in Fig. 1, it is necessary to have a relationship " _vdc1 > _vdc2 " in which the first DC voltage _vdc1 is greater than the second DC voltage _vdc2 . While one of the first switch section 11 and the second switch section 12 is controlled to be on, the other is controlled to be off.

In order to perform bidirectional voltage conversion between the first DC voltage _vdc1 and the second DC voltage _vdc2 regardless of the magnitude relationship between the first DC voltage _vdc1 and the second DC voltage _vdc2 , it is sufficient to provide two semiconductor valve devices capable of bidirectional current interruption or four semiconductor valve devices capable of unidirectional current interruption as the main power converter.

For example, when two semiconductor valve devices capable of bidirectional current interruption are provided as the main power converter to perform bidirectional voltage conversion between the first DC voltage _vdc1 and the second DC voltage _vdc2 regardless of the magnitude relationship between the first DC voltage _vdc1 and the second DC voltage _vdc2 , the first switch section 11 and the second switch section 12 in the chopper circuit 1 shown in FIG. 1 can be configured with semiconductor valve devices capable of bidirectional current interruption.

Furthermore, for example, in order to perform bidirectional voltage conversion between the first DC voltage _vdc1 and the second DC voltage _vdc2 regardless of the magnitude relationship between the first DC voltage _vdc1 and the second DC voltage _vdc2 , in the case of providing four semiconductor valve devices capable of unidirectional current interruption, the semiconductor valve devices may be arranged as shown in FIGS. 3 and 4.

Figure 3 is a circuit diagram showing a chopper circuit according to a first modified example of the first embodiment of the present disclosure.

The chopper circuit 1 according to the first modification of the first embodiment has four semiconductor valve devices as the main power converter 10, namely, first switch units 11-1 and 11-2 and second switch units 12-1 and 12-2. The first switch units 11-1 and 11-2 and the second switch units 12-1 and 12-2 are all semiconductor valve devices capable of unidirectional current interruption, and are composed of a semiconductor switching element that conducts in one direction when turned on and a feedback diode connected in reverse parallel to the semiconductor switching element. The first switch unit 11-1 is provided between the first external connection terminal T ₁ and the connection point P _1. The first switch unit 11-2 is provided between the ground terminal G ₁ of the first external connection terminal and the connection point P _2. The second switch unit 12-1 is provided between the connection point P ₁ and the second external connection terminal T ₂ . The second switch section 12-2 is provided between the connection point _P2 and the ground terminal _G2 of the second external connection terminal. The second switch section 12-1 is connected in series to the first switch section 11-1 at the connection point _P1 so that the conduction direction when on is aligned with that of the first switch section 11-1. The second switch section 12-2 is connected in series to the first switch section 11-2 at the connection point _P2 so that the conduction direction when on is aligned with that of the first switch section 11-2. In order to perform bidirectional voltage conversion between the first DC voltage _vdc1 and the second DC voltage _vdc2 under the relationship " _vdc1 > _vdc2 " in which the first DC voltage _vdc1 is greater than the second DC voltage _vdc2 , the first switch section 11-2 and the second switch section 12-2 are always on, and while one of the first switch section 11-1 and the second switch section 12-1 is controlled to be on, the other is controlled to be off. In this case, the chopper circuit 1 in FIG. 3 is equivalent to the chopper circuit in FIG. 1. On the other hand, in order to perform bidirectional voltage conversion between the first DC voltage _vdc1 and the second DC voltage _vdc2 under the relationship " _vdc1 < _vdc2 " in which the first DC voltage _vdc1 is smaller than the second DC voltage _vdc2 , the first switch section 11-1 and the second switch section 12-1 are always kept on, and while either the first switch section 11-2 or the second switch section 12-2 is controlled to be on, the other is controlled to be off.

Figure 4 is a circuit diagram showing a chopper circuit according to a second modified example of the first embodiment of the present disclosure.

The chopper circuit 1 according to the second modification of the first embodiment has four semiconductor valve devices, namely, first switch units 11-1 and 11-2 and second switch units 12-1 and 12-2, as the main power converter 10. The first switch units 11-1 and 11-2 and the second switch units 12-1 and 12-2 are all semiconductor valve devices capable of unidirectional current interruption, and are composed of a semiconductor switching element that conducts in one direction when on and a feedback diode connected in anti-parallel to the semiconductor switching element. The first switch unit 11-1 and the first switch unit 11-2 are provided between the first external connection terminal T ₁ and the connection point P _1. The first switch unit 11-1 is provided so that the conduction direction when on is opposite to that of the first switch unit 11-2. The first switch unit 11-1 and the first switch unit 11-2 may be provided interchangeably. The second switch section 12-1 and the second switch section 12-2 are provided between the connection point _P1 and the second external connection terminal _T2 . The second switch section 12-1 is provided so that the conduction direction when on is opposite to that of the second switch section 12-2. The second switch section 12-1 and the second switch section 12-2 may be provided in an interchangeable manner. The second switch section 12-1 is provided so that the conduction direction when on is the same as that of the first switch section 11-1. The second switch section 12-2 is provided so that the conduction direction when on is the same as that of the first switch section 11-2. In order to perform bidirectional voltage conversion between the first DC voltage _vdc1 and the second DC voltage _vdc2 under the relationship " _vdc1 > _vdc2 " in which the first DC voltage _vdc1 is greater than the second DC voltage _vdc2 , the first switch section 11-2 and the second switch section 12-2 are always kept on, and while either the first switch section 11-1 or the second switch section 12-1 is controlled to be on, the other is controlled to be off. On the other hand, in order to perform bidirectional voltage conversion between the first DC voltage _vdc1 and the second DC voltage _vdc2 under the relationship " _vdc1 < _vdc2 " in which the first DC voltage _vdc1 is smaller than the second DC voltage _vdc2 , the first switch section 11-1 and the second switch section 12-1 are always kept on, and while either the first switch section 11-2 or the second switch section 12-2 is controlled to be on, the other is controlled to be off.

In addition, an overvoltage suppression snubber circuit may be connected in parallel to each semiconductor valve device of the first switch section 11 and the second switch section 12 in the chopper circuit 1 according to the first embodiment shown in FIG. 1, the first switch section 11-1 and 11-2 and the second switch section 12-1 and 12-2 in the chopper circuit 1 according to the first modified example of the first embodiment shown in FIG. 3 and the second modified example of the first embodiment shown in FIG. 4.

As described above, the semiconductor power converter 13 and the inductor 14 are provided on the same wiring branched from the connection point _P1 on the wiring connecting the first switch section 11 and the second switch section 12 in the main power converter 10. In the examples shown in Figs. 1, 3 and 4, the inductor 14 is arranged between the connection point _P1 and the semiconductor power converter 13, and the second external connection terminals _T2 and _G2 are arranged at the connection point _P2 of the auxiliary power converter 19 on the side opposite to the side to which the inductor 14 is connected (i.e., on the side opposite to the side to which the inductor 14 is connected of the set of the multiple semiconductor power converters 13), but the arrangement order of these semiconductor power converters 13 and inductors 14 can be designed arbitrarily on the wiring between the connection point _P1 and the connection point _P2 . Fig. 5 is a circuit diagram showing an example of the arrangement of the semiconductor power converters and inductors in the chopper circuits according to the first and second embodiments of the present disclosure. The layout example of the semiconductor power converter and the inductor shown in FIG. 5 can be applied to the chopper circuit according to the first embodiment shown in FIG. 1, FIG. 3, and FIG. 4, and the chopper circuit according to the second embodiment shown in FIG. 14 described later. However, when the layout example of the semiconductor power converter and the inductor shown in FIG. 5 is applied to the chopper circuit according to the second embodiment, the connection points P ₁ and P ₂ shown in FIG. 5 are replaced with connection points P ₃ and P ₅ , respectively. For the semiconductor power converter 13 in FIG. 5, in order to facilitate understanding, the capacitor C (capacitor 132) in the semiconductor power converter 13 shown in FIG. 1 to FIG. 4 and FIG. 14 is written outside the semiconductor power converter 13. Also, each of the N semiconductor power converters 13 (where N is a natural number) is represented by cell 1, ..., cell j, ..., cell N. In the example shown in FIG. 5(A), the inductor 14 is placed between the connection point P ₁ and cell 1 of the semiconductor power converter 13. In the example shown in Fig. 5(B), the inductor 14 is disposed between the cell N of the semiconductor power converter 13 and the connection point _P2 . In the example shown in Fig. 5(C), the inductor 14 is disposed on the opposite side of the cell N of the semiconductor power converter 13 to the connection point _P2 .

Next, we will explain the operating principle of the chopper circuit 1 according to the first embodiment of the present disclosure.

The operation of the chopper circuit 1 according to the first modified example of the first embodiment shown in FIG. 3 and the operation of the chopper circuit 1 according to the second modified example of the first embodiment shown in FIG. 4 can be considered to be similar to the operation of the chopper circuit 1 according to the first embodiment shown in FIG. 1, so here, the operation of the chopper circuit 1 according to the first embodiment shown in FIG. 1 will be described. Below, a continuous control system with zero control delay is assumed.

The semiconductor power converter control unit 15 controls the power conversion operation of the semiconductor power converter 13 so as to output a current having a DC component and an AC component. In particular, the semiconductor power converter control unit 15 performs current control to cause the semiconductor power converter 13 to output an inductor current that follows a current command value consisting of a DC component command value and an AC component command value. In this current control, the semiconductor power converter control unit 15 performs lead compensation control to compensate for the phase lag of the inductor current relative to the current command value. For this reason, the AC component command value of the current command value has a lead compensation component to compensate for the phase lag of the inductor current relative to the current command value. This lead compensation component is generated based on at least the proportional gain in the current control and the inductance of the inductor 14.

In _the chopper circuit 1 according to the first embodiment shown in FIG.

At this time, in the chopper circuit 1 according to the first embodiment shown in FIG. 1, the circuit equation shown in Equation 2 holds. In Equation 2, L represents the inductance of the inductor 14, R represents the resistance of the chopper circuit 1, i represents the inductor current flowing through the inductor 14, and v represents the voltage on the first DC side of the auxiliary power converter 19 (i.e., the sum of the voltages on the first DC sides of the semiconductor power converters 13).

On the other hand, the voltage v on the first DC side of the auxiliary power converter 19 is given by Equation 3. In Equation 3, K _P represents a proportional gain in the current control of the semiconductor power converter control unit 15, _KI represents an integral gain in the current control of the semiconductor power converter control unit 15, and i ^* represents a current command value in the current control of the semiconductor power converter control unit 15.

Substituting equation 3 into equation 2 gives equation 4.

By performing a Laplace transform on both sides of equation 4, equation 5 is obtained. In equation 5, I(s) represents the Laplace transform of the inductor current i, and I ^* (s) represents the Laplace transform of the current command value i ^* .

Here, the integral gain K _I is defined as in Equation 6.

Substituting equation 6 into equation 5 gives equation 7, which is the response of a first-order system.

As described later, the current command value i ^* in the current control of the semiconductor power converter control unit 15 is composed of a DC component command value i _dc ^* and an AC component command value i _ac ^* . As can be seen from Equation 7, the DC component included in the inductor current i follows the DC component command value i _dc ^* without any steady-state deviation. On the other hand, the AC component included in the inductor current i has an amplitude reduction and a phase delay with respect to the AC component command value i _ac ^* . Therefore, in the embodiment of the present disclosure, the semiconductor power converter control unit 15 performs lead compensation control in the current control to compensate for the phase delay of the inductor current i with respect to the current command value i ^* . The current command value i ^* is specified as shown in Equation 8.

In Equation 8, the second term on the right side L/ _KP ·d( _iac ^* )/dt corresponds to a lead compensation component, and this component is included in the AC command value. The lead compensation component L/ _KP ·d( _iac ^* )/dt includes at least a proportional gain _KP in the current control and an inductance L of the inductor 14.

From Equations 7 and 8, in the steady state, the relationship expressed by Equation 9 can be realized between the inductor current i, the DC component command value i _dc ^* , and the AC component command value i _ac ^* .

Next, the forms of the AC component command values included in the current command value will be listed. The operation of the chopper circuit 1 according to the first modified example of the first embodiment shown in FIG. 3 and the operation of the chopper circuit 1 according to the second modified example of the first embodiment shown in FIG. 4 can be considered to be similar to the operation of the chopper circuit 1 according to the first embodiment shown in FIG. 1, so here, the forms of the AC component command values included in the current command value in the chopper circuit 1 according to the first embodiment shown in FIG. 1 will be listed.

First, the AC component command value according to the first form in the chopper circuit 1 according to the first embodiment will be described. The AC component command value according to the first form has a fundamental wave AC component having a fundamental frequency and a tertiary wave AC component having a frequency three times the fundamental frequency.

6 is a diagram showing ideal waveforms of each part when current control is performed based on a current command value including an AC component command value according to the first embodiment of the present disclosure in the chopper circuit, where (A) shows the relationship between a triangular wave and a modulated wave used in the switch control part, (B) shows the voltage appearing at both ends of the first switch part, (C) shows the voltage appearing at both ends of the second switch part, (D) shows the output voltage of the auxiliary power converter, (E) shows the inductor current, and (F) shows the current flowing through the first switch part and the second switch part. In FIG. 6(A), the triangular wave v _tri is shown by a solid line, and the modulated wave d is shown by a dashed line. In FIG. 6(F), the current i ₁ flowing through the first switch part 11 is shown by a solid line, and the current i ₂ flowing through the second switch part 12 is shown by a dashed line.

The on/off timing of the first switch unit 11 and the second switch unit 12 is determined by the switch control unit 16 based on a comparison result between the triangular wave v _tri , which has a minimum value of 0 and a maximum value of 1, and the modulation wave d. The method of determining the modulation wave d in the AC component command value according to the first embodiment will be described in detail later, but it is determined based on the relationship between the first DC voltage v _dc1 and the second DC voltage v _dc2 . The switch control unit 16 controls one of the first switch unit 11 and the second switch unit 12 to be on and the other to be off. For example, as shown in FIG. 6A, when the triangular wave v _tri is smaller than the modulation wave d, the switch control unit 16 controls the first switch unit 11 to be on and the second switch unit 12 to be off. In this case, the voltage appearing across the first switch section 11 becomes v _S1 =0 as shown in Fig. 6(B), and the voltage appearing across the second switch section 12 becomes v _S2 =v _dc1 -v _dc2 as shown in Fig. 6(C). For example, when the triangular wave v _tri is larger than the modulation wave d as shown in Fig. 6(A), the switch control section 16 controls the first switch section 11 to be turned off and the second switch section 12 to be turned on. In this case, the voltage appearing across the first switch section 11 becomes v _S1 =v _dc1 -v _dc2 as shown in Fig. 6(B), and the voltage appearing across the second switch section 12 becomes v _S2 =0 as shown in Fig. 6(C).

In the first form, the inductor current i includes a DC component, a fundamental AC component having a fundamental frequency, and a tertiary AC component having a frequency three times the fundamental frequency. If the carrier frequency of the triangular wave is _fSM , the inductor current i is expressed as in Equation 10. In Equation 10, _Iac represents the amplitude of the AC component ( _Iac >0), and _Idc represents the amplitude of the DC component ( _Idc >0). In Equation 10, the amplitude of the tertiary AC component is set to, for example, 0.4 times the amplitude of the fundamental AC component as an example, but the value of the amplitude of the tertiary AC component can be set to any value.

The reason why the inductor current i includes the DC component _Idc and the AC component _Iac will be explained as follows. As shown in Fig. 9, which will be described later, the output voltage v of the auxiliary power converter 19 includes a voltage term _vf that outputs _vdc1 when the first switch unit 11 is on and outputs _vdc2 when the second switch unit 12 is on as shown in Equation 11 as feedforward control (Fig. 6(D)). The voltage term _vf is larger than the other voltage terms included in the output voltage v of the auxiliary power converter 19, and therefore the output voltage v of the auxiliary power converter 19 can be considered to be approximately equal to the voltage term _vf (v ≈ _vf ).

From equation 11 and FIG. 6D, the voltage term _vf equivalently includes a DC voltage, an AC voltage having the same frequency as the carrier frequency _fSM , and an AC voltage having the same frequency as _3fSM .

On the other hand, in order to keep the DC capacitor voltage of the capacitor C in each semiconductor power converter 13 constant in the auxiliary power converter 19, it is necessary to control the auxiliary power converter 19 so that the power " _vf x i" of the auxiliary power converter 19 is zero on average. To achieve this, the semiconductor power converter control unit 15 controls the power conversion operation of the semiconductor power converter 13 (auxiliary power converter 19) so that a current in which a DC component _Idc is superimposed on the AC component _Iac flows as the inductor current i to the inductor 14. As shown in Figures 6(D) and 6(E), when the direction of the power flow is from the first DC voltage _vdc1 side to the second DC voltage _vdc2 side, the AC components included in the voltage term _vf and the inductor current i are both in phase with each other, that is, the fundamental wave AC component and the tertiary wave AC component, and therefore both form a positive active power " _vf x i". Therefore, if the DC components contained in the voltage term _vf and the inductor current i form negative active power, the power " _vf ×i" of the auxiliary power converter 19 can be set to zero on average. Since the DC component contained in the voltage term _vf is positive according to Fig. 6(D) and equation 11, the polarity of the DC component contained in the inductor current i is negative.

On the other hand, when the direction of the power flow is from the second DC voltage _vdc2 side to the first DC voltage _vdc1 side, the phase of the AC component included in the inductor current i is shifted by 180 degrees from the case of the power flow from the first DC voltage _vdc1 side to the second DC voltage _vdc2 side. That is, since the AC components included in the voltage term _vf and the inductor current i are in opposite phase, they form a negative active power " _vf x i". Therefore, if the DC components included in the voltage term _vf and the inductor current i form a positive active power, the power " _vf x i" of the auxiliary power converter 19 can be made zero on average. Since the DC component included in the voltage term _vf is negative according to FIG. 6(D) and Equation 11, the polarity of the DC component included in the inductor current i becomes positive. Therefore, when the direction of power flow is from the second DC voltage _vdc2 side to the first DC voltage _vdc1 side, the inductor current i is expressed as in Equation 12.

In Fig. 6, the phase at the moment when the inductor current i switches from negative to positive is α (where 0<α<π/2), and the phase at the moment when the inductor current i switches from positive to negative is π-α. In one period (0≦θ≦2π), when "α≦θ≦π-α", the first switch unit 11 is on and the second switch unit 12 is off, so that the current flowing through the first switch unit 11 is _i = i, and the current flowing through the second switch unit 12 is _i = 0, as shown in Fig. 6(F). On the other hand, when "0≦θ≦α" and "π-α≦θ≦2π", the first switch unit 11 is off and the second switch unit 12 is on, so that the current flowing through the first switch unit 11 is _i = 0, and the current flowing through the second switch unit 12 is _i = -i, as shown in Fig. 6(F). Moreover, from FIG. 6E and equation 11, the DC component I _dc of the inductor current i can be expressed as equation 13 using the AC components I _ac and α.

In addition, with regard to the dead time period in which both the first switch unit 11 and the second switch unit 12 are off, when the inductor current i is positive (i>0), the current flows through the feedback diode of the second switch unit 12, so the current flowing through the first switch unit 11 is _i1 = 0, and the current flowing through the second switch unit 12 is _i2 = -i. On the other hand, when the inductor current i is negative (i<0), the current flows through the feedback diode of the first switch unit 11, so the current flowing through the first switch unit 11 is _i1 = i, and the current flowing through the second switch unit 12 is _i2 = 0. Furthermore, according to Equations 10 and 12, Equation 13 always holds true even if the direction of the power flow changes.

On the other hand, when focusing on the turn-on and turn-off of the first switch section 11 and the second switch section 12, the current _i1 flowing through the first switch section 11 and the current _i2 flowing through the second switch section 12 during a voltage change are both zero. This means that the first switch section 11 and the second switch section 12 in an ideal state do not generate switching loss.

This is the reason why the inductor current i contains a DC component I _dc and an AC component I _ac .

Next, a method for determining the modulated wave d and the phase α in the first form in which the AC command value includes the fundamental wave AC component and the tertiary wave AC component will be described. In the following, an ideal state in which the loss of each converter is zero is assumed. If the average power on the first DC voltage _vdc1 side is _Pdc1 and the average power on the second DC voltage _vdc2 side is _Pdc2 , then Equation 14 holds steadily between the two.

The average power _Pdc1 on the first DC voltage _vdc1 side and the average power _Pdc2 on the second DC voltage _vdc2 side can be expressed as in Equation 15 and Equation 16, where _I1 and _I2 are the average values over one cycle of the current _i1 flowing through the first switch section 11 and the current _i2 flowing through the second switch section 12, respectively.

From FIG. 6(F) and equations 10 and 13, the current i ₁ flowing through the first switch portion 11 can be expressed by equation 17.

Similarly, from FIG. 6(F) and equations 12 and 13, the current i ₂ flowing through the second switch section 12 can be expressed by equation 18.

From equations 14 to 18, equation 19 holds for phase α.

It can be seen from Equation 19 that the phase α is determined by the first DC voltage _vdc1 and the second DC voltage _vdc2 , and does not depend on the AC component _Iac of the inductor current i. This means that the phase α is a value that does not depend on the amount of transmitted power.

To simplify equation 19, we apply the approximation shown in equation 20.

As a result, equation 19 can be transformed into equation 21.

Equation 11 is a quadratic equation with respect to phase α, and solving it for phase α gives us equation 22.

The phase condition "0 < α < π/2" means that the phase α can be expressed as in Equation 23.

6A, the slope of the triangular wave v _tri between phase π/2 and phase 3π/2 is "1/π", and the value of the triangular wave v _tri at phase π is "0.5". The value of the triangular wave v _tri at phase π-α is "d". From these relationships, d can be expressed as in Equation 24.

As described above, the modulation wave d is determined based on Equation 24, and the phase α is determined based on Equation 23. When the triangular wave v _tri is smaller than the modulation wave d determined based on Equation 24, the switch control unit 16 controls the first switch unit 11 to be ON and the second switch unit 12 to be OFF. When the triangular wave v _tri is larger than the modulation wave d determined based on Equation 24, the switch control unit 16 controls the first switch unit 11 to be OFF and the second switch unit 12 to be ON. The switch control unit 16 executes switching from ON to OFF and switching from OFF to ON for the first switch unit 11 and the second switch unit 12 at the phase α and phase π-α at which the current output by the semiconductor power converter 13 is controlled to zero by the semiconductor power converter control unit 15.

Next, we will explain the AC component command value in the second form in the chopper circuit 1 according to the first embodiment. The AC component command value in the second form has a trapezoidal wave AC component.

7 is a diagram showing ideal waveforms of each part when current control is performed based on a current command value including an AC component command value according to the second form in the chopper circuit according to the first embodiment of the present disclosure, where (A) shows the relationship between a triangular wave and a modulated wave used in the switch control part, (B) shows the voltage appearing at both ends of the first switch part, (C) shows the voltage appearing at both ends of the second switch part, (D) shows the output voltage of the auxiliary power converter, (E) shows the inductor current, and (F) shows the current flowing through the first switch part and the second switch part. In FIG. 6(A), the triangular wave v _tri is shown by a solid line, and the modulated wave d is shown by a dashed line. In FIG. 6(F), the current i ₁ flowing through the first switch part is shown by a solid line, and the current i ₂ flowing through the second switch part is shown by a dashed line.

The on/off timing of the first switch unit 11 and the second switch unit 12 is determined by the switch control unit 16 based on a comparison result between the triangular wave v _tri , which has a minimum value of 0 and a maximum value of 1, and the modulation wave d. The method of determining the modulation wave d in the AC component command value according to the second embodiment will be described in detail later, but it is determined based on the relationship between the first DC voltage v _dc1 and the second DC voltage v _dc2 . The switch control unit 16 controls one of the first switch unit 11 and the second switch unit 12 to be on and the other to be off. For example, as shown in FIG. 7A, when the triangular wave v _tri is smaller than the modulation wave d, the switch control unit 16 controls the first switch unit 11 to be on and the second switch unit 12 to be off. In this case, the voltage appearing across the first switch section 11 becomes v _S1 =0 as shown in Fig. 7(B), and the voltage appearing across the second switch section 12 becomes v _S2 =v _dc1 -v _dc2 as shown in Fig. 6(C). For example, when the triangular wave v _tri is larger than the modulation wave d as shown in Fig. 7(A), the switch control section 16 controls the first switch section 11 to be turned off and the second switch section 12 to be turned on. In this case, the voltage appearing across the first switch section 11 becomes v _S1 =v _dc1 -v _dc2 as shown in Fig. 7(B), and the voltage appearing across the second switch section 12 becomes v _S2 =0 as shown in Fig. 7(C).

Here, we explain how to determine the modulation wave d and phase α in the second form in which the AC command value includes a trapezoidal AC component. In the following, we assume an ideal state in which the loss in each converter is zero.

The inductor current i, which has a trapezoidal wave AC component, is expressed as in Equation 25, where 0<φ<π/2.

Let α be the phase angle where i = 0 when 0 < θ < φ. From Equation 25, α is given by Equation 26.

From Equations 25 and 26, the average value _I1 of the current flowing through the first switch section 11 in one period can be expressed by Equation 27.

Similarly, from equations 25 and 26, the average value I ₂ of the current flowing through the second switch section 12 in one period can be expressed by equation 28.

From equations 14 to 16, 27, and 28, equation 29 holds for phase α.

Transforming equation 29 gives us equation 30.

Equation 30 is a quadratic equation with respect to phase α, and solving it for phase α (where 0<α<π/2) gives Equation 31.

7A, the slope of the triangular wave v _tri between phase π/2 and phase 3π/2 is "1/π", and the value of the triangular wave v _tri at phase π is "0.5". The value of the triangular wave v _tri at phase π-α is "d". From these relationships, d in the second form can be expressed as in Equation 24, similar to the first form.

In the above-mentioned method of determining the phase α in the first and second forms of the AC component command value, an ideal state in which the loss of each converter is zero is assumed. Under this ideal state, when the phase of the current output by the semiconductor power converter 13 by the semiconductor power converter control unit 15 is α and π-α, the current output by the semiconductor power converter 13 becomes zero. However, in the actual chopper circuit 1, since there is a loss in each converter in the chopper circuit 1, even when the phase of the current output by the semiconductor power converter 13 by the semiconductor power converter control unit 15 is α and π-α, the current output by the semiconductor power converter 13 does not become zero, and a small current flows. Therefore, the switch control unit 16 executes the switching from on to off and the switching from off to on for the first switch unit 11 and the second switch unit 12 when the current output by the semiconductor power converter 13 by the semiconductor power converter control unit 15 is controlled to a predetermined value or less at phase α and phase π-α. Here, the predetermined value is a value sufficiently smaller than the rated current of the semiconductor power converter 13. As an example, the predetermined value is, for example, about 0% to 10% of the rated current of the semiconductor power converter 13, but depending on the application environment of the chopper circuit 1, it may be a value that exceeds 10% of the rated current of the semiconductor power converter 13.

Next, the control in the semiconductor power converter control unit 15 will be described in more detail. The operation of the chopper circuit 1 according to the first modified example of the first embodiment shown in FIG. 3 and the operation of the chopper circuit 1 according to the second modified example of the first embodiment shown in FIG. 4 can be considered to be similar to the operation of the chopper circuit 1 according to the first embodiment shown in FIG. 1, so here, the DC capacitor voltage control in the semiconductor power converter control unit 15 in the chopper circuit 1 according to the first embodiment shown in FIG. 1 will be described.

The DC capacitor voltage control system in the semiconductor power converter control unit 15 is composed of a DC voltage central control system, an inductor current control system, and an individual balance control system.

Figure 8 is a block diagram showing a DC capacitor voltage control system in a semiconductor power converter control unit in a chopper circuit according to the first embodiment of the present disclosure, where (A) shows a DC voltage central control system, (B) shows an inductor current control system, and (C) shows an individual balance control system.

Focusing on the operation of the auxiliary power converter 19 shown in Fig. 1, in the case where the direction of the power flow is from the first DC voltage _vdc1 side to the second DC voltage _vdc2 side, when the first switch unit 11 is on and the second switch unit 12 is off, the auxiliary power converter 19 absorbs power from the first DC voltage _vdc1 side, and when the first switch unit 11 is off and the second switch unit 12 is on, the auxiliary power converter 19 discharges power to the second DC voltage _vdc2 side. Since the relationship of Equation 14 holds in a steady state, the average value of the power flowing in and out of the auxiliary power converter 19 in one period is zero. In other words, since no steady flow of power in and out of the auxiliary power converter 19 occurs, the DC component of the DC capacitor voltage in the capacitor in each semiconductor power converter 13 (chopper cell) does not ideally fluctuate. However, in reality, the voltage fluctuates due to the influence of transient fluctuations and converter losses, so the semiconductor power converter control section 15 executes DC capacitor voltage control for the capacitors in each semiconductor power converter 13 (chopper cell).

As shown in Fig. 8A, the DC voltage central control system configures a feedback loop that makes the arithmetic average value _vCave of the DC capacitor voltages used in all semiconductor power converters 13 (chopper cells) follow the capacitor voltage command value _vC ^* . Specifically, this is realized by adjusting the _DC component iDc contained in the inductor current i. _vCave is expressed by Equation 32.

When v _C ^* > v _Cave , the DC component command value i _dc ^* increases, and the active power flowing into each semiconductor power converter 13 increases. As a result, the DC capacitor voltage increases. On the other hand, when v _C ^* < v _Cave , the DC component command value i _dc ^* decreases, and power flows out of each semiconductor power converter 13. As a result, the DC capacitor voltage decreases. Note that v _Cave contains an AC component whose main component is f _CM , so the AC component is removed using a moving average filter or the like before being provided to the DC capacitor control system.

As shown in Fig. 8(B), the inductor current control system aims to make the inductor current i follow the current command value i ^* as described above. The DC component command value i _dc ^* in Fig. 8(B) is the DC current command value given in Fig. 8(A). The AC current command value i _ac ^* is expressed as Equation 33 in the first mode in which the AC component command value includes the fundamental wave AC component and the tertiary wave AC component, and is expressed as Equation 34 in the second mode in which the AC component command value has a trapezoidal wave AC. Here, θ=2πf _SM t.

In the inductor current control system, the above-mentioned lead compensation control is performed, so that the lead compensation component shown in Equation 8 is added to the AC current command value shown in Equation 33 or Equation 34, and finally, the voltage command value v ^* is output.

8C, the individual balance control system realizes voltage balance by forming effective power between the output voltage v and the inductor current i of each semiconductor power converter 13. Finally, it outputs a voltage command value v _Bj ^* .

9 is a block diagram showing a voltage command value for each semiconductor power converter (chopper cell) in the chopper circuit according to the first embodiment of the present disclosure. In FIG. 9, v _f ^* represents a voltage command value of a feedforward term, and for example, v _f ^* is given by Equation 11.

However, during the dead time period when the first switch unit 11 and the second switch unit 12 are both off, if the inductor current i is greater than zero (i>0), then " _vf ^* = _vdc2 ," and if the inductor current i is less than zero (i<0), then " _vf ^* = _vdc1 ." Each chopper cell voltage command value _vj ^* (j:1-N) is normalized by the DC capacitor voltage, and then compared with a triangular wave of carrier frequency _fSA whose maximum value is 1 and minimum value is 0.

The above is the configuration of the DC capacitor voltage control system of the chopper circuit 1 according to the first embodiment of the present disclosure.

Next, we will explain the simulation results of the chopper circuit 1 according to the first embodiment of the present disclosure.

FIG. 10 is a diagram showing circuit constants used in the simulation of the chopper circuit according to the first embodiment of the present disclosure. "PSCAD/EMTDC" was used for the simulation. The number N of semiconductor power converters 13 (number of chopper cells) provided in the chopper circuit 1 was set to 3. The first DC voltage v _dc1 was set to 1.5 [kV], the second DC voltage v _dc2 was set to 0.75 [kV], and the DC capacitor voltage V _C of the capacitor in each semiconductor power converter 13 (chopper cell) was set to 0.6 [kV]. The carrier frequency f _SM of the main power converter 10 was set to 450 [Hz], and the carrier frequency f _SA of the auxiliary power converter 19 was set to 7.5 [kHz]. Since phase shift PWM is applied to each chopper cell, the equivalent carrier frequency of the auxiliary power converter 19 is 22.5 [kHz] (=Nf _SA ). Note that this simulation is intended to confirm the principle, and therefore an ideal state is assumed. That is, an analog control system with zero control delay was assumed, and an ideal switch with zero dead time was used.

11 is a diagram showing a simulation waveform when 152 [kW] of power is transmitted from the first DC voltage side to the second DC voltage side when current control is performed based on a current command value including an AC component command value according to the first form in the chopper circuit according to the first embodiment of the present disclosure, where (A) shows an inductor current, (B) shows a voltage appearing at both ends of the first switch unit and the second switch unit, (C) shows a current flowing through the first switch unit, (D) shows a current flowing through the second switch unit, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). In FIG. 11(B), the voltage v _S1 appearing at both ends of the first switch unit 11 is shown by a solid line, and the voltage v _S2 appearing at both ends of the second switch unit 12 is shown by a dashed line. In FIG. 11E, DC capacitor voltages v _C1 , v _C2 and v _C3 of each semiconductor power converter 13 (chopper cell) are indicated by a solid line, a dashed line and a dashed dotted line, respectively.

Focusing on the inductor current i shown in Fig. 11(A), it can be seen that a negative DC current is superimposed on the AC components of the fundamental wave frequency of 450 [Hz] and the third wave frequency of 1350 [Hz]. Focusing on the voltages _vS1 and vS2 (forward voltages of the semiconductor valve device) appearing at both ends of the first switch section 11 and the second switch section 12 shown in Fig. 11(B), when _vS1 = 0, which means that _the first switch section 11 is in the on state, the current flowing through the first switch section 11 is _i1 = i as shown in Fig. 11(C), and the current flowing through the second switch section 12 is _i2 = 0 as shown in Fig. 11(D). When the second switch unit 12 is in the ON state (v _S1 =v _dc1 -v _dc2 =750V), the current flowing through the first switch unit 11 is i ₁ =0 as shown in FIG. 11C, and the current flowing through the second switch unit 12 is i ₂ =-i as shown in FIG. 11D. From this simulation result, it can be seen that by applying the lead compensation control expressed by Equation 8 and giving the modulation factor d expressed by Equation 24, the first switch unit 11 and the second switch unit 12 can achieve soft switching operation (i.e., switching operation at a timing when the flowing current is a small predetermined value or less (for example, zero)) both at turn-off and turn-on. Therefore, no switching loss occurs in the first switch unit 11 and the second switch unit 12.

In addition, no step-like current change occurs in either the current _i1 flowing through the first switch section 11 shown in Fig. 11(C) or the current _i2 flowing through the second switch section 12 shown in Fig. 11(D). Therefore, no overvoltage due to a step-like current occurs in the chopper circuit 1.

11(E), the DC capacitor voltages _vC1 , _vC2 , and _vC3 include DC and AC components, and it can be seen that the DC components closely follow the command value of 600 V. As for the AC components, an AC component of 450 Hz exists, but its magnitude is sufficiently small compared to the DC component.

12 is a diagram showing a simulation waveform when 152 [kW] of power is transmitted from the second DC voltage side to the first DC voltage side when current control is performed based on a current command value including an AC component command value according to the second form in the chopper circuit according to the first embodiment of the present disclosure, where (A) shows an inductor current, (B) shows a voltage appearing at both ends of the first switch unit and the second switch unit, (C) shows a current flowing through the first switch unit, (D) shows a current flowing through the second switch unit, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). In FIG. 12(B), the voltage v _S1 appearing at both ends of the first switch unit 11 is shown by a solid line, and the voltage v _S2 appearing at both ends of the second switch unit 12 is shown by a dashed line. In FIG. 12E, DC capacitor voltages v _C1 , v _C2 and v _C3 of each semiconductor power converter 13 (chopper cell) are indicated by a solid line, a dashed line and a dashed dotted line, respectively.

As shown in Fig. 12(A), a positive DC current is superimposed on the inductor current i. Focusing on the voltages v _S1 and v _S2 (forward voltages of the semiconductor valve device) appearing at both ends of the first switch section 11 and the second switch section 12 shown in Fig. 12(B), when v _S1 =0, which indicates that the first switch section 11 is in the ON state, the current i ₁ =i flows through the first switch section 11 as shown in Fig. 12(C), and the current i ₂ =0 flows through the second switch section 12 as shown in Fig. 12(D). When "v _S1 =v _dc1 -v _dc2 =750 [V]", which indicates that the second switch section 12 is in the ON state, the current i ₁ =0 flows through the first switch section 11 as shown in Fig. 12(C), and the current i ₂ =-i flows through the second switch section 12 as shown in Fig. 12(D). From this simulation result, it can be seen that by applying the lead compensation control expressed by Equation 8 and providing the modulation factor d expressed by Equation 24, soft switching operation (i.e., switching operation at a timing when the flowing current is a minute predetermined value or less (e.g., zero)) can be realized both at turn-off and turn-on in the first switch section 11 and the second switch section 12. Therefore, no switching loss occurs in the first switch section 11 and the second switch section 12.

In addition, no step-like current change occurs in either the current _i1 flowing through the first switch section 11 shown in Fig. 12(C) or the current _i2 flowing through the second switch section 12 shown in Fig. 12(D). Therefore, no overvoltage due to a step-like current occurs in the chopper circuit 1.

As shown in Fig. 12(E), the DC capacitor voltages _vC1 , _vC2 , and _vC3 include DC and AC components, and it can be seen that the DC components closely follow the command value of 600 V. The AC components have a shape closer to a triangular wave than that shown in Fig. 11(E).

13 is a diagram showing a simulation waveform in a case where the direction of power transmission is reversed between the first DC voltage side and the second DC voltage side for 2.2 [ms] when current control is performed based on a current command value including an AC component command value according to the second form in the chopper circuit according to the first embodiment of the present disclosure, where (A) shows an inductor current, (B) shows a voltage appearing at both ends of the first switch unit and the second switch unit, (C) shows a current flowing through the first switch unit, (D) shows a current flowing through the second switch unit, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). In FIG. 13(B), the voltage v _S1 appearing at both ends of the first switch unit 11 is shown by a solid line, and the voltage v _S2 appearing at both ends of the second switch unit 12 is shown by a dashed line. In FIG. 13E, DC capacitor voltages v _C1 , v _C2 and v _C3 of each semiconductor power converter 13 (chopper cell) are indicated by a solid line, a dashed line and a dashed dotted line, respectively.

As shown in FIG. 13, even when the direction of power transmission is changed rapidly at 2.2 ms, the chopper circuit 1 operates satisfactorily without causing overvoltage or overcurrent. This shows that the auxiliary power converter 19 has a high-speed current control function.

The above is a description of the chopper circuit according to the first embodiment of the present disclosure. Next, we will explain the chopper circuit according to the second embodiment of the present disclosure.

Figure 14 is a circuit diagram showing a chopper circuit according to a second embodiment of the present disclosure.

A chopper circuit 2 according to the second embodiment of the present disclosure performs bidirectional voltage conversion between a first DC voltage _vdc1 between a pair of first external connection terminals _T1 and _G1 and a second DC voltage _vdc2 between a pair of second external connection terminals _T2 and _G2 . A DC power supply is connected to one of the first external connection terminals _T1 and _G1 and the second external connection terminals _T2 and _G2 , and a load or another DC power supply is connected to the other.

For example, when a DC power supply is connected to the first external connection terminals _T1 and _G1 and a load is connected to the second external connection terminals _T2 and _G2 , the chopper circuit 2 operates as a step-down chopper. In this case, the voltage output by the DC power supply is the first DC voltage _vdc1 , and the voltage applied to the load is the second DC voltage _vdc2 .

For example, when a load is connected to the first external connection terminals _T1 and _G1 and a DC power supply is connected to the second external connection terminals _T2 and _G2 , the chopper circuit 2 operates as a boost chopper. In this case, the voltage applied to the load is the first DC voltage _vdc1 , and the voltage output by the DC power supply is the second DC voltage _vdc2 .

Also, for example, a DC power supply may be connected to the first external connection terminals _T1 and _G1 , and another DC power supply may be connected to the second external connection terminals _T2 and _G2 .

The chopper circuit 2 includes a first switch section 21, a second switch section 22, a third switch section 23, a fourth switch section 24, a semiconductor power converter 25, and an inductor 26. The chopper circuit 2 also includes a semiconductor power converter control section 27 and a switch control section 28 as its control system.

The first switch section 21, the second switch section 22, the third switch section 23, and the fourth switch section 24 are semiconductor valve devices capable of unidirectional current interruption. The first switch section 21, the second switch section 22, the third switch section 23, and the fourth switch section 24 are composed of a semiconductor switching element that conducts in one direction when turned on and a feedback diode connected in reverse parallel to the semiconductor switching element. Examples of semiconductor switching elements include IGBTs, SiC (Silicon Carbide)-MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors), thyristors, GTOs (Gate Turn-OFF Thyristors), and transistors, but the type of semiconductor switching element itself does not limit the present invention, and other semiconductor elements may be used.

In addition, an overvoltage suppression snubber circuit may be connected in parallel to each of the semiconductor valve devices of the first switch section 21, the second switch section 22, the third switch section 23, and the fourth switch section 24.

The first switch section 21, the second switch section 22, the third switch section 23, and the fourth switch section 24 are connected in series to one another so that the conduction directions when on are the same. The connection point between the first switch section 21 and the second switch section 22 is denoted by _P3 , the connection point between the second switch section 22 and the third switch section 23 is denoted by _P4 , and the connection point between the third switch section 23 and the fourth switch section 24 is denoted by _P5 . The connection point of the fourth switch section 24 on the opposite side to the side to which the third switch section 23 is connected is denoted by _P6 .

The terminal opposite to the connection point _P3 between the first switch section 21 and the second switch section 22 is defined as the positive terminal _T1 of the first external connection terminals. The terminal opposite to the connection point _P5 between the third switch section 23 and the fourth switch section 24 is defined as the ground terminal _G1 of the first external connection terminals. The positive terminal _T1 and the ground terminal _G1 constitute a pair of first external connection terminals.

A positive terminal _T2 of the second external connection terminals is provided on a wiring extending from a connection point _P4 between the second switch section 22 and the third switch section 23. A ground terminal G2 of the second external connection terminals is provided on a wiring extending from a connection point _P6 of the fourth switch section 24 on the side opposite to the side to _which the third switch section 23 is connected. The positive terminal _T2 and the ground terminal _G2 form a pair of second external connection terminals.

Here, the forward voltage of the first switch section 21 (i.e., the potential difference between the positive terminal _T1 of the first external connection terminal and the connection point _P3 ) is represented by _vS1 . The forward voltage of the second switch section 22 (i.e., the potential difference between the connection points _P3 and _P4 ) is represented by _vS2 . The forward voltage of the third switch section 23 (i.e., the potential difference between the connection points _P4 and _P5 ) is represented by _vS3 . The forward voltage of the fourth switch section 24 (i.e., the potential difference between the connection points _P5 and _P6 ) is represented by _vS4 .

In addition, the current flowing between the positive terminal _T1 of the first external connection terminal and the connection point _P3 is represented by _i1 and referred to as "the current flowing on the first DC voltage _vdc1 side," and the current flowing between the connection point _P4 and the positive terminal _T2 of the second external connection terminal is represented by _i2 and referred to as "the current flowing on the second DC voltage _vdc2 side."

In this specification, a set of power converters consisting of a first switch unit 21, a second switch unit 22, a third switch unit 23, and a fourth switch unit 24 is referred to as a main power converter 20. As described below, while either one of the set of the first switch unit 21 and the third switch unit 23 and the set of the second switch unit 22 and the fourth switch unit 24 is controlled to be on, the other set is controlled to be off.

A variable controlled voltage source using a semiconductor power converter 25 and an inductor 26 are provided on a wiring connecting a connection point _P3 between the first switch section 21 and the second switch section 22 and a connection point _P5 between the third switch section 23 and the fourth switch section 24.

The semiconductor power converter 25 is provided on the wiring connecting the connection point _P3 and the connection point _P5 , either singly or in a state where a plurality of semiconductor power converters 25 are cascaded to each other. In this specification, the single semiconductor power converter 25 or the plurality of semiconductor power converters 25 is referred to as an auxiliary power converter 29. In this specification, when there is one semiconductor power converter 25, the side to which an inductor 26 described later is connected is referred to as a "first DC side", and when a plurality of semiconductor power converters 25 are cascaded to each other, the side to which another semiconductor power converter 25 different from the semiconductor power converter 25 is connected is also referred to as a "first DC side". In addition, the DC side opposite to the "first DC side" is referred to as a "second DC side". As an example, FIG. 14 shows a case where a plurality of semiconductor power converters 25 (N, where N is an integer of 2 or more) are cascaded to each other on the first DC side. Hereinafter, the number of cascades of semiconductor power converters 25 is represented by j (where j is a natural number from 1 to N). By simply adjusting the number of cascade-connected semiconductor power converters 25 as appropriate, the chopper circuit 2 can easily be made to withstand high voltage.

As shown in FIG. 2, the semiconductor power converter 25 is configured as a bidirectional chopper cell having a DC-DC converter 131 and a capacitor 132. That is, the semiconductor power converter 25 is configured as a chopper cell including two semiconductor switches connected in series and a DC capacitor connected in parallel to the two semiconductor switches, with each terminal of one of the two semiconductor switches serving as an output terminal. The DC-DC converter 131 and the capacitor 132 are as described with reference to FIG. 2. When the chopper circuit 2 is operated, the DC-DC converter 131 is operated to initially charge the capacitor 132. Here, the voltage of the DC capacitor of each semiconductor power converter 25 is v _Cj , and the voltage of the first DC side of the auxiliary power converter 29 is v. Although details will be described later, the inductor current i is controlled using the auxiliary power converter 29, so that the inductor 26 and the auxiliary power converter 29 operate as a controlled current source. Note that in FIG. 14, the auxiliary power converter is realized by cascading a plurality of semiconductor power converters 25 (chopper cells), but any semiconductor power converter having a similar function can be used instead.

The inductor 26 is connected in series to the semiconductor power converter 25 on a wiring connecting a connection point _P3 between the first switch section 21 and the second switch section 22 and a connection point _P5 between the third switch section 23 and the fourth switch section 24. The inductor current flowing through the inductor 26 is represented as i.

Therefore, the semiconductor power converter 25 and the inductor 26 are provided on the same wiring branched from a connection point _P3 on the wiring connecting the first switch section 21 and the second switch section 22 in the main power converter 20. In the example shown in Fig. 14, the inductor 26 is provided between the connection point _P3 and the semiconductor power converter 25, and the second external connection terminals _T2 and _G2 are provided at a connection point _P6 on the auxiliary power converter 29 opposite to the side to which the inductor 26 is connected (i.e., the side opposite to the side to which the inductor 26 is connected of the set of multiple semiconductor power converters 25), but the arrangement order of these semiconductor power converters 25 and the inductor 26 can be arbitrarily designed as shown in Fig. 5 on the wiring on which the semiconductor power converter 25 is provided between the connection point _P3 and the connection point _P5 , as in the first embodiment described above.

The semiconductor power converter control unit 27 controls the power conversion operation of the semiconductor power converter 25 so as to output a current having a DC component and an AC component. In particular, the semiconductor power converter control unit 27 performs current control to cause the semiconductor power converter 25 to output an inductor current that follows a current command value consisting of a DC component command value and an AC component command value. In this current control, the semiconductor power converter control unit 27 performs lead compensation control to compensate for the phase delay of the inductor current relative to the current command value. For this reason, the AC component command value of the current command value has a lead compensation component to compensate for the phase delay of the inductor current relative to the current command value. This lead compensation component is generated based on at least the proportional gain in the current control and the inductance of the inductor 26. Details of the lead compensation control in the current control by the semiconductor power converter control unit 27 will be described later.

The switch control unit 28 controls one of the pair of the first switch unit 21 and the third switch unit 23 and the pair of the second switch unit 22 and the fourth switch unit 24 to ON and controls the other pair to OFF. When the semiconductor power converter control unit 27 controls the value of the current output by the semiconductor power converter 25 to be equal to or lower than a predetermined value, the switch control unit 28 executes switching from ON to OFF and switching from OFF to ON for each pair of the pair of the first switch unit 21 and the third switch unit 23 and the pair of the second switch unit 22 and the fourth switch unit 24. Here, the predetermined value is a value sufficiently smaller than the rated current of the semiconductor power converter 25. As an example, the predetermined value is, for example, a value of about 0% to 10% of the rated current of the semiconductor power converter 25, but depending on the application environment of the chopper circuit 2, it may be a value exceeding 10% of the rated current of the semiconductor power converter 25.

The arithmetic processing device (not shown) has a semiconductor power converter control unit 27 and a switch control unit 28. The semiconductor power converter control unit 27 and the switch control unit 28 are functional modules realized by, for example, a computer program executed on the arithmetic processing device. For example, when the semiconductor power converter control unit 27 and the switch control unit 28 are constructed in the form of a computer program, the functions of the semiconductor power converter control unit 27 and the switch control unit 28 can be realized by operating the arithmetic processing device according to the computer program. The computer program for executing the processing of the semiconductor power converter control unit 27 and the switch control unit 28 may be provided in a form recorded on a computer-readable recording medium such as a semiconductor memory, a magnetic recording medium, or an optical recording medium. Alternatively, the semiconductor power converter control unit 27 and the switch control unit 28 may be realized as a semiconductor integrated circuit in which a computer program for realizing the functions of each unit is written. The operation of the semiconductor power converter control unit 27 and the switch control unit 28 will be described in detail later.

Next, the operating principle of the chopper circuit 2 according to the second embodiment of the present disclosure will be described. The operating principle of the semiconductor power converter control unit 27 and the switch control unit 28 is similar to the operating principle of the semiconductor power converter control unit 15 and the switch control unit 16 in the first embodiment. Below, a continuous control system with zero control delay is assumed.

The chopper circuit 2 according to the second embodiment can perform bidirectional voltage conversion between a first DC voltage _vdc1 between a pair of first external connection terminals _T1 and _G1 and a second DC voltage _vdc2 between a pair of second external connection terminals _T2 and _G2 . However, in the chopper circuit 2 according to the second embodiment, it is necessary to have a relationship " _vdc1 > _vdc2 " in which the first DC voltage _vdc1 is greater than the second DC voltage _vdc2 .

The semiconductor power converter control unit 27 controls the power conversion operation of the semiconductor power converter 25 so as to output a current having a DC component and an AC component. In particular, the semiconductor power converter control unit 27 performs current control to cause the semiconductor power converter 25 to output an inductor current that follows a current command value consisting of a DC component command value and an AC component command value. In this current control, the semiconductor power converter control unit 27 performs lead compensation control to compensate for the phase lag of the inductor current relative to the current command value. For this reason, the AC component command value of the current command value has a lead compensation component to compensate for the phase lag of the inductor current relative to the current command value. This lead compensation component is generated based on at least the proportional gain in the current control and the inductance of the inductor 26.

The current command value i ^* in the current control of the semiconductor power converter control unit 27 is composed of a DC component command value i _dc ^* and an AC component command value i _ac ^* . In the second embodiment, as in the first embodiment, the AC component command value i _ac ^* may be of a first form in which the AC component command value i _ac ^* includes a fundamental wave AC component and a tertiary wave AC component, or of a second form in which the AC component command value i _ac ^* includes a trapezoidal wave AC component.

First, the AC component command value according to the first form in the chopper circuit 2 according to the second embodiment will be described. The AC component command value according to the first form has a fundamental wave AC component having a fundamental frequency and a tertiary wave AC component having a frequency three times the fundamental frequency.

15 is a diagram showing ideal waveforms of each part when current control is performed based on a current command value including an AC component command value according to the first form in the chopper circuit according to the second embodiment of the present disclosure, where (A) shows the relationship between a triangular wave and a modulated wave used in the switch control part, (B) shows the voltage appearing at both ends of the first switch part and the voltage appearing at both ends of the second switch part, (C) shows the output voltage of the auxiliary power converter, (D) shows the inductor current, (E) shows the current flowing on the first DC voltage side, and (F) shows the current flowing on the second DC voltage side. In FIG. 15(A), the triangular wave v _tri is shown by a solid line, and the modulated wave d is shown by a dashed line. In FIG. 15(F), the current i ₁ flowing on the first DC voltage v _dc1 side is shown by a solid line, and the current i ₂ flowing on the second DC voltage v _dc2 side is shown by a dashed line.

The operation of the chopper circuit 2 according to the second embodiment differs between the case where the second DC voltage _vdc2 is less than half the first DC voltage _vdc1 ( _vdc2 < _0.5vdc1 ) and the case where it is more than half the first DC voltage _vdc1 ( _vdc2 > _0.5vdc1 ), but the operating principle can be considered similar. The case where the second DC voltage _vdc2 is less than half the first DC voltage _vdc1 ( _vdc2 < _0.5vdc1 ) will be described below.

The on/off timing of the first switch section 21 and the second switch section 22 is determined by the switch control section 28 based on a comparison result between the triangular wave v _tri , which has a minimum value of 0 and a maximum value of 1, and the modulation wave d. The method of determining the modulation wave d will be described in detail later, but it is determined based on the relationship between the first DC voltage v _dc1 and the second DC voltage v _dc2 . The switch control section 28 controls one of the pair of the first switch section 21 and the third switch section 23 and the pair of the second switch section 22 and the fourth switch section 24 to be on, and controls the other pair to be off. For example, as shown in FIG. 15A, when the triangular wave v _tri is smaller than the modulation wave d, the switch control section 28 controls the pair of the first switch section 21 and the third switch section 23 to be on, and controls the pair of the second switch section 22 and the fourth switch section 24 to be off. In this case, as shown in Fig. 15B, the voltage appearing across the first switch section 21 is v _S1 =0, and the voltage appearing across the second switch section 22 is v _S2 =v _dc1 -v _dc2 . For example, when the triangular wave v _tri is larger than the modulation wave d as shown in Fig. 15A, the switch control section 28 controls the pair of the first switch section 21 and the third switch section 23 to be turned off, and controls the pair of the second switch section 22 and the fourth switch section 24 to be turned on. In this case, as shown in Fig. 15B, the voltage appearing across the first switch section 21 is v _S1 =v _dc1 -v _dc2 , and the voltage appearing across the second switch section 22 is v _S2 =0.

The waveforms of the respective parts of the chopper circuit 2 according to the second embodiment are basically the same as those of the respective parts of the chopper circuit 1 according to the first embodiment, except for the current _i1 flowing between the positive terminal _T1 of the first external connection terminal and the connection point _P3 shown in Fig. 15(E) and the current _i2 flowing between the connection point _P4 and the positive terminal _T2 of the second external connection terminal shown in Fig. 15(F). In the chopper circuit 1 according to the first embodiment, when the first switch section 11 is on, the current flowing through the first switch section 11 is _i1 = i, and the current flowing through the second switch section 12 is _i2 = 0. When the second switch section 12 is on, the current flowing through the first switch section 11 is _i1 = 0, and the current flowing through the second switch section 12 is _i2 = -i. In contrast, in the chopper circuit 2 according to the second embodiment, when the pair of the first switch section 21 and the third switch section 23 is on, the current _i1 flowing between the positive terminal _T1 of the first external connection terminal and the connection point _P3 , and the current _i2 flowing between the connection point _P4 and the positive terminal _T2 of the second external connection terminal are both i ( _i1 = _i2 = i), and when the pair of the second switch section 22 and the fourth switch section 24 is on, the current flowing between the positive terminal _T1 of the first external connection terminal and the connection point _P3 is _i1 = 0, and the current flowing between the connection point _P4 and the positive terminal _T2 of the second external connection terminal is _i2 = -i.

By referring to Equations 14 to 20, the relational equation for the phase α of the chopper circuit 2 according to the second embodiment is derived as shown in Equation 35.

Equation 35 is a quadratic equation with respect to phase α, and solving it for phase α (where 0<α<π/2) gives Equation 36.

15A, the slope of the triangular wave v _tri from phase π/2 to phase 3π/2 is "1/π", and the value of the triangular wave v _tri at phase π is "0.5". The value of the triangular wave v _tri at phase π-α is "d". From these relationships, d can be expressed as in Equation 24, similar to the first embodiment.

As described above, the modulating wave d is determined based on Equation 24, and the phase α is determined based on Equation 36. When the triangular wave v _tri is smaller than the modulating wave d determined based on Equation 24, the switch control section 28 controls the pair of the first switch section 21 and the third switch section 23 to be turned on, and controls the pair of the second switch section 22 and the fourth switch section 24 to be turned off. When the triangular wave v _tri is larger than the modulating wave d determined based on Equation 24, the switch control section 28 controls the pair of the first switch section 21 and the third switch section 23 to be turned off, and controls the pair of the second switch section 22 and the fourth switch section 24 to be turned on. The switch control unit 28 executes on-to-off and off-to-on switching for the pair of the first switch unit 21 and the third switch unit 23 and the pair of the second switch unit 22 and the fourth switch unit 24 when the current output by the semiconductor power converter 25 is controlled to zero by the semiconductor power converter control unit 27 at phase α and phase π-α.

Next, a second form of the AC component command value in the chopper circuit 2 according to the second embodiment will be described. The AC component command value according to the second form has a trapezoidal wave AC component.

16 is a diagram showing ideal waveforms of each part when current control is performed based on a current command value including an AC component command value according to the second form in the chopper circuit according to the second embodiment of the present disclosure, where (A) shows the relationship between a triangular wave and a modulated wave used in the switch control part, (B) shows the voltage appearing at both ends of the first switch part and the voltage appearing at both ends of the second switch part, (C) shows the output voltage of the auxiliary power converter, (D) shows the inductor current, (E) shows the current flowing on the first DC voltage side, and (F) shows the current flowing on the second DC voltage side. In FIG. 16(A), the triangular wave v _tri is shown by a solid line, and the modulated wave d is shown by a dashed line. In FIG. 16(F), the current i ₁ flowing on the first DC voltage v _dc1 side is shown by a solid line, and the current i ₂ flowing on the second DC voltage v _dc2 side is shown by a dashed line.

The on/off timing of the first switch section 21 and the second switch section 22 is determined by the switch control section 28 based on a comparison result between the triangular wave v _tri , which has a minimum value of 0 and a maximum value of 1, and the modulation wave d. The method of determining the modulation wave d will be described in detail later, but it is determined based on the relationship between the first DC voltage v _dc1 and the second DC voltage v _dc2 . The switch control section 28 controls one of the pair of the first switch section 21 and the third switch section 23 and the pair of the second switch section 22 and the fourth switch section 24 to be on, and controls the other pair to be off. For example, as shown in FIG. 16A, when the triangular wave v _tri is smaller than the modulation wave d, the switch control section 28 controls the pair of the first switch section 21 and the third switch section 23 to be on, and controls the pair of the second switch section 22 and the fourth switch section 24 to be off. In this case, as shown in Fig. 16B, the voltage appearing across the first switch section 21 is v _S1 =0, and the voltage appearing across the second switch section 22 is v _S2 =v _dc1 -v _dc2 . For example, when the triangular wave v _tri is larger than the modulation wave d as shown in Fig. 16A, the switch control section 28 controls the pair of the first switch section 21 and the third switch section 23 to be turned off, and controls the pair of the second switch section 22 and the fourth switch section 24 to be turned on. In this case, as shown in Fig. 16B, the voltage appearing across the first switch section 21 is v _S1 =v _dc1 -v _dc2 , and the voltage appearing across the second switch section 22 is v _S2 =0.

Here, we explain how to determine the modulation wave d and phase α in the second form in which the AC command value includes a trapezoidal AC component.

The waveforms of each part of the chopper circuit 2 according to the second embodiment are basically the same as those of each part of the chopper circuit 1 according to the first embodiment, except for the current _i1 flowing between the positive terminal _T1 of the first external connection terminal and the connection point _P3 shown in Fig. 16(E) and the current _i2 flowing between the connection point _P4 and the positive terminal _T2 of the second external connection terminal shown in Fig. 16(F). In the chopper circuit 1 according to the first embodiment, when the first switch unit 11 is on, the current flowing through the first switch unit 11 is _i1 = i, and the current flowing through the second switch unit 12 is _i2 = 0. When the second switch unit 12 is on, the current flowing through the first switch unit 11 is _i1 = 0, and the current flowing through the second switch unit 12 is _i2 = -i. In contrast, in the chopper circuit 2 according to the second embodiment, when the pair of the first switch section 21 and the third switch section 23 is on, the current i1 flowing on the first DC voltage _vdc1 side (the current flowing between the positive terminal _T1 of the first external connection terminal and the connection point _P3 ) and the current _i2 flowing on the second DC voltage _vdc2 side (the current flowing between the connection point _P4 and the positive terminal _T2 of the second external connection terminal) are _both i ( _i1 = _i2 = i), and when the pair of the second switch section 22 and the fourth switch section 24 is on, the current flowing between the positive terminal _T1 of the first external connection terminal and the connection point _P3 is _i1 = 0, and the current flowing between the connection point _P4 and the positive terminal _T2 of the second external connection terminal is _i2 = -i.

By referring to Equations 25 to 29, the relational equation for the phase α of the chopper circuit 2 according to the second embodiment is derived as shown in Equation 37.

Transforming equation 37 gives us equation 38.

Equation 38 is a quadratic equation with respect to phase α, and solving it for phase α (where 0<α<π/2) gives Equation 39.

16A, the slope of the triangular wave v _tri from phase π/2 to phase 3π/2 is "1/π", and the value of the triangular wave v _tri at phase π is "0.5". The value of the triangular wave v _tri at phase π-α is "d". From these relationships, d can be expressed as in Equation 24, similar to the first embodiment.

As described above, the modulating wave d is determined based on Equation 24, and the phase α is determined based on Equation 39. When the triangular wave v _tri is smaller than the modulating wave d determined based on Equation 24, the switch control section 28 controls the pair of the first switch section 21 and the third switch section 23 to be turned on, and controls the pair of the second switch section 22 and the fourth switch section 24 to be turned off. When the triangular wave v _tri is larger than the modulating wave d determined based on Equation 24, the switch control section 28 controls the pair of the first switch section 21 and the third switch section 23 to be turned off, and controls the pair of the second switch section 22 and the fourth switch section 24 to be turned on. The switch control unit 28 executes on-to-off and off-to-on switching for the pair of the first switch unit 21 and the third switch unit 23 and the pair of the second switch unit 22 and the fourth switch unit 24 when the current output by the semiconductor power converter 25 is controlled to zero by the semiconductor power converter control unit 27 at phase α and phase π-α.

Next, we will explain the simulation results of the chopper circuit 2 according to the second embodiment of the present disclosure.

FIG. 17 is a diagram showing circuit constants used in the simulation of the chopper circuit according to the second embodiment of the present disclosure. "PSCAD/EMTDC" was used for the simulation. The number N of semiconductor power converters 13 (number of chopper cells) provided in the chopper circuit 1 was set to 3. The first DC voltage v _dc1 was set to 1.5 [kV], the second DC voltage v _dc2 was set to 0.6 [kV], and the DC capacitor voltage V _C of the capacitor in each semiconductor power converter 13 (chopper cell) was set to 0.6 [kV]. The carrier frequency f _SM of the main power converter 10 was set to 450 [Hz], and the carrier frequency f _SA of the auxiliary power converter 19 was set to 7.5 [kHz]. Since phase shift PWM is applied to each chopper cell, the equivalent carrier frequency of the auxiliary power converter 19 is 22.5 [kHz] (=Nf _SA ). Note that this simulation is intended to confirm the principle, and therefore an ideal state is assumed. That is, an analog control system with zero control delay was assumed, and an ideal switch with zero dead time was used.

18 is a diagram showing a simulation waveform when 200 [kW] of power is transmitted from the second DC voltage side to the first DC voltage side when current control is performed based on a current command value including an AC component command value according to the first form in the chopper circuit according to the second embodiment of the present disclosure, where (A) shows an inductor current, (B) shows a voltage appearing at both ends of each of the first switch unit and the second switch unit, (C) shows a current flowing on the first DC voltage side, (D) shows a current flowing on the second DC voltage side, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). In FIG. 18(B), the voltage v _S1 appearing at both ends of the first switch unit 11 is shown by a solid line, and the voltage v _S2 appearing at both ends of the second switch unit 12 is shown by a dashed line. In FIG. 18E, DC capacitor voltages v _C1 , v _C2 and v _C3 of each semiconductor power converter 13 (chopper cell) are indicated by a solid line, a dashed line and a dashed dotted line, respectively.

Focusing on the inductor current i shown in FIG. 18A, it can be seen that a negative DC current is superimposed on the AC components of the fundamental wave frequency of 450 [Hz] and the tertiary wave frequency of 1350 [Hz]. Focusing on the voltages v _S1 and v _S2 (forward voltages of the semiconductor valve device) appearing at both ends of the first switch section 21 and the second switch section 22 in the first unit shown in Figure 18 (B), when the pair of the first switch section 21 and the third switch section 23 is in the on state, v _S1 = v _S3 = 0, as shown in Figure 18 (C), the current i ₁ flowing on the first DC voltage v _dc1 side and the current i ₂ flowing on the second DC voltage v _dc2 side are both i (i ₁ = i ₂ = i ), and when the pair of the second switch section 22 and the fourth switch section 24 is in the on state, "v _S1 = v _dc1 - v _dc2 = 900 [V]", the current i ₁ flowing on the first DC voltage v _dc1 side and the current i 2 flowing on the second DC voltage v _dc2 side are _-i , as shown in Figure 18 (C). From the simulation results, it can be seen that by applying the modulation factor d expressed by Equation 24, soft switching operation (i.e., switching operation at a timing when the flowing current is a minute, predetermined value or less (e.g., zero)) can be realized both at turn-off and turn-on in the first switch section 21, the second switch section 22, the third switch section 23, and the fourth switch section 24. Therefore, no switching loss occurs in the first switch section 21, the second switch section 22, the third switch section 23, and the fourth switch section 24.

Furthermore, no step-like current change occurs in either the current _i1 flowing on the first DC voltage _vdc1 side in the chopper circuit 2 shown in Fig. 18(C) or the current _i2 flowing on the second DC voltage _v2 side in the chopper circuit 2 shown in Fig. 18(D). Therefore, no overvoltage due to a step-like current occurs in the chopper circuit 1.

18(E), the DC capacitor voltages _vC1 , _vC2 , and _vC3 include DC and AC components, and it can be seen that the DC components closely follow the command value of 600 V. As for the AC components, an AC component of 450 Hz exists, but its magnitude is sufficiently small compared to the DC component.

19 is a diagram showing a simulation waveform when 200 [kW] of power is transmitted from the first DC voltage side to the second DC voltage side when current control is performed based on a current command value including an AC component command value according to the second form in the chopper circuit according to the second embodiment of the present disclosure, where (A) shows an inductor current, (B) shows a voltage appearing at both ends of the first switch unit and the second switch unit, (C) shows a current flowing on the first DC voltage side, (D) shows a current flowing on the second DC voltage side, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). In FIG. 19(B), the voltage v _S1 appearing at both ends of the first switch unit 11 is shown by a solid line, and the voltage v _S2 appearing at both ends of the second switch unit 12 is shown by a dashed line. In FIG. 19E, DC capacitor voltages v _C1 , v _C2 and v _C3 of each semiconductor power converter 13 (chopper cell) are indicated by a solid line, a dashed line and a dashed dotted line, respectively.

As shown in Fig. 19(A), a positive DC current is superimposed on the inductor current i. In addition, it can be seen that the phase α expressed by Equation 39 does not change even if the direction of power transmission is changed compared to Fig. 18. Focusing on the voltages v S1 and v _S2 (forward voltages of the semiconductor valve device) appearing at both ends of the first switch section 21 and the second switch section 22 shown in Fig. 19( _B ), when v _S1 =0, which is when the first switch section 11 is in the on state, the current flowing to the first DC voltage v _dc1 side becomes i ₁ =i as shown in Fig. 19(C), and the current flowing to the second switch section 22 becomes i ₂ =i as shown in Fig. 19(D). When the second switch unit 22 is in the ON state, i ₁ =0 flows through _{the first DC voltage v dc1 as} shown in FIG. 19C, and i 2 =i flows through the second DC voltage v _dc2 as shown in FIG. 19D. From the simulation results, it can be seen that soft switching operation (i.e., switching operation at a timing when _the flowing current is a minute predetermined value or less (e.g., _{zero )} ) can be realized in the first switch unit 21 and the second switch unit 22 at both turn-off and turn-on by applying the lead compensation control expressed by Equation 8 and giving the modulation factor d expressed by Equation 24. The same is true for the third switch unit 23 and the fourth switch unit 24. Therefore, no switching loss occurs in the first switch unit 21, the second switch unit 22, the third switch unit 23, and the fourth switch unit 24.

Furthermore, no step-like current change occurs in either the current _i1 flowing on the first DC voltage _vdc1 side (current flowing between the positive terminal _T1 of the first external connection terminal and the connection point _P3 ) shown in Fig. 19(C) or the current i2 flowing on the second DC voltage _vdc2 side (current flowing between the connection point _P4 and the positive terminal _T2 of the second external connection terminal) shown in Fig. ₁₉ (D). Therefore, no overvoltage due to a step-like current occurs in the chopper circuit 2.

As shown in Fig. 19(E), the DC capacitor voltages _vC1 , _vC2 , and _vC3 include DC and AC components, and it can be seen that the DC components closely follow the command value of 600 V. The AC components have a shape closer to a triangular wave than that shown in Fig. 18(E).

20 is a diagram showing a simulation waveform in a case where the direction of power transmission is reversed between the first DC voltage side and the second DC voltage side for 2.2 [ms] when current control is performed based on a current command value including an AC component command value according to the second form in the chopper circuit according to the second embodiment of the present disclosure, where (A) shows an inductor current, (B) shows a voltage appearing at both ends of the first switch unit and the second switch unit, (C) shows a current flowing on the first DC voltage side, (D) shows a current flowing on the second DC voltage side, and (E) shows a DC capacitor voltage of a capacitor in a semiconductor power converter (chopper cell). In FIG. 20(B), the voltage v _S1 appearing at both ends of the first switch unit 21 is shown by a solid line, and the voltage v _S2 appearing at both ends of the second switch unit 22 is shown by a dashed line. In FIG. 13E, DC capacitor voltages v _C1 , v _C2 and v _C3 of each semiconductor power converter 13 (chopper cell) are indicated by a solid line, a dashed line and a dashed dotted line, respectively.

As shown in FIG. 20, even when the direction of power transmission is changed rapidly at 2.2 ms, the chopper circuit 2 operates satisfactorily without causing overvoltage or overcurrent. This shows that the auxiliary power converter 19 has a high-speed current control function.

In this way, according to the first and second embodiments of the present disclosure, a chopper circuit that does not require a filter circuit for smoothing the DC current, is small and lightweight, is easy to process, and is highly versatile can be realized. That is, in order to create the "timing at which the inductor current becomes zero" required to realize the "soft switching operation" that switches the switch unit in the chopper circuits shown in Figures 1, 3, 4, and 14, a PI control method including lead compensation control is used in controlling the inductor current, so that the calculation processing is easy. In addition, since the PI control method including lead compensation control can be applied regardless of the number of chopper circuits, it is highly versatile. In addition, by using an AC component command value in which a fundamental wave AC component is superimposed with a tertiary wave AC component or an AC component command value having a trapezoidal wave AC component for the current command value of the inductor current, the DC current flowing on each of the first DC voltage side and the second DC voltage side can be sufficiently smoothed, so that it is not necessary to connect a large and heavy filter circuit to the chopper circuit, and it is possible to reduce the size and weight of the device including the chopper circuit.

For example, when the first and second embodiments of the present disclosure are applied to a chopper circuit installed in a railway vehicle, DC current smoothing, i.e., low noise, can be realized, and it becomes easier to achieve the low noise performance required for preventing inductive interference in railway vehicles. In addition, since the filter circuit (mainly a reactor) that was conventionally required to complement the low noise performance can be eliminated or reduced in inductance, the power conversion device as a whole can be made smaller and lighter. In addition, when obtaining the effective current value of the DC current output from the chopper circuit compared to the conventional one, the first and second embodiments of the present disclosure can reduce the current peak value, which increases the freedom of power device selection and improves the power factor. The first and second embodiments of the present disclosure can be applied not only to railway vehicles but also to other moving bodies, and the above-mentioned effects can be obtained in such moving bodies as well.

REFERENCE SIGNS LIST 1 Chopper circuit 2 Chopper circuit 10 Main power converter 11, 11-1, 11-2 First switch section 12, 12-1, 12-2 Second switch section 13 Semiconductor power converter 14 Inductor 15 Semiconductor power converter control section 16 Switch control section 19 Auxiliary power converter 20 Main power converter 21 First switch section 22 Second switch section 23 Third switch section 24 Fourth switch section 25 Semiconductor power converter 26 Inductor 27 Semiconductor power converter control section 28 Switch control section 29 Auxiliary power converter 131 DCDC converter 132 Capacitor

Claims (9)

A chopper circuit for converting a voltage between a first DC voltage at a first external connection terminal and a second DC voltage at a second external connection terminal,
a first switch unit having a first external connection terminal;
a second switch section connected in series to the first switch section so that the conduction direction when the second switch section is on is aligned with that of the first switch section, and having a second external connection terminal on an opposite side to the side to which the first switch section is connected;
one or a plurality of semiconductor power converters connected in cascade to each other, the semiconductor power converters being provided on wiring branched from a wiring connecting the first switch unit and the second switch unit;
an inductor connected in series to the semiconductor power converter on a wiring branched from a wiring connecting the first switch unit and the second switch unit;
a semiconductor power converter control unit that performs current control to cause the semiconductor power converter to output an inductor current that follows a current command value including a DC component command value and an AC component command value;
a switch control unit that controls one of the first switch unit and the second switch unit to ON and controls the other to OFF, and that switches the first switch unit and the second switch unit from ON to OFF and from OFF to ON when a value of an inductor current output by the semiconductor power converter is controlled by the semiconductor power converter control unit to be equal to or lower than a predetermined value;
Equipped with
The semiconductor power converter control unit performs lead compensation control in the current control to compensate for a phase lag of the inductor current with respect to the current command value. A chopper circuit for converting a voltage between a first DC voltage at a first external connection terminal and a second DC voltage at a second external connection terminal,
a first switch unit having a first external connection terminal;
a second switch section connected in series to the first switch section so that the conduction direction when the second switch section is on is aligned with that of the first switch section, and having a second external connection terminal on an opposite side to the side to which the first switch section is connected;
one or a plurality of semiconductor power converters connected in cascade to each other, the semiconductor power converters being provided on wiring branched from a wiring connecting the first switch unit and the second switch unit;
an inductor connected in series to the semiconductor power converter on a wiring branched from a wiring connecting the first switch unit and the second switch unit;
a semiconductor power converter control unit that performs current control to cause the semiconductor power converter to output an inductor current that follows a current command value including a DC component command value and an AC component command value;
a switch control unit that controls one of the first switch unit and the second switch unit to ON and controls the other to OFF, and that switches the first switch unit and the second switch unit from ON to OFF and from OFF to ON when a value of an inductor current output by the semiconductor power converter is controlled by the semiconductor power converter control unit to be equal to or lower than a predetermined value;
Equipped with
The first switch unit and the second switch unit are provided in two units each,
The semiconductor power converter control unit performs lead compensation control in the current control to compensate for a phase lag of the inductor current with respect to the current command value. A chopper circuit for converting a voltage between a first DC voltage between a pair of first external connection terminals and a second DC voltage between a pair of second external connection terminals,
a first switch section, a second switch section, a third switch section, and a fourth switch section which are connected in series to each other so that their conduction directions when turned on are aligned;
one or a plurality of semiconductor power converters connected in cascade to each other, the semiconductor power converters being provided on wiring connecting a connection point between the first switch section and the second switch section and a connection point between the third switch section and the fourth switch section;
an inductor connected in series to the semiconductor power converter on a wiring that connects a connection point between the first switch unit and the second switch unit and a connection point between the third switch unit and the fourth switch unit;
a semiconductor power converter control unit that performs current control to cause the semiconductor power converter to output an inductor current that follows a current command value including a DC component command value and an AC component command value;
a switch control unit that controls one of the pair of the first switch unit and the third switch unit and the pair of the second switch unit and the fourth switch unit to ON and controls the other pair to OFF, and that switches each of the pair of the first switch unit and the third switch unit and the pair of the second switch unit and the fourth switch unit from ON to OFF and from OFF to ON when a value of an inductor current output by the semiconductor power converter is controlled by the semiconductor power converter control unit to be equal to or lower than a predetermined value;
Equipped with
a terminal on an opposite side to a connection side between the first switch section and the second switch section, and a terminal on an opposite side to a connection side between the third switch section and the fourth switch section are defined as the pair of first external connection terminals,
a terminal on a connection side between the second switch section and the third switch section and a terminal of the fourth switch section on an opposite side to a connection side with the third switch section are defined as the pair of second external connection terminals,
The semiconductor power converter control unit performs lead compensation control in the current control to compensate for a phase lag of the inductor current with respect to the current command value. The chopper circuit according to any one of claims 1 to 3, wherein the AC component command value has a lead compensation component for compensating for a phase lag of the inductor current relative to the current command value. The chopper circuit of claim 4, wherein the lead compensation component is generated based on at least a proportional gain in the current control and the inductance of the inductor. The chopper circuit according to any one of claims 1 to 5, wherein the AC component command value has a fundamental wave AC component having a fundamental frequency and a tertiary wave AC component having a frequency three times the fundamental frequency. The chopper circuit according to any one of claims 1 to 5, wherein the AC command value has a trapezoidal AC component. The chopper circuit according to any one of claims 1 to 7, wherein the semiconductor power converter is composed of a chopper cell that includes two semiconductor switches connected in series and a DC capacitor connected in parallel to the two semiconductor switches, and has each terminal of one of the two semiconductor switches as an output terminal. Each of the semiconductor switches is
A semiconductor switching element that passes current in one direction when turned on;
a feedback diode connected in anti-parallel to the semiconductor switching element;
9. The chopper circuit of claim 8, comprising:

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