WO2011089483A1 - Dc to dc power converter - Google Patents

Dc to dc power converter Download PDF

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Publication number
WO2011089483A1
WO2011089483A1 PCT/IB2010/053633 IB2010053633W WO2011089483A1 WO 2011089483 A1 WO2011089483 A1 WO 2011089483A1 IB 2010053633 W IB2010053633 W IB 2010053633W WO 2011089483 A1 WO2011089483 A1 WO 2011089483A1
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Prior art keywords
dc
voltage
output
power converter
dc power
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PCT/IB2010/053633
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French (fr)
Inventor
Duraikkannan Varadarajan
Original Assignee
Duraikkannan Varadarajan
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Priority to IN00156/CHE/2010 priority Critical
Priority to IN156CH2010 priority
Application filed by Duraikkannan Varadarajan filed Critical Duraikkannan Varadarajan
Publication of WO2011089483A1 publication Critical patent/WO2011089483A1/en

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Classifications

    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00Conversion of dc power input into dc power output
    • H02M3/02Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00Conversion of dc power input into dc power output
    • H02M3/02Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/06Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider
    • H02M3/07Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider using capacitors charged and discharged alternately by semiconductor devices with control electrode, e.g. charge pumps

Abstract

A DC to DC power converter (22) and a method for boosting and bucking an input voltage (Vs) are disclosed. The DC to DC power converter includes one or more output converter stages electrically coupled to an input voltage source through an inductor (24). Each output converter stage comprises a capacitor (40, 42, 44) and at least two switches (46, 48, 50, 52) for switching at a desired duty cycle in an ON stage and an OFF stage for generating an applied voltage, such that the input voltage source (12) continuously delivers power to a load (26) and the capacitor. The one or more output stages enable a DC to DC voltage conversion to any desired step -up ratio or step -down for the output voltage. In another embodiment, output converter stages are switched such that bidirectional power flow is achieved.

Description

DC TO DC POWER CONVERTER FIELD OF INVENTION

The invention relates generally to DC to DC power converter and more specifically to circuit topologies for boosting and bucking a DC input voltage.

BACKGROUND OF THE INVENTION

A DC to DC power converter is an electronic circuit which converts a source of direct current (DC) from one voltage level to another. Typical DC power sources include batteries, solar panels, fuel cell, rectifiers and DC generators and it is often desirable to step-up or step-down the voltage from these power sources.

Battery powered systems often stack cells in series to achieve higher voltage. Stacking of cells leads to unbalanced charging and discharging of individual cells which results in early battery failure. Also, sufficient stacking of cells is not possible in many high voltage applications due to lack of space. Anyhow cells have to be connected in parallel or series to increase AH (ampere hour) capacity of the battery. Parallel connected cells produces source voltage of one cell but the abovementioned problem of unbalanced charging or discharging does not happen. A boost converter is used to step-up the voltage of such parallel connected cells. A boost converter is a DC to DC converter with an output voltage greater than the source voltage. A boost converter is sometimes called a step-up converter since it "steps up" the source voltage. Boost converters can increase the voltage and reduce the number of series connected cells. A buck converter is the reverse of the boost converter, i.e. it is used for reducing DC voltage from one level to another, and therefore is also called a "step-down" converter.

Electronic switch-mode DC to DC converters are effective converters of one DC voltage level to another, by storing the input energy temporarily and then releasing that energy to the output at a different voltage. The storage may be in either magnetic field storage components (inductors, transformers) or electric field storage components (capacitors). This conversion method is more power efficient. The typical switch-mode boost converter is shown in FIG. 1 as indicated by reference numeral 10 and it consists of two distinct states based on the state of the switch SI indicated by reference numeral 20. In the On-state, the switch SI is closed and energy is stored in the inductor L indicated by reference numeral 14, resulting in an increase in the inductor current and in the Off-state, the switch S 1 is open and the energy stored in the inductor 14 is delivered to the capacitor CI indicated by reference numeral 18 through the diode 16 and the load R indicated by reference numeral 19. During Off-state the inductor energy as well as the source energy is delivered to the load and the capacitor. During On-state source does not deliver energy to the load or capacitor. Off-state time has to be decreased to get higher output voltages. Reduced Off-state results in more power delivered by storing energy in the inductor and less power flowing directly from the source to the load. Because of this, more energy needs to be stored in the inductor by way of big inductance or more current ripple especially at high step-up voltages. This boost converter is generally not preferred for an output voltage of more than 3 to 4 times input voltage.

In the conventional boost converters, inductor size can be reduced by operating the switch at higher frequencies. But higher switching frequencies lead to more losses in the switch and the inductor resulting in poor efficiency. Also, inductor size has to be kept high even at higher switching frequencies due to hysteresis losses, eddy current losses and the losses due to proximity effect. Another aspect to be considered is the current ripple, higher inductor current ripple, which is also source current ripple in this converter, tends to reduce the battery life i.e. generally power source is a battery source. Therefore, generally the current ripple is kept low and the inductance is kept high in the conventional DC boost power converters.

Due to the above limitations of the conventional DC boost circuits, it is not practical to design DC boost circuits for high boost ratios which require huge inductance. Higher boost ratios are typically achieved by using a full bridge inverter, a transformer stage to boost voltage and a diode rectifier stage. This is more complicated and less efficient. Also, these circuits generate more EMI (electromagnetic interference) noises. It is therefore desirable to have circuit topologies for the electronic switch mode DC to DC converters that address the above disadvantages while retaining the functionality of boosting or bucking the DC voltage. SUMMARY OF THE INVENTION

According to one embodiment, a DC to DC power converter connected to an input voltage source is disclosed. The DC to DC power converter includes an inductor connected to the input voltage source. The DC to DC power converter further includes one or more output converter stages coupled to the input voltage source through the inductor, each output converter stages comprises a capacitor and at least two switches for switching at a desired duty cycle in an ON stage and an OFF stage for generating an applied voltage, wherein the one or more output stages enable a DC to DC voltage conversion to a desired output voltage. The input voltage source continuously delivers power to a load through the inductor.

According to another embodiment, a method for generating an applied voltage in the DC to DC power converter is disclosed. The method includes a boosting operation by switching the at least two switches of each output converter stages such that voltage across one capacitor appears as the applied voltage in the ON time and voltage across two or more capacitors in series appears as the applied voltage in the OFF time. A similar approach is used for a bucking operation. In another embodiment, output converter stages are switched such that bidirectional power flow is achieved.

BRIEF DESCRIPTION OF DRAWINGS

These and other features, aspects, and advantages of the present invention will become better understood when the following detailed description is read with reference to the accompanying drawings in which like characters represent like parts throughout the drawings, wherein:

FIG. 1 is a circuit diagram representation of a conventional prior art DC boost converter circuit;

FIG. 2 is a circuit diagram representation of an exemplary three level DC boost converter; FIG. 3 is a circuit diagram representation of an experimental set-up based on the DC boost converter topology of FIG. 2;

FIG. 4 is an oscilloscope output showing the switching pulses during the experimental set-up;

FIG. 5 is an oscilloscope output showing the inductor current profile during the experimental set-up;

FIG. 6 is a diagrammatic representation of an exemplary building block circuit that is used as a boost unit;

FIG. 7 is a circuit diagram representation of another exemplary DC boost power converter that steps-up the input voltage close to two times;

FIG. 8 is a circuit diagram representation of another exemplary DC power converter embodiment that is used to charge a battery source;

FIG. 9 is a circuit diagram representation for charging a battery source; and

FIG. 10 is circuit diagram representation for a bi-directional multilevel converter for battery management.

DETAILED DESCRIPTION OF INVENTION

The primary embodiment of the present invention is a DC to DC power converter comprises an inductor connected to an input voltage source; an interface circuit comprises a diode and a capacitor in series combination, said diode is connected to the inductor; plurality of output stages connected to the input voltage source through the inductor and the interface circuit, each output stage comprises plurality of series combination of a diode and a switch coupled to a capacitor, said capacitor connects two adjacent output stages or connects last output stage with the input voltage source through a diode; and a load connected across the capacitors of the output stages and the capacitor of the interface circuit, said load receives an output voltage from the output stages; wherein said switches of the output stages are switched based on predefined duty cycle for generating an applied voltage to produce predefined output voltage. In yet another embodiment of the present invention the input voltage source continuously delivers power to the load through the inductor during ON and OFF time of the duty cycle. In still another embodiment of the present invention the output stage diodes block reverse voltage. In still another embodiment of the present invention the DC to DC power converter with one stage, comprises an output stage having a plurality of switches with anti-parallel diode coupled to a capacitor.

In still another embodiment of the present invention the diodes of the output converter stage can be replaced by thyristors or GTOs (gate turn-off thyristors) to reduce voltage rating of the series connected switches.

In still another embodiment of the present invention the predefined output voltage is sum of the voltages across the capacitors of the output stages and the capacitor of the interface circuit.

In still another embodiment of the present invention the input voltage source is selected from a group comprising battery, photovoltaic cell, fuel cell, output voltage of a power converter and output of a rectifier.

In still another embodiment of the present invention the number of output stages is varied to achieve predefined output voltage.

In still another embodiment of the present invention the output stage optionally comprises a switch with anti-parallel diode in place of a diode, said parallel combination provides reverse current from the load to the voltage source.

In still another embodiment of the present invention the interface circuit optionally comprises two switches with anti -parallel diodes to enable conduction in both directions.

In still another embodiment of the present invention the DC to DC converter optionally comprises two switches with anti-parallel diodes in place of the diode connected between the input voltage source and the last output stage to enable bi-directional conduction. In still another embodiment of the present invention the output stages provides bidirectional DC power flow between the input voltage source and the load.

In still another embodiment of the present invention the current from an input source is drawn at unity power factor using the DC to DC power converter.

Another embodiment of the present invention is a method for generating predefined output voltage in a DC to DC power converter, said method comprising acts of performing switching operation selectively on switches of output stages of the DC to DC power converters at predetermined duty cycle to generate an applied voltage, said applied voltage is of one capacitor voltage during ON time and two or more capacitors voltages during OFF time, said applied voltage leads to a predefined output voltage across a load; wherein the input voltage source delivers continuous power to the load through the inductor during both ON period and OFF period of the duty cycle.

In yet another embodiment plurality of output stages coupled in series combination.

In still another embodiment the output voltage is the voltage across capacitors of the output converter stages and the capacitor of an interface circuit.

In still another embodiment power flow in the DC to DC power converter is bidirectional between the input voltage source and the load.

In still another embodiment the applied voltage in the DC to DC bidirectional power converter is of one or more capacitor voltage during the ON time and zero voltage during the OFF time for reverse power flow from the load to the voltage source.

Different embodiments described herein relate generally to electronic switch mode DC to DC power converter topologies for boosting and bucking of DC voltage. In the embodiments for the boost converter a DC source voltage may be stepped-up to any higher voltages. Energy stored in the inductor of these exemplary DC power converters is considerably less compared to the conventional boost converter leading to smaller inductor sizes. Also, energy storage in the inductor of these exemplary DC power converters does not increase proportionately with the step-up ratio. These circuit topologies can be used to boost voltages from voltage sources such as battery, photovoltaic cells etc or voltages from another power converter. Also, the same principle can be used for charging the battery if that is the voltage source, without employing any additional storage elements.

According to one embodiment of the invention FIG. 2 is a diagrammatic representation of the electric circuit topology for an exemplary DC boost converter 22. Here a three level boost converter is described wherein a maximum output voltage close to but less than three times input voltage can be obtained. Similarly higher level converters can be built. For example a four level converter can boost an input voltage to a maximum output voltage close to but less than four times input voltage. As shown in FIG. 2, Vs indicated by reference numeral 12 is an input voltage for the DC boost converter, that could be from an input voltage source like a battery or it could be the output voltage of another converter. CI, C2, C3 indicated by reference numerals 40, 42, 44 are capacitors whose total series voltage is the output voltage of the DC boost converter which would be less than three times the input voltage. Dl to D6 are diodes indicated by reference numerals 28, 30, 32, 34, 36 and 38 and SI, S2, S3, S4 indicated by reference numerals 46, 48, 50, 52 are gate controlled switches with reverse diodes. R is a resistive load indicated by reference numeral 26, L is an inductor indicated by reference numeral 24, iL represents inductor current, iO represents output (load) current. In the description of the embodiment as given below, Ts is the switching period for the switches SI , S2, S3 and S4. Switching period is the total time (ON-time + OFF-time) of one switching cycle. Duty cycle is represented by d and is the fraction of a switching period that a switch is in ON state.

In the exemplary embodiment of FIG. 2, there are six stages in one cycle of the DC boost converter operation. The following descriptions can be easily understood considering inductor current as a current source charging the capacitors CI , CI, C2, C3, C3, C2 in that order and this cycle repeats. Note that CI , C2 & C3 are charged for equal times. Stage 1 : Start-up stage: Switch S4 is in ON state and switches S3 and SI are OFF. Switch S2 is ON for the period d x Ts and OFF for the period (1-d) x Ts. When S2 is switched ON, the inductor current increases and the voltage across CI appears as Va, referred herein as applied voltage and indicated by reference numeral 13 which is less than Vs. When S2 is switched OFF the inductor energy forces the current through D5, CI , C2, D4 and S4. The combined voltage of CI & C2 appears as Va, and the inductor current decreases. The voltages across CI & C2 reach a steady value when the inductor volt-seconds balance happens. The product of inductor voltage and time for one switching cycle is zero for a steady inductor current, and at one particular output voltage the inductor volt-seconds balance occurs. Alternately, it can be said that the output voltage adjusts such that this inductor volt-seconds balance occurs. Thus, the power flows from source to CI and load during ON time and to CI , C2 and load during OFF time.

In all the stages described below when the appropriate switch is ON, one or single capacitor voltage is the applied voltage Va, and when the switch is OFF voltage of two capacitors in series is the applied voltage Va.

Stage 2: Stage 1 is repeated so that the equal energy flows from voltage source to all the capacitors in one cycle of six stages.

Stage 3: Switch SI is in ON state and S2 and S3 in OFF state. Switch S4 is ON for the period d x Ts and OFF for the period (1-d) x Ts. Power flows from source to C2 and load during ON time and to C2, C3 and load during OFF time.

A delay time referred herein as "dead time" is required before S 1 is switched ON after S2 is switched OFF to avoid both switches being ON simultaneously which results in shorting of source through the inductor leading to high currents. Maximum duty cycle depends on this dead time. This dead time requirement is taken into consideration in all the stages of the exemplary operation.

Stage 4: Switch S4 and S2 are in OFF state and SI is in ON state. Switch S3 is ON for the period d x Ts and OFF for the period (1-d) x Ts. Power flows from source to C3 and load during ON time and to C3, C2 and load during OFF time. Stage 5: Stage 4 is repeated so that the equal energy flows from source to all the capacitors in one cycle of six stages.

Stage 6: S3 is in OFF state, S4 is in ON state and S2 is in OFF state. Switch SI is switched ON for the period d x Ts and OFF for the period (1-d) x Ts. Power flows from source to C2 and load during ON time and to C2, CI and load during OFF time.

After Stage 6, Stage 1 follows and the sequence repeats. The switches are operated such that after a full cycle of six stages all the capacitors are equally charged from the source. It may be noted here that the voltage across C2 will be more because it is charged with one extra OFF time compared to CI & C3. This is easily corrected by adding more stages of stage 1 and stage 4 appropriately in some of the cycles. Many other sequences of switching are possible with the same concepts.

Steady state equation for the above operation is discussed below. It may be noted that all the switches, diodes, inductor and capacitors are assumed to be ideal. It is also assumed that the voltage across CI , C2 & C3 are same and equal to Vc. The step-up ratio can be increased or decreased by increasing or decreasing the duty cycle d. For each of the above stages following steady state volt-seconds balance equation can be written for the inductor:

(Vs - Vc) d Ts = (2 Vc - Vs) (1-d) Ts

Vc = Vs/ (2-d) d = (2Vc - Vs)/Vc

During OFF state of the above stages the inductor current falls. This change in the inductor current decides the current ripple. Thus, the current ripple is directly proportional to the OFF time and the capacitor voltages. The minimum OFF time is decided by the dead time requirement of the switches used, which is the time delay between switching OFF of one switch to switching ON of another switch respectively in the same unit. The minimum inductor current ripple that occurs at the maximum step-up, is therefore decided by the dead-time and not by the switching frequency. The switching frequency is decided based on the capacitor size which any how generally much higher than required by the operation of the boost converter due to other considerations.

In the embodiment described herein, at steady state, the voltage across the inductor is switched between (Vs - Vc) and (Vs - 2Vc) where Vc is the voltage across one capacitor which is less than the source voltage. By switching this way steady state inductor volt- seconds balance is achieved with the source delivering power always. Also this switching leads to minimum inductor current ripple at the maximum ON-time i.e. maximum step- up. The energy stored in the inductor does not increase in proportion to step-up ratio.

In situation where regulated output voltage is required, i.e. the output voltage needs to be steady, even though the current ripple may be higher as OFF time is increased to regulate the output voltage it is still less compared to conventional boost converter because the source always delivers power during ON as well as OFF intervals, while the source delivers power in the conventional boost converter only during the OFF interval.

The voltage stress on the top switch of the last stage (S3 in FIG. 2) and the bottom switch of the first stage (S2 in FIG. 2) will be maximum. The series diode along with these switches (for example D3 is series diode for S3) can be replaced with a slower switch (switches with more turn ON & turn OFF times) like a SCR (Thyristor) or GTO (Gate Turn-Off Thyristors). By using these slower switches that block high forward voltages, the voltage rating of the MOSFET (metal-oxide-semiconductor field-effect transistor) or IGBT (Insulated Gate Bipolar Thyristor), used in the place of S3 & S2 can be reduced. The inductor size will be determined by the dead time of these MOSFET/IGBT switches for unregulated maximum output voltage. SCR/GTO can be turned one cycle before S3 is to be switched ON and turned OFF in the next cycle after S3 switching cycle is over. Since these switches in series take high voltage stress MOSFET/IGBT switches of lesser voltage rating can be used for S3 & S2 in FIG. 2, in higher level converters, leading to low cost and lesser losses in the switches.

According to another aspect of the invention a switching scheme as described herein above is disclosed. More particularly, the method for switching includes providing one or more switches in the output converter stages. The switches are operated in such a manner that one capacitor voltage appear as the applied voltage in the ON time and the voltage of two or more capacitors in series appears as applied voltage during OFF time. Further, the switches are operated in cyclic manner where each capacitor is charged one after other by the input voltage source and all the capacitor voltages are kept same. The switching scheme described herein leads to minimum current ripple at the maximum duty cycle. The said switching scheme enables the power flow from input to output continuously i.e. both during ON & OFF intervals resulting in lesser current ripple in the inductor and the source.

A power converter with single output stage produces a maximum output voltage close to but less than twice the input voltage, a power converter with two stacked stages produces a maximum output voltage close to but less than three times of input voltage and so on with additional stacked stages. Also, the output voltage using the switching scheme described herein can be regulated by varying the duty cycle by closed loop control methods. The DC to DC power converter as described herein when operated at the maximum step- up i.e. maximum ON time, the size of the inductor is determined by the dead-time, i.e. by the delay time required to be given between switching OFF of one switch and switching ON of another switch connected in series, and not by the switching frequency.

The DC boost converter circuit topology of FIG. 2 is implemented as an experimental set-up as shown by the circuit diagram 23 in FIG. 3. MOSFET switches 47, 49, 51 , and 53 are used. The other elements as shown in FIG. 3 are same as shown and described in FIG. 2.

The following components were used in the experiment:

Vs - Voltage source 23V

L - 0.58mH

SI, S2, S3, S4 - MOSFETs IR640

Dl, D2, D3, D4, D5, D6 - Diodes U16C40

CI, C2, C3 - Capacitors of the rating 470uF, 40 V

Load - Resistor of the rating 100 Ohms Under operation the MOSFT switches were switched at 6KHZ with a duty cycle of 0.89 during the stages the switches were supposed to be switched ON & OFF as explained before. One additional cycle of stage 1 & stage 4 were inserted for every 4 full cycles (6 stages).

The measured outputs were as follows:

Measured output voltage = 55 V

Voltage across CI = 18.5

Voltage across CI = 18.6

Voltage across CI = 18.7

The output voltage is less than the theoretically expected value of 62V because of the voltage drops in MOSFETs and Diodes.

The switching pulses as obtained during the experiment are shown in FIG. 4 as the output from an oscilloscope indicated by reference numeral 54. The switching pulses are shown on Y axis 55 for switches SI , S2, S3 & S4 and indicated by 56, 58, 60, and 62 respectively in channel 1 , 2, 3 & 4 wherein high level is ON time and zero level is OFF time. Y axis scale for all the signals are 5Volts (V)/division. The X axis is shown by reference numeral 57 and indicates the time for the switching pulses in micro-seconds; the X axis scale is 200 microsecond/division.

The inductor current profile for the circuit of FIG.3 is shown in FIG. 5 by reference numeral 70 The Y axis is indicated by reference numeral 71 whose scale is 1 Ampere (A)/division and X axis 73 indicates the time with the scale lOOmicrosec/division. The reference numeral 72 indicates the current profile. The average current is 1.41 A and the peak to peak ripple is 0.86 A. The inductor current ripple is given by the following formula: Inductor current ripple Δί = 2 x Vc x (1-d) Ts/L

For given values the theoretical inductor current ripple Δί = 0.9 A

The current ripple for the given inductor is therefore very low which will be further reduced when the duty cycle is close to 1. A series diode with the switches shown in Fig. 2 is required if the switch has anti-parallel diode or it can not block reverse voltage. If reverse voltage blocking switches such as GTOs (Gate Turn-Off Thyristors) are used then diodes Dl, D2, D3 & D4 are not required and they can be shorted. Thus the different outputs of the experimental results validated the circuit topology as described in FIG. 2 above.

FIG. 6 shows an building block circuit 80 or output converter stage that is used as a boost unit that can be added as many times to obtain three-level, four-level or any higher level boosting of the input voltage to meet the required output voltage. As shown the boost unit includes two switches Sx and Sx+1 depicted by reference numerals 82 and 84, and a capacitor Cx depicted by reference numeral 86.

Another exemplary embodiment of the DC power converter is explained in reference with FIG. 7. The circuit diagram depicted by reference numeral 90 in FIG. 7 represents a two level boost converter that steps-up the input voltage Vs depicted by reference numeral 12, to a maximum output voltage close to two times. This exemplary embodiment uses IGBT as the switches (depicted by reference numerals 46, 48). Other circuit elements like the inductor 24, capacitors 40, 42, diodes 32, 34, and Load R depicted by reference numeral 26 have the same functionality as described in reference to FIG. 2. The switching principle is also similar to multilevel boosting as discussed in reference to FIG. 2 with added simplification. In this exemplary embodiment, SI & S2 are switched at nearly 50% duty cycle with required dead time to achieve close to twice the input voltage. By reducing the duty cycle lesser output voltages can be obtained if so desired.

The exemplary embodiments of the DC booster circuit topologies described herein provide several advantages over the conventional DC boost converter. These include reduced size of the inductor, negligible losses in the inductor, low input current ripple as the voltage source continuously delivers power both during ON & OFF intervals of all the stages, very low source current ripple, which is same as the inductor current ripple. The battery life is not deteriorated if the voltage source is a battery unlike in the conventional DC boost converters. Even with very low switching frequency, much better performance is obtained using smaller inductor, and switching losses are negligibly small. Lesser voltage switches can be used leading to lesser conduction losses. Economical MOSFETs can be used as switches. EMI (electromagnetic interference) generated by this boost converter is considerably less because of smaller inductor, lesser inductor current ripple and lesser switching voltages.

In yet another aspect of the invention, the above topologies can also be used to charge a battery source without adding any additional storage elements. An exemplary implementation depicted by reference numeral 92 for the same is shown in FIG.8. In FIG. 8, switches S5 depicted by reference numeral 94, S6 depicted by reference numeral 96 are added to the original three level boost circuit as shown in FIG. 2 and other components are same as in FIG. 2, to charge the battery source 12. Switches S5 & S6 are switched to step down the load voltage for charging the battery source. When both the switches S5 & S6 are OFF inductor current flows through D5. Another exemplary circuit topology 98 to charge a battery source using a similar switching arrangement and switching schemes as the boost topology of FIG 2 is described in FIG. 9. In this embodiment, the direction of switches SI, S2, S3 & S4 and diodes Dl , D2, D3, D4, D5 & D6 is reversed, current flows in the opposite direction compared to the DC boost converter described in Figs. 2 and 3. Switches S55 & S66 indicated by reference numeral 94 & 96, Diode D7 indicated by reference numeral 104 and capacitor C4 indicated by reference numeral 100. With this topology inductor size is smaller than that of Fig. 8 as this circuit is operated as three buck converters with lesser step-down ratio. The voltages across all the capacitors (CI, C2 & C3) are assumed to be same say Vc. In this embodiment, Vc is always more than load voltage (Vc4). Each capacitor voltage is switched one by one to obtain stepped down voltage at the load as describe below:

Stage 1 : Switches S22 & S55 are switched ON, and switches SI 1 , S33, S44 & S66 are OFF. The voltage across CI (Vc) appears as applied voltage Va indicated by reference numeral 13. Since Vc is more than load voltage (Vc4) the inductor current increases in the direction shown.

Stage 2: S22 & S55 is turned OFF. All other switches' status remains same as above. Now the inductor energy makes the current to freewheel through diode D7. Similarly C2 (by switching ON & OFF SI 1 & S44) and C3 (by switching ON & OFF S33 & S66) are switched to the load with required duty cycle.

In yet another embodiment, a bi-directional multilevel converter 106 for battery management is shown in FIG.10. In the boost mode, where the power flows from battery 12 to load 26, switches S5 & S6 (reference numeral 94 and 96) are kept ON always. SI, S2, S3 & S4 (reference numerals 46, 48, 50, 52) are switched as described in reference to FIG. 2 above for boost operation. All other switches are OFF.

In the battery charging mode (power flow is from load to battery), switches S55 (reference numeral 107) and S22 (reference numeral 110) are switched to charge the battery from CI (reference numeral 40). Switches Sl l (reference numeral 108) and S44 (reference numeral 114) are switched to charge the battery from C2 (reference numeral 42) and S33 (reference numeral 1 12) and S66 (reference numeral 1 16) are switched to charge the battery from C3 (reference numeral 44). All other switches are kept OFF. The capacitor voltages are kept same by appropriate control methods.

The Boost converter concept explained through FIG. 2 can further be extended to draw a unity power factor input current from an AC source. For this the rectified AC source should be the input source of the Boost converter. The switch S7 shown in FIG. 10 indicated by reference numeral 105 is used to inject zero applied voltage. The switch S7 can be used for FIG. 2 as is used in FIG. 10. A switching strategy, wherein zero voltage or the voltage of one or more capacitors is the applied voltage during ON time and the voltage of one or more capacitors is the applied voltage during OFF time, can be used. A higher boost ratio for a given number of output stages can be achieved using this switching strategy which can be used for unity power factor operation. A voltage controller which regulates the output voltage to a reference voltage produces a DC reference current for an inner current control loop. The current controller reference current is produced by multiplying the DC reference current with a full wave rectified waveform in phase with the input AC source voltage. The current controller controls the actual inductor current to follow the reference current by varying the ON & OFF times explained in the Boost converter operation. The different DC converter topologies as described in the invention are simpler and more efficient than the conventional DC converters, and allow for different combinations to step-up or step-down the voltages based on the end application. The maximum output voltages can be close to twice the source voltage, or alternately any output voltages higher than the source voltage can be achieved through these embodiments. Also these embodiments advantageously allow a continuous power flow from the input voltage source. Very low Inductor current ripple occur at the maximum step-up in these embodiments. Also, the inductor size does not increase proportionately with the step-up ratio. Another advantage as shown in select embodiments is that a bidirectional power flow between the input source and output is possible. The above mentioned DC converter embodiments can be used in several applications, including but not limited to hybrid electric vehicles (HEV), solar power converters, fuel cell power converters, uninterrupted power supplies, lighting systems, various portable electronic devices such as cellular phones and laptop computers etc., which are supplied with power from batteries primarily. While considerable emphasis has been placed herein on the various components of the preferred embodiment, it will be appreciated that many alterations can be made and that many modifications can be made in the preferred embodiment without departing from the principles of the invention. These and other changes in the preferred embodiment as well as other embodiments of the invention will be apparent to those skilled in the art from the disclosure herein, whereby it is to be distinctly understood that the foregoing descriptive matter is to be interpreted merely as illustrative of the invention and not as a limitation.

Claims

We claim
1. A DC to DC power converter comprises:
an inductor connected to an input voltage source;
an interface circuit comprises a diode and a capacitor in series combination, said diode is connected to the inductor;
plurality of output stages connected to the input voltage source through the inductor and the interface circuit, each output stage comprises plurality of series combination of a diode and a switch coupled to a capacitor, said capacitor connects two adjacent output stages and said capacitor of the last output stage connects to the input voltage source through a diode; and
a load connected across the capacitors of the output stages and the capacitor of the interface circuit, said load receives an output voltage from the output stages;
wherein said switches of the output stages are switched based on predefined duty cycle for generating an applied voltage to produce a predefined output voltage.
2. The DC to DC power converter of claim 1, wherein the input voltage source continuously delivers power to the load through the inductor during ON and OFF time of the duty cycle.
3. The DC to DC power converter of claim 1, wherein the output stages diodes blocks reverse voltage.
4. The DC to DC power converter of claim 1 , wherein the DC to DC power converter with one output stage, comprises said output stage having plurality of switches with anti-parallel diode coupled to a capacitor.
5. The DC to DC power converter of claim 1 , wherein the diodes of the output stages can be replaced by thyristors or GTOs (gate turn-off thyristors) to reduce voltage rating of the series connected switches.
6. The DC to DC power converter of claim 1 , wherein the predefined output voltage is sum of the voltages across the capacitors of the output stages and the capacitor of the interface circuit.
7. The DC to DC power converter of claim 1 , wherein the input voltage source is selected from a group comprising battery, photovoltaic cell, fuel cell, output voltage of a power converter and output of a rectifier.
8. The DC to DC power converter of claim 1, wherein the number of output stages are varied to achieve a predefined output voltage.
9. The DC to DC power converter of claim 1, wherein the output stages optionally comprises a switch with anti-parallel diode in place of a diode, said parallel combination allows reverse current from the load to the voltage source.
10. The DC to DC power converter of claim 1, wherein the interface circuit optionally comprises two switches with anti-parallel diodes to enable conduction in both directions.
11. The DC to DC power converter of claim 1 , wherein the DC to DC converter optionally comprises two switches with anti-parallel diodes in place of the diode connected between the input voltage source and the last output stage to enable bidirectional conduction.
12. The DC to DC power converter of claims 9 to 11 , wherein the output stages provides bidirectional DC power flow between the input voltage source and the load.
13. The DC to DC power converter of claim 1 , wherein the current from an input source is drawn at unity power factor using the DC to DC power converter.
14. A method for generating predefined output voltage in a DC to DC power converter, said method comprising acts of:
performing switching operation selectively on switches of output stages of the DC to DC power converters at a predetermined duty cycle to generate an applied voltage, said applied voltage is of one capacitor voltage during ON time and two or more capacitors voltages during OFF time, said applied voltage leads to a predefined output voltage across a load; wherein the input voltage source delivers continuous power to the load through the inductor during both ON period and OFF period of the duty cycle.
15. The method as claimed in claim 14, wherein plurality of output stages coupled in series combination.
16. The method as claimed in claim 14, wherein the output voltage is the voltage across capacitors of the output converter stages and the capacitor of an interface circuit.
17. The method as claimed in claim 14, wherein power flow in the DC to DC power converter is bidirectional between the input voltage source and the load.
18. The method as claimed in claim 17, wherein the applied voltage in the DC to DC bidirectional power converter is of one or more capacitor voltage during the ON time and zero voltage during the OFF time for reverse power flow from the load to the voltage source.
PCT/IB2010/053633 2010-01-24 2010-08-11 Dc to dc power converter WO2011089483A1 (en)

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