JP7336137B2 - power converter - Google Patents

Description

The present invention relates to power converters.

Among the power conversion devices disclosed in Patent Document 1, in particular, as [11th embodiment], the insulation type bipolar device disclosed in paragraphs 0200 to 0240 of the specification and FIGS. 31 to 36 of the drawings bidirectional DC/DC converters and isolated bidirectional DC/AC inverters.

In this power converter, the output voltage can be determined by the "phase relationship" of the waveforms driving the switches S1 and S2 and the switches S4a and S4b, and the output voltage can be freely controlled to be positive or negative.

Moreover, in this power conversion device, when a power source or the like is connected instead of a load as shown in FIG.

Such a power conversion device of Patent Document 1 and the insulated bipolar bidirectional DC/DC converter and the insulated bidirectional DC/AC inverter of the embodiments of the present invention are hereinafter referred to as "power conversion device". The power conversion device can be used as both functions of an isolated bipolar bidirectional DC/DC converter and an isolated bidirectional DC/AC inverter, or can be used as either one of them. In addition, it is not always necessary to use the motor in both directions, and it can be used in either power running or regeneration direction depending on the application.

JP 2012-210104 A

It is desirable to suppress power loss in power converters. It is desirable to suppress noise in power converters.

A power converter according to an aspect of the present invention includes a primary side circuit that outputs alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a direct current power supply. A power conversion device receives a voltage based on the output of a primary circuit, and outputs a voltage having the same polarity as the input voltage by switching the conduction state of a secondary side switch element, or outputs a voltage with the same polarity as the input voltage. A secondary side circuit for switching whether to output a voltage of reverse polarity is provided. In the power converter, whether to advance or delay the switching timing of the conduction state of the secondary side switch element with respect to the switching timing of the conduction state of the primary side switch element is determined by the output terminal of the DC power supply and the output terminal of the primary side circuit. , a control circuit for switching according to the polarity of the current at the output end of the secondary side circuit or the output end of the power converter.

The primary side circuit includes a first switch element and a second switch element connected in series between both ends of the DC power supply, and a first capacitance element and a second electrostatic capacitance element connected in series between both ends of the DC power supply. and a capacitive element. One end of the output of the primary circuit may be connected to a connection point between the first switch element and the second switch element. The other end of the output of the primary side circuit may be connected to the connection point between the first capacitance element and the second capacitance element.

The capacitance value of the first capacitance element may be substantially equal to the capacitance value of the second capacitance element.

The primary side circuit includes a first switch element and a second switch element connected in series between both ends of the DC power supply, and a capacitance element connected to one end of the DC power supply and one end of the output of the primary side circuit. , may be a half-bridge circuit comprising: The other end of the output of the primary circuit may be connected to the connection point between the first switch element and the second switch element.

The DC power supply may comprise a first DC power supply and a second DC power supply connected in series. The primary side circuit may be a half bridge circuit comprising a first switch element and a second switch element connected in series between both ends of a first DC power supply and a second DC power supply connected in series. One end of the output of the primary circuit may be connected to a connection point between the first switch element and the second switch element. The other end of the output of the primary circuit may be connected to a connection point between the first DC power supply and the second DC power supply.

The DC power supply may comprise a first DC power supply and a second DC power supply connected in series. The primary side circuit includes a first switch element and a second switch element connected in series between both ends of a first DC power supply and a second DC power supply connected in series, and a first DC power supply and a second DC power supply. and a capacitive element connected between the connection point and one end of the output of the primary side circuit. The other end of the output of the primary circuit may be connected to the connection point between the first switch element and the second switch element.

The primary side circuit includes a first switch element and a second switch element connected in series between both ends of the DC power supply, a third switch element and a fourth switch element connected in series between both ends of the DC power supply, may be a full bridge circuit comprising One end of the output of the primary circuit may be connected to a connection point between the first switch element and the second switch element. The other end of the output of the primary circuit may be connected to a connection point between the third switch element and the fourth switch element.

Between the connection point between the first switch element and the second switch element and one end of the output of the primary circuit, and between the connection point between the third switch element and the fourth switch element and the output of the primary circuit A capacitive element may be provided between at least one of the ends.

The secondary circuit may comprise a transformer comprising a primary winding to which the output of the primary circuit is input, a first secondary winding and a second secondary winding. The secondary side switch element is positioned between one end of the output of the secondary side circuit and one end of the first secondary winding, or between the other end of the output of the secondary side circuit and the other end of the first secondary winding. A first bidirectional switch element may be provided between the ends. The secondary side switch element is positioned between one end of the output of the secondary side circuit and one end of the second secondary winding, or between the other end of the output of the secondary side circuit and the other end of the second secondary winding. A second bidirectional switch element may be provided between the ends. The secondary circuit switches one end and the other end of the first secondary winding to one end of the output of the secondary circuit by switching the conductive state of the first bidirectional switch element and the second bidirectional switch element. and the other end, or connecting one end and the other end of the second secondary winding to the one end and the other end of the output of the secondary side circuit, respectively.

The secondary circuit may comprise a transformer comprising a primary winding to which the output of the primary circuit is input, a first secondary winding and a second secondary winding. The secondary side switch element includes a first unidirectional switch element provided between one end of the output of the secondary side circuit and one end of the first secondary winding; and a first diode element connected in parallel with opposite polarity. The secondary side switch elements are a second unidirectional switch element provided between the other end of the output of the secondary side circuit and the other end of the first secondary winding, and a second unidirectional switch element. and a second diode element connected in parallel with the opposite polarity to the switch element. The secondary switch element includes a third unidirectional switch element provided between one end of the output of the secondary circuit and one end of the second secondary winding, and and a third diode element connected in parallel with opposite polarity. The secondary switch elements include a fourth unidirectional switch element provided between the other end of the output of the secondary circuit and the other end of the second secondary winding, and a fourth unidirectional switch element. and a fourth diode element connected in parallel with a reverse polarity to the switch element. The first unidirectional switch element and the second unidirectional switch element may be connected in series with opposite polarities across the first secondary winding. The third unidirectional switch element and the fourth unidirectional switch element may be connected in series with opposite polarities across the second secondary winding.

The secondary circuit may include a transformer including a primary winding to which the output of the primary circuit is input, and a secondary winding. The secondary side switch element may comprise a first bidirectional switch element provided between one end of the output of the secondary side circuit and one end of the secondary winding. The secondary side switch element may comprise a second bidirectional switch element provided between the other end of the output of the secondary side circuit and the other end of the secondary winding. The secondary side switch element may comprise a third bidirectional switch element provided between one end of the output of the secondary side circuit and the other end of the secondary winding. A fourth bidirectional switch element may be provided between the other end of the output of the secondary circuit and one end of the secondary winding.

The power converter may comprise a smoothing circuit connected to the output side of the secondary circuit. The smoothing circuit may comprise an inductor element connected between one of the smoothing circuit's inputs and one of the smoothing circuit's outputs.

The smoothing circuit may comprise a capacitive element connected in parallel with the output of the smoothing circuit.

The load connected to the output of the power converter may be a resistive load.

The load connected to the output of the power converter may be a load that supplies power to the power converter.

The load connected to the output of the power converter may be a load that consumes power output from the power converter and supplies power to the power converter.

The control circuit may include a primary side control circuit that outputs to the primary side circuit a reference signal indicating the switching timing of the conduction state of the primary side switch element. The control circuit may include a determination circuit that determines the polarity of the current at the output terminal of the DC power supply, the output terminal of the primary circuit, the output terminal of the secondary circuit, or the output terminal of the power converter. The control circuit outputs a signal having an advance time difference or a delay time difference with respect to the reference signal according to the determination result of the determination circuit as a signal indicating switching timing of the conduction state of the secondary side switch element to the secondary side circuit. A secondary side control circuit may be provided.

The control circuit may include a determination circuit that determines the polarity of the current at the output terminal of the DC power supply, the output terminal of the primary circuit, the output terminal of the secondary circuit, or the output terminal of the power converter. The control circuit outputs a reference signal or a signal obtained by inverting the reference signal according to the determination result of the determination circuit to the primary side circuit as a primary side control signal for controlling the primary side switch element. Be prepared. A signal obtained by inverting a signal having a lead time difference with respect to the reference signal or a signal having a lag time difference with respect to the reference signal according to the decision result of the decision circuit is used as a signal indicating the switching timing of the conduction state of the secondary side switch element. A secondary side control circuit that outputs to the secondary side circuit may be provided.

At least part of the control circuitry may be implemented by a programmable device or processor.

The power conversion device may include a current sensor that is connected to the output side of the secondary circuit and detects current at the output end of the secondary circuit. The control circuit may switch whether to advance or delay the switching timing of the conductive state of the secondary side switch element with respect to the switching timing of the conductive state of the primary side switch element according to the polarity of the current detected by the current sensor.

The power conversion device may include a smoothing circuit connected to the output side of the secondary circuit and smoothing the output voltage of the secondary circuit. The power conversion device may include a current sensor that detects the current at the output end of the smoothing circuit. The control circuit may switch whether to advance or delay the switching timing of the conductive state of the secondary side switch element with respect to the switching timing of the conductive state of the primary side switch element according to the polarity of the current detected by the current sensor.

The power conversion device may include a smoothing circuit connected to the output side of the secondary circuit and smoothing the output voltage of the secondary circuit. The power conversion device may include a voltage sensor that detects the voltage at the output terminal of the smoothing circuit. The control circuit may switch whether to advance or delay the switching timing of the conductive state of the secondary side switch element with respect to the switching timing of the conductive state of the primary side switch element according to the polarity of the voltage detected by the voltage sensor.

The power conversion device may include a current sensor that detects current between the output terminal of the DC power supply or the output terminal of the primary side circuit and the secondary side circuit. The control circuit may switch whether to advance or delay the switching timing of the conductive state of the secondary side switch element with respect to the switching timing of the conductive state of the primary side switch element according to the polarity of the current detected by the current sensor.

The control circuit may include a primary side control circuit that generates a primary side control signal indicating switching timing of the conduction state of the primary side switch element based on the reference signal. The control circuit may include a determination circuit that determines the polarity of the external signal that indicates the output target value of the power converter. The control circuit outputs a signal having an advance time difference or a delay time difference with respect to the reference signal according to the determination result of the determination circuit as a signal indicating switching timing of the conduction state of the secondary side switch element to the secondary side circuit. A secondary side control circuit may be provided.

The power converter may comprise a low pass filter that attenuates high frequency components of the output signal of the current sensor. The control circuit may determine the polarity of the current based on the output of the low pass filter.

The power converter may comprise a low pass filter that attenuates high frequency components of the output signal of the voltage sensor. The control circuit may determine the polarity of the current based on the output of the low pass filter.

The control circuit may comprise a hysteresis comparator that determines the polarity of the current at the output of the DC power supply, the output of the primary circuit, the output of the secondary circuit, or the output of the power converter.

The control circuit may comprise a comparator that determines the polarity of the current at the output of the secondary circuit or at the output of the power converter. The control circuit may include a holding circuit that holds the output of the comparator or the inversion of the output of the comparator according to the output of the comparator at the switching timing of the conduction state of the primary side switch element. Based on the output of the holding circuit, the control circuit advances or delays the switching timing of the conduction state of the secondary side switch element with respect to the switching timing of the conduction state of the primary side switch element according to the output of the holding circuit. A secondary side control circuit that switches between

The control circuit may comprise a comparator that determines the polarity of the current at the output of the DC power supply or at the output of the primary circuit. The control circuit may comprise a hold circuit that holds the output of the comparator. Based on the output of the holding circuit, the control circuit advances or delays the switching timing of the conduction state of the secondary side switch element with respect to the switching timing of the conduction state of the primary side switch element according to the output of the holding circuit. A secondary side control circuit that switches between

The secondary circuit includes a transformer including a primary winding to which the output of the primary circuit is input, a secondary winding connected to a secondary switching element, and one or more tertiary windings. Be prepared. The secondary circuit may comprise one or more power supply circuits formed by one or more tertiary windings and one or more rectifying circuits connected to the one or more tertiary windings.

A power conversion device includes a control circuit that operates with power supplied from at least one of one or more power supply circuits, or a driver that drives at least one of a primary side switch element and a secondary side switch element. good.

A power conversion device may comprise the current sensor described above that operates on power supplied from at least one of the one or more power supply circuits.

A power conversion device may comprise the voltage sensor described above that operates on power supplied from at least one of the one or more power supply circuits.

According to the power conversion device described above, it is possible to obtain effects such as reduction in power loss and reduction in switching noise as compared with the conventional power conversion device.

BRIEF DESCRIPTION OF THE DRAWINGS It is a figure which shows the structural example of the power converter device which concerns on 1st Embodiment. It is a figure which shows the mode of operation|movement of a primary side circuit. FIG. 2 is a diagram showing an example of a circuit for providing dead time and its input/output relationship in a primary side circuit; It is a figure which shows the operation|movement in the state in which the primary side circuit and the secondary side circuit were connected. FIG. 4 is a diagram showing the relationship between a reference signal and a control signal that controls switching elements of a secondary circuit; 3 is a diagram showing the relationship between the output voltage of the primary circuit, the ON/OFF state of the switch element of the secondary circuit, and the output voltage of the secondary circuit; FIG. FIG. 5 is a diagram showing the relationship between a typical duty ratio of the output voltage waveform of the secondary side circuit and the output voltage of the secondary side circuit when the time difference _TD is leading; FIG. 5 is a diagram showing the relationship between a typical duty ratio of the output voltage waveform of the secondary side circuit and the output voltage of the secondary side circuit when the time difference _TD is a delay; FIG. 3 is a diagram showing how the output voltage of a secondary circuit is output to a load as an output voltage through a smoothing circuit; This is an example in which an inductive load such as a motor, charging means, or the like is connected to the output side of the smoothing circuit. FIG. 3 is a diagram showing the relationship between the polarity of the current flowing between the secondary circuit and the smoothing circuit and the direction of the current of the DC power supply; 4 is a diagram showing the direction of the current of the DC power supply with respect to the polarity of the current in the output of the primary circuit; FIG. FIG. 4 is a diagram showing a current path related to the direction of the current of the DC power supply when the power supply current state transitions from charging to discharging before and after the control signal of the primary circuit transitions. FIG. 4 is a diagram showing a current path related to the direction of the current of the DC power supply when the power supply current state transitions from charging to discharging before and after the control signal of the primary circuit transitions. FIG. 4 is a diagram showing a current path related to the direction of the current of the DC power supply when the power supply current state transitions from discharging to charging before and after the control signal of the primary side circuit transitions. FIG. 4 is a diagram showing a current path related to the direction of the current of the DC power supply when the power supply current state transitions from discharging to charging before and after the control signal of the primary side circuit transitions. FIG. 4 is a diagram summarizing whether the direction of the current of the DC power supply is discharging or charging. FIG. 2 is a diagram showing an equivalent circuit of a primary side circuit including an exciting inductor on the primary side of a transformer; FIG. 10 is a diagram showing an example in which the duty ratio of the control signal is other than 0.5 and the time difference is leading; FIG. 10 is a diagram showing an example of a case where the duty ratio of the control signal is other than 0.5 and the time difference is a delay; FIG. 10 is a diagram showing an example in which the duty ratios of the control signals of the primary side circuit and the secondary side circuit are different and the time difference advances; FIG. 10 is a diagram showing an example in which the duty ratios of the control signals of the primary side circuit and the secondary side circuit are different and the time difference is a delay; FIG. 10 is a diagram showing another configuration example of the control circuit according to the second embodiment; FIG. 10 is a diagram showing another configuration example of the control circuit according to the second embodiment; FIG. 10 is a diagram showing another configuration example of the control circuit according to the second embodiment; FIG. 4 is a diagram showing the relationship between the polarity of the current flowing between the secondary circuit and the smoothing circuit and the direction of the current of the DC power supply; FIG. 11 is a diagram showing an example of a configuration of a determination circuit according to a third embodiment; FIG. It is a figure which shows the state of the voltage of each part, and the current centering on a smoothing circuit. FIG. 3 is a diagram showing an example of a duty ratio of an output voltage waveform of a typical secondary circuit, an output voltage of the secondary circuit, and an example of waveforms of an output current of the secondary circuit; FIG. 10 is a diagram showing an example in which means for holding an output is added to the comparator of the determination circuit; FIG. 11 is a diagram showing an example of a configuration in which a low-pass filter is arranged between a current sensor and a determination circuit according to a fourth embodiment; FIG. 4 is a diagram showing an example in which a current sensor is configured to detect an output current of a smoothing circuit; FIG. 10 is a diagram showing an example in which a voltage sensor is configured at the output of a load; 4 is a diagram showing an example in which a current sensor is configured at the output end of a DC power supply 8; FIG. 3 is a diagram showing a configuration example of a determination circuit in an example in which a current sensor is configured at an output terminal of a DC power supply 8; FIG. It is a figure which shows the example which comprised the current sensor between the primary side circuit and the secondary side circuit. FIG. 4 is a diagram showing a configuration example of a determination circuit in an example in which a current sensor is configured between a primary side circuit and a secondary side circuit; It is a figure which shows the example of a half bridge type as a structure of a primary side circuit which concerns on 5th Embodiment. It is a figure which shows the example of a full bridge type as a structure of a primary side circuit. It is a figure which shows the example of a full bridge type as a structure of a primary side circuit. FIG. 11 is a diagram showing a specific example of a bidirectional switch element according to the sixth embodiment; FIG. 12 is a diagram showing an example of a secondary side circuit configured by bidirectional switch elements according to a seventh embodiment; It is a figure which shows the example of the secondary side circuit comprised by a switch element. It is a figure which shows the example of the secondary side circuit comprised by a switch element. It is a figure which shows the example of the secondary side circuit comprised by a switch element. It is a figure which shows the example of the secondary side circuit comprised by a switch element. FIG. 2 is a diagram showing an example of a secondary side circuit in which the secondary side of a transformer T is configured with a single turn; FIG. 22 is a diagram illustrating an example of an embodiment in which an insulated DC power supply is added according to the eighth embodiment; FIG. 10 is a diagram showing an example of actual measurement of the output current and the output voltage of the primary side circuit when the time difference _TD is advanced; FIG. 10 is a diagram showing an example of actual measurement of the output current and the output voltage of the primary side circuit when the time difference _TD is set as a delay;

[First Embodiment] (Description of the whole based on a basic configuration example)
In the first embodiment, the on/off control of the switching element of the secondary side circuit with respect to the on/off control of the switching element of the primary side circuit is performed with either of the two types of time differences, delay or advance. 1 is an example of a power converter that switches based on the direction of current flow within the power converter. Such a configuration has the effect of reducing power loss and switching noise.

1 to 16B show an example of a power converter according to the first embodiment.

FIG. 1 shows an example of the overall configuration of a power converter 100 according to the first embodiment. FIG. 2 shows the operation of the primary circuit 10, and FIG. 3 shows an example of a dead time generating circuit 110 for providing dead time in the primary circuit 10 and its input/output relationship. there is FIG. 4 shows how the primary side circuit 10 and the secondary side circuit 20 are connected.

FIG. 5 shows the relationship between the reference signal and the control signal that controls the switch element of the secondary circuit 20. As shown in FIG. FIG. 6 shows the relationship between the output voltage of the primary side circuit 10, the on/off state of the switch element of the secondary side circuit 20, and the output voltage of the secondary side circuit 20. As shown in FIG. 7A and 7B show the relationship between the typical duty ratio of the output voltage waveform of the secondary circuit 20 and the output voltage V2 of the secondary circuit 20. FIG.

FIG. 8 shows how the output voltage of the secondary circuit 20 is output to the load 9 through the smoothing circuit 30 as the output voltage. FIG. 9 shows an example in which an inductive load such as a motor and charging means are connected to the output side of the smoothing circuit 30 .

FIG. 10 shows the relationship between the polarity of the current flowing between the secondary circuit 20 and the smoothing circuit 30 and the direction of the current of the DC power supply 8 . FIG. 11 shows the direction of the current in the DC power supply 8 with respect to the polarity of the current in the output of the primary circuit 10 . 12A, 12B, 12C, and 12D show current paths related to the direction of the current in the DC power supply 8 before and after the control signal of the primary circuit 10 transitions. FIG. 13 shows whether the direction of the current of the DC power supply 8 is discharging or charging.

FIG. 14 shows an equivalent circuit of the primary side circuit 10 including the exciting inductor on the primary side of the transformer. FIGS. 15A and 15B show examples in which the duty ratio of the control signal is other than 0.5 and the time difference _TD is leading and lagging. 16A and 16B show cases where the duty ratios of the control signals of the primary side circuit 10 and the secondary side circuit 20 are different and the time difference is leading and lagging.

FIG. 1 shows an example of the configuration of a power converter 100 according to the first embodiment, comprising a DC power supply 8, a primary side circuit 10, a secondary side circuit 20, a smoothing circuit 30, a load 9 and a control circuit 40. showing. An outline of each circuit is described below.

The primary side circuit 10 receives a DC power supply 8 having a potential difference of _VDD between both ends thereof, and the voltage of the DC power supply 8 is applied to two switch elements S1Hi, which are two switch elements connected in series to both poles of the DC power supply 8. and S1Lo are turned on/off to output a rectangular wave voltage with a duty ratio of _D0 . Note that the duty ratio _D0 of the reference signal is generally set to about 0.5 because it is transmitted as an AC voltage by a transformer T, which will be described later. However, if the value that can be taken is at least greater than 0 and less than 1, since an AC component is included, the DC voltage is not continuously applied to the input until the core of the transformer T is magnetically saturated. Operation is possible under In the following description, for the sake of simplicity, the duty ratio D0 is assumed to be _0.5 unless otherwise considered separately. However, as will be described later, D ₀ is not limited to 0.5.

A diode element D1Hi is connected in parallel with the switching element S1Hi, and a diode element D1Lo is connected in parallel with the switching element S1Lo as shown in the primary side circuit 10 of FIG. is one end of the output of Here, the potential of one end of the output of the primary side circuit 10 when the negative electrode side of the DC power supply 8 is taken as a reference potential is V1Hi. In addition, a capacitance element 16a with a capacitance value of C1Hi and a capacitance element 16b with a capacitance value of C1Lo are also arranged in series across the DC power supply 8, and the two capacitance elements 16a and 16b The other end of the voltage output of the primary circuit 10 is the portion where the capacitance elements 16b are connected to each other. The potential of this portion with the negative electrode side of the DC power supply 8 as the reference potential is V1Lo. The potential of V1Hi with V1Lo as the reference potential is set to V1.

The two switch elements S1Hi and S1Lo of the primary side circuit 10 have terminals for inputting a control signal for switching on/off, and the on/off state is changed by the control signal from the primary side driver Drv _S1 . controlled. The primary side driver Drv _S1 is a circuit that generates control signals for two switch elements S1Hi and S1Lo between different potentials according to the primary side control signal Sig _S1 . ON/OFF control of these switch elements is performed via the primary side driver Drv _S1 based on the control signal Sig _S1 generated by the control circuit 40 .

A current flowing from the primary side circuit 10 to the secondary side circuit 20 is assumed to be i1 as shown in FIG.

The secondary side circuit 20 receives the output voltage V1 of the primary side circuit 10, is insulated through the transformer T, and outputs two voltages of either the same sign or the inverted sign by turning on/off the switch elements S2Hi and S2Lo. This is the portion that is output as the output voltage V2 of the secondary circuit 20 . For convenience of explanation, the secondary side circuit 20 includes the primary side of the transformer T as well. Control of the switch elements of the secondary circuit 20 is performed via the secondary driver Drv _S2 based on the control signal Sig _S2 generated by the control circuit 40 . For convenience, the potentials of the two output terminals of the secondary circuit 20 are V2Hi and V2Lo as shown in FIG. Note that the reference potentials for V2Hi and V2Lo are determined on a case-by-case basis, and no particular portion is determined as the reference potential.

Also, let the current flowing from the secondary circuit 20 to the smoothing circuit 30 be i2 as shown in FIG.

The smoothing circuit 30 is composed of an inductor element 31 and an electrostatic capacitance element 32 arranged as necessary, receives the output voltage V2 of the secondary circuit 20, and converts the output voltage V2 of the secondary circuit 20 into a high frequency signal. This is a circuit that removes the component and obtains an output voltage V3, which is a DC voltage component, as an output. The capacitance element 32 of the smoothing circuit 30 can be omitted. In order to indicate this, in the drawings showing the smoothing circuit 30, the electrostatic capacitive element 32 is marked with [ ]. For convenience, the potentials of the two output terminals of the smoothing circuit 30 are defined as V3Hi and V3Lo as shown in FIG. Note that the reference potentials for V3Hi and V3Lo are determined on a case-by-case basis, and no particular portion is determined as the reference potential.

Also, let the current flowing from the smoothing circuit 30 to the load 9 be i3 as shown in FIG.

The control circuit 40 generates a control signal Sig _S1 for controlling the switching element of the primary side circuit 10 and a control signal Sig _S2 for controlling the switching element of the secondary side circuit 20 according to the output target value Vi. This is the circuit that controls the circuit.

As an example in which the inside of the control circuit 40 is divided into three circuits, (A) a primary side control circuit 41, (B) a secondary side control circuit 42, and (C) a determination circuit 43 will be described. It should be noted that the example of division into these three circuits is for the convenience of simple description, and does not limit the division position of each circuit or the division into three circuits.

(A) Based on the reference signal, the primary side control circuit 41 generates a control signal Sig _S1 for controlling the ON/OFF of the switch element of the primary side circuit 10, and sends the signal to the primary side driver Drv _S1 . This is a circuit that outputs Here, the reference signal is a logic signal having a period _Tp and a duty ratio _D0 . The lower limit of the period _Tp of the reference signal is mainly determined according to the response time required for on/off switching of the switching element used. In a switching power supply device including the power conversion device 100, the cycle _Tp is generally used in a range of several hundred microseconds to several hundred ns, and expressed in frequency in a range of several kHz to several MHz.

In addition, _D0 is a duty ratio, and it is necessary to switch the logic level within the period _Tp of the reference signal for operation, and is at least greater than 0 and less than 1. In actual use, the duty ratio D ₀ is, for example, 0.5, and the description in the example of the present embodiment is made with D ₀ =0.5. It is not limited to ₀ = 0.5.

This embodiment shows an example in which the reference signal is directly used as the control signal Sig _S1 that is the output of the primary side control circuit 41 .

(B) The secondary side control circuit 42 is a circuit that generates a control signal Sig _S2 for controlling on/off of the switch element of the secondary side circuit 20 and outputs it to the secondary side driver Drv _S2 . .

First, there is an output target value Vi as an input. This output target value Vi is a signal for controlling the output voltage value V3 of the smoothing circuit 30 in the present embodiment. As an example, a signal that is proportional to V3, and the proportionality factor according to this example is A ₀ , then V3=A ₀ ·Vi during steady state operation. For the sake of simplicity, the target output value Vi will be described as a signal proportional to the output voltage V3 of the smoothing circuit 30. FIG. However, it is also possible to design the relationship between Vi and V3, for example, by giving them a linear function relationship with a constant term, or a more generalized functional relationship with a one-to-one correspondence.

Next, a triangular wave is generated in the secondary-side control circuit 42, and the comparators COMP1 and COMP2 output the triangular wave, the output target value Vi, and the signal -Vi that is the sign of Vi inverted by the inverting circuit AMP1, respectively. It compares and outputs the judgment result of magnitude relation.

The triangular wave generating means 108 is means for generating a waveform having upper and lower amplitude limits at the rising and falling portions of the reference signal, synchronized with the same period _Tp as that of the reference signal. Synchronization is performed so that the rising edge of the reference signal and the upper limit of the triangular wave, and the falling edge of the reference signal and the lower limit of the triangular wave are substantially coincident, and are described between the reference signal and the triangular wave in the secondary side control circuit 42 of FIG. The term "synchronization" expresses such a relationship.

In such a synchronous relationship, the output of the comparator COMP1 becomes a logic signal that rises in the advance direction with a time lag corresponding to the output target value Vi with respect to the rise of the reference signal and falls in the lag direction. Let the time difference here be _TD . The output of the comparator COMP2 is a logic signal that rises in the leading direction and falls in the lagging direction with a time difference _TD corresponding to the output target value Vi with respect to the falling edge of the reference signal.

The output of the comparator COMP1 and the output of the comparator COMP2, which have these relationships, are input to the leading signal generation circuit 101 and the lagging signal generation circuit 102 arranged at the subsequent stage.

The lead signal generation circuit 101 and the lag signal generation circuit 102 are circuits that set or reset the logic signal of the output X according to the rise or fall of the outputs of the comparators COMP1 and COMP2, respectively.

A logic circuit whose output transitions according to the rise and fall of a logic signal can be designed by combining basic logic circuits such as RS flip-flops, and details of the specific configuration are omitted.

The output Sig _S2lead of the lead signal generation circuit 101 and the output Sig _S2lag of the delay signal generation circuit 102 are input to the switching means SW1, one of which is selected according to the determination signal output from the determination circuit 43, and the secondary It is output as the side control signal Sig _S2 .

(C) As the judgment circuit 43, an example configured by a comparator COMP3 is shown. A signal Sign corresponding to the current i2 detected by the current sensor CT arranged at the output of the secondary circuit 20 is inputted to one input of the comparator COMP3. Then, it compares the magnitude relationship with the signal Sign with respect to the reference potential, and outputs it as the determination signal Sel. This determination signal Sel is output as a logic signal associated with the direction of the current i2, for example, + if the signal Sign corresponding to the current i2 is positive, and - if it is negative. The determination signal Sel switches the secondary side control signal Sig _S2 selected as an output by the switching means SW1 of the secondary side control circuit 42 . The switching means SW1 can be configured by, for example, a combination of a selector operating at a logic level, a logic circuit, or an analog switch. Details of specific configurations are omitted.

Although an example of the configuration handling logic signals has been given here, at least part of the parts of the configuration handling logic signals may be, for example, PLD (Programmable Logic Device), FPGA (Field Programmable Gate Array), CPU (Central Processing Unit) or other programmable device or processor, and can be realized by a program written in HDL (Hardware Description Language) or various computer languages.

A control signal Sig _S1 that controls the switch element of the primary circuit 10 and a control signal Sig _S2 that controls the switch element of the secondary circuit 20 are two control signals generated and output by the control circuit 40 . These two control signals are also synchronous with the period _Tp of the reference signal, have the same period _Tp , and have a duty ratio _D0 .

In the configuration example shown in FIG. 1, the reference signal is output as the control signal Sig _S1 . On the other hand, the control signal Sig _S2 is output as a signal having rise and fall with a time difference _TD which is either advanced or delayed with respect to the rise and fall of the reference signal. The time difference _TD is controlled to have a value corresponding to the output target value Vi. A negative value is not used for this time difference _TD , and when consideration in the direction of the time axis is required, it is expressed as lead or lag. On that premise, the lower limit of the value of T _D is 0, and the upper limit is either D ₀ ·T _p or (1−D ₀ )·T _p , whichever is smaller, which is meaningful in terms of operation. The upper limit of this time difference _TD is defined as _TDMax . For example, if D ₀ =0.5, T _DMax , which is the lesser of D ₀ ·T _p or (1−D ₀ )·T _p , is 0.5·T _p .

If a DC voltage is given to the output target value Vi and the output voltage is controlled to be constant, it will operate as a DC/DC converter. Also, if an AC voltage is applied to Vi to produce an output voltage that varies with time, for example, several times or more with respect to the period _Tp of the reference signal, it operates as a DC/AC inverter.

For example, the DC/AC inverter outputs a sine wave having a frequency of 50 Hz or 60 Hz as the output voltage V3 of the smoothing circuit 30 with respect to the period T _p =100 μs (frequency 10 kHz) of the reference signal, thereby simulating the commercial power supply. It has uses as a device.

However, these times and frequencies are only examples, and do not limit the application or range of use of the power conversion device 100 .

As shown in FIG. 1, when a power-consuming load such as a resistor or motor powering operation is connected to the output side of the smoothing circuit 30, the output current i3 is in the same direction as the output voltage V3. (A direction in which the higher potential of V3Hi and V3Lo flows, powering when viewed from the power conversion device 100).

The power conversion device 100 also smoothes the output current i3 of the smoothing circuit 30 when the load 9 is a regenerative operation state of an inductive load such as a motor, a charging device that charges a storage battery, a solar inverter, or the like. A bi-directional power conversion device that can also be operated in the direction in which the output voltage V3 of the circuit 30 is supplied (the direction in which the higher potential of V3Hi and V3Lo is applied, and the power conversion device 100 is regenerated). is.

Furthermore, the present embodiment can include an operation of connecting a commercial AC power supply as the load 9 and charging a storage battery as the DC power supply 8 (regenerative operation of the DC/AC inverter).

FIG. 2 shows how the primary side circuit 10 operates.

FIG. 2A shows a state in which the switching element S1Hi is turned on and S1Lo is turned off, and FIG. 2B shows a state in which the switching element S1Hi is turned off and S1Lo is turned on.

In this specification and the accompanying drawings, in order to distinguish from Hi and Lo used for switch elements S1Hi, S1Lo, etc., two states of logic signals generally referred to as Hi and Lo are referred to as + and - are used to represent. Also, when each of the switch elements is used, the switch element is used in a state in which the diode element is provided in parallel in the direction shown in the figure. This switch element is a switch that can be turned on/off with respect to the current in the opposite direction to the diode element provided in parallel. On the other hand, it is not always necessary to control the on/off of the current in the forward direction for the diode element provided in parallel. However, in such a case, on/off control may be possible. In other words, the switch element used in the example of the configuration of the primary side circuit 10 must be able to flow in the desired direction of current when it is turned on. However, it is not necessary to be able to block the current in the opposite direction.

Such a switch element is represented by the switch symbol in FIG. 2, and the direction of the diode element is also aligned with the direction shown in FIG. That is, it is the direction in which the side of the switch symbol in FIG.

For example, the configuration of the switch element includes a configuration example in which a unidirectional switch element such as a MOS-FET, J-FET, bipolar transistor, IGBT, SiC device, or GaN transistor is provided in parallel with a diode element. Also, P-channel elements and PNP elements can be used instead of N-channel elements and NPN elements, or they can be used together.

Furthermore, the diode element provided in parallel with the switch element includes a diode built in the switch element itself and a parasitic diode, and in this case, it is not necessary to separately attach the diode element. By using such a switch element, the mounting area and cost can be reduced.

FIG. 2C is a diagram showing a case where the control signal Sig _S1 alternately transitions between + and -. 2 shows how a square-wave output voltage of ±V _DD /2 is obtained as the output voltage V1 (=V1Hi−V1Lo) of .

Here, the term "rectangular wave" is also used as a term to express a waveform having a step that is output by turning on/off a switch element, and in this specification, not only an ideal rectangular wave but also a broader term We will use For example, a square wave includes a waveform having overshoot or ringing at the rising or falling portion of the waveform due to parasitic capacitance components, parasitic inductor components, etc. in the circuit. Also, in an ideal rectangular wave, a rectangular wave includes a waveform that has a slope caused by the supply or consumption of energy to a capacitance element or an inductor element in the flat part with a constant amplitude. It is assumed that

Although the transformer T including the primary side of the transformer T is summarized as the secondary side circuit 20, it is not essential that the transformer T is not included in the primary side circuit 10. ing. In terms of circuit configuration, it is of course possible to dispose the transformer T on the primary side, and as long as the signal flow can be maintained, it is arbitrary from where to where to include each circuit.

The configuration example of FIG. 1 is an example in which the reference signal is used as it is as Sig _S1 , and the reference signal is a logic signal with a period _Tp and a duty ratio _D0 . Therefore, in this case, the output voltage V1 of the primary side circuit 10 shown in FIG. 2(C) becomes a rectangular wave with a period of T _p and a duty ratio of D ₀ .

The primary side circuit 10 is a circuit that outputs a rectangular wave voltage with a duty ratio of _D0 from a DC power supply 8 whose DC voltage is _VDD . First, regarding the primary side circuit 10, the operation of obtaining such a rectangular wave voltage will be described. For the sake of simplicity, the duty ratio D ₀ is assumed to be 0.5 here, and the case of D ₀ ≠0.5 will be described later.

The two switch elements S1Hi and S1Lo of the primary circuit 10 are arranged in series with both poles of the DC power supply 8, and by turning on one of them, the voltage of V1Hi changes to either the positive or negative terminal voltage of the DC power supply 8. are connected so that

In particular, when the duty ratio _D0 is 0.5, V1Lo can be approximated to a state in which the voltage V _DD of the DC power supply 8 is divided by 1/2.

In the example of the first embodiment, in order to simplify the explanation, the case where the capacitance values of the capacitance elements 16a and 16b are C1Hi=C1Lo is described. Note that this embodiment is not limited to the case where the capacitance values of the capacitance elements 16a and 16b are C1Hi=C1Lo, and a configuration example when C1Hi≠C1Lo will be described later. A fifth embodiment will be described.

The primary side driver Drv _S1 is a switch element driver, and based on the logic signal Sig _S1 from the control circuit 40, outputs signals for controlling the on/off of the switch elements of the primary side circuit 10 at different potentials. Generate.

When the logic signal Sig _S1 generated by the control circuit 40 is +, the primary side driver Drv _S1 turns on the switch element S1Hi by the control signal Sig _S1Hi of the switch element S1Hi, and the control signal Sig _S1Lo of the switch element S1Lo. to turn off the switch element S1Lo.

When the logic signal Sig _S1 is negative, the primary side driver Drv _S1 turns off the switch element S1Hi by the control signal Sig _S1Hi of the switch element S1Hi, and turns off the switch element _{S1Lo by the control signal Sig S1Lo} of the switch element S1Lo. control on.

Since the two switch elements S1Hi and S1Lo are connected in series with the DC power supply 8, when both are turned on at the same time, both poles of the DC power supply 8 are short-circuited and a through current is generated through the two switch elements S1Hi and S1Lo. It may cause damage to switching elements.

For example, in an actual element such as a semiconductor switch, a shoot-through current may occur because the time required for the ON/OFF transition of the switching element cannot be zero. In order to avoid this, when the on/off state of the two switch elements is changed, a period called dead time is provided in which the two switch elements are once turned off. As a result, control is performed such that one is turned off before the other is turned on so that a through current does not flow.

FIG. 3 shows an example of a dead time generation circuit 110 for providing dead time to the control signal of the switch element of the primary side circuit 10. As shown in FIG. AND1 and AND2 are 2-input AND gates, and INV1 is an inverting gate. The rise of Sig _S1Hi is delayed with respect to Sig _S1 by a time corresponding to the time constant of the resistance element 111 with the resistance value R1 and the capacitance element 112 with the capacitance value C1. Similarly, the fall of Sig _S1Hi is delayed with respect to Sig _S1 by a time corresponding to resistance element 113 having resistance value R2 and capacitance element 114 having capacitance value C2. The state of the dead time is shown in the portion surrounded by the square dotted line in FIG.

With such a configuration, Sig _S1Hi has a dead time in the rising portion and falling portion with Sig _S1 , and in other sections, Sig _S1Hi operates according to the same logic as Sig _S1 , and Sig _S1Lo operates according to the same logic as Sig S1. It operates with logic that is the inverse of _S1 .

The illustrated dead time generation circuit 110 is a circuit that utilizes a time constant by a resistance element and a capacitance element. It is also possible to manage the rise and fall of the signal and provide a dead time.

Let _D1 denote the duty ratio of Sig _S1Hi and Sig _S1Lo .

During the dead time, although there is a period during which Sig _S1Hi and Sig _S1Lo are turned off at the same time, the state of the circuit is almost the same as one of the switching elements being turned on by the diode element D1Hi or D1Lo becoming conductive. Therefore, it is more appropriate to consider that the duty ratio of the output voltage V1 of the primary side circuit 10 is _D0 instead of D1.

Note that if the value of _D0 is small with respect to the time constant of the dead time generation circuit 110 and the time required for the ON/OFF transition of the switch element used, the output pulse of the square wave of the output voltage V2 may disappear. . Therefore, it is necessary to select _D0 and the time constant so that the required output voltage V1 is output. For example, D ₀ ·T _p and (1−D ₀ )·T _p should be set to be larger than the time constant of the dead time generation circuit 110 . The upper and lower limits of _D0 in an actual circuit are the threshold value and noise margin on the input side of the logic circuit, the resistance values of the resistance elements 111 and 113 that determine the time constant, and the capacitance elements 112 and 113. It is also influenced by variations in the capacitance value of 114 and the like. Therefore, when _D0 is used at a value as close as possible to the upper and lower limits, the resistance values of the resistance elements 111 and 113 and the capacitance element 112 and static The capacitance value of the capacitance element 114 may be adjusted.

Further, if the dead time is long, the time during which the diode element D1Hi or D1Lo is conductive becomes long, and the power loss due to the voltage generated in the conductive diode element becomes long. Therefore, it is desirable that the time is as short as possible, longer than the time required for the switch element to be used to transition from off to on or from on to off, because this will reduce the power loss due to the conductive diode element.

For convenience of explanation, the dead time generation circuit 110 is provided in the primary side driver Drv _S1 , but it may be provided on the control circuit 40 side. Further, the primary side driver Drv _S1 itself may be arranged on the control circuit 40 side, and two control signals for the switch elements S1Hi and S1Lo may be output from the control circuit 40 to the primary side circuit 10 . Similarly, the secondary side driver Drv _S2 may be arranged on the control circuit 40 side.

In order to keep the description as concise as possible without detracting from the essence of the embodiments, Sig _S1Hi is in principle the same logic as Sig _S1 , except when dead-time specific conditions are considered. , and Sig _S1Lo is treated as operating in the inverse logic of Sig _S1 .

FIG. 4 shows that the output voltage V1 of the primary side circuit 10, which is the rectangular voltage waveform generated in FIG. It shows how it is output as the output voltage V2 of the circuit 20. FIG.

Although the secondary circuit 20 also includes the primary side of the transformer T, the secondary circuit 20 including the primary side of the transformer T will be called the secondary circuit 20 for the sake of convenience.

The switch elements S2Hi and S2Lo are switch elements that allow the same flow of current in both directions when they are on, and have the characteristic of being able to cut off the current in both directions in the same way when they are off. It is a switch element. Such a switch element is distinguished from the switch element described in the primary side circuit 10, and is hereinafter referred to as a "bidirectional switch element". In the configuration example of the first embodiment, the bidirectional switch elements S2Hi and S2Lo are different from the switch elements S1Hi and S1Lo of the primary side circuit 10, and are diode elements that allow current to flow in one direction. Do not set in parallel. On the other hand, for the switch elements S1Hi and S1Lo used in the primary circuit 10, bidirectional switch elements may be used on the premise that diode elements are attached in parallel in the illustrated direction.

In the drawings attached to this specification, the bidirectional switch elements are represented by the symbols used as S2Hi and S2Lo switch elements in FIG.

The transformer T in this example is a transformer in which the number of turns of the primary side winding w1 is N1, and the number of turns of the two windings w21 and w22 on the secondary side is both N2.

The secondary side circuit 20 has a secondary side driver Drv _S2 that performs on/off control of switch elements at different potentials, and one of the bidirectional switch elements S2Hi and S2Lo is turned off by the logic signal Sig _S2 . is controlled so that the other is turned on when .

The output voltage V1 of the primary side circuit 10 is input to the primary side of the transformer T. As shown in FIG.

On the secondary side of the transformer T, two windings w21 and w22 are connected in series, and the connection point is one end of the output as the output V2Lo. The output voltage V2 of the secondary circuit 20, which is the potential of V2Hi viewed from V2Lo, is as follows depending on which of the bidirectional switch elements S2Hi and S2Lo is turned on. where m=N2/N1.
・Turn on only the bidirectional switch element S2Hi: V2=mV1 (Fig. 4(A))
・Turn on only the bidirectional switch element S2Lo: V2=-mV1 (Fig. 4(B))
In particular, if N1=N2, then m=1, and
・Turn on only the bidirectional switch element S2Hi: V2=V1 (Fig. 4(A))
・Turn on only the bidirectional switch element S2Lo: V2=-V1 (Fig. 4(B))
becomes.

That is, one of the two bidirectional switch elements S2Hi and S2Lo of the secondary circuit 20 is turned on and the other is turned off. By doing so, the voltage obtained by multiplying the output voltage V1 of the primary side circuit 10 by m is output as the output voltage V2 of the secondary side circuit 20, or the output voltage of the secondary side circuit 20 is output by inverting the sign. Controls whether to output as V2.

The effects of the number of turns N1 of the winding w1 on the primary side of the transformer T and the number of turns N2 of the windings w21 and w22 on the secondary side of the transformer T appear only in the following two points. The two points are the ratio between the output voltage V1 of the primary circuit 10 and the output voltage V2 of the secondary circuit 20 (V2=mV1, etc.), and the output current i1 of the primary circuit 10 and the secondary It is a ratio of the output current i2 of the side circuit 20 (i2=i1/m, etc.).

Furthermore, it is also possible to set the number of turns of the windings w21 and w22 on the secondary side to N21 and N22, respectively, so that N21≠N22. In this case, the output voltage V2 of the secondary side circuit 20, which is the potential of V2Hi viewed from V2Lo, is as follows depending on which of the bidirectional switch elements S2Hi and S2Lo is turned on. Here, m1=N21/N1 and m2=N22/N1.
・Turn on only the bidirectional switch element S2Hi: V2=m1・V1
・Turn on only the bidirectional switch element S2Lo: V2=-m2・V1

In the following, for the sake of simplicity, the case of N1=N2 and N21=N22=N2 will be described. However, even if N1≠N2 or N21≠N22, the essence of the present embodiment does not change, and it is possible to design a transformer T where N1≠N2 or N21≠N22.

On/off control of the bidirectional switch element of the secondary circuit 20 is also performed by the logic signal Sig _S2 synchronized with the reference signal and having a duty ratio of 0.5. However, this Sig _S2 is a logic signal having a time difference _TD with respect to the rise of the reference signal in accordance with the output target value Vi.

This time difference _TD can be provided in two ways, one in the leading direction and the other in the lagging direction, with respect to the rise of the reference signal. _In this _{embodiment ,} the lead signal generation circuit 101 shown in FIG. Generating Sig _S2lag . The control signals Sig _S2lead and Sig _S2lag are used by being switched by the switching means SW1.

FIG. 5 shows the relationship between the reference signal in FIG. 1, the secondary-side switch control signal Sig _S2lead generated by the lead signal generation circuit 101, and the secondary-side switch control signal Sig _S2lag generated by the lag signal generation circuit . showing.

The secondary side control circuit 42 in the control circuit 40 in FIG. 1 and the _triangular wave shown as a waveform in FIG. has upper and lower amplitude limits.

The voltage value of this triangular wave and the output target value Vi are compared by the comparator COMP1. Also, the voltage value of this triangular wave and -Vi obtained by inverting the output target value Vi by AMP1 are compared by the comparator COMP2.

The upper and lower amplitude limits of the triangular wave correspond to the rising and falling portions of the reference signal, respectively. Therefore, the output of the comparator COMP1 becomes a logic signal that rises in the lead direction and falls in the lag direction with a time difference corresponding to the output target value Vi with respect to the rise of the reference signal. Let the time difference here be _TD . The output of the comparator COMP2 is a logic signal that rises in the leading direction and falls in the lagging direction with a time difference _TD corresponding to the output target value Vi with respect to the falling edge of the reference signal.

Therefore, the logic signal Sig _S2lead having the lead time difference _TD with respect to the reference signal may be a logic signal having + and - under the following conditions.

・From the rise of the output of the comparator COMP1 to the rise of the output of the comparator COMP2 is positive.
・From the rise of the output of the comparator COMP2 to the rise of the output of the comparator COMP1 is negative.
Similarly, the logic signal Sig _S2lag having a delay time difference _TD with respect to the reference signal may be a logic signal having + and - under the following conditions.
• From the fall of the output of the comparator COMP1 to the fall of the output of the comparator COMP2 is positive.
・From the fall of the output of the comparator COMP2 to the fall of the output of the comparator COMP1 is negative.

Here, an example of generating a logic signal that leads or lags the reference signal by a time difference _TD has been given, but other implementation means include the following configuration.

For example, using programmable devices and processors such as PLD (Programmable Logic Device), FPGA (Field Programmable Gate Array) and CPU (Central Processing Unit), the time difference _TD is calculated for the output target value Vi as digital numerical data. Calculated by A logic signal having a desired time difference with respect to the reference signal can also be generated by means such as counting clock signals with respect to the reference signal according to the calculated numerical value.

Further, since the reference signal is a periodic signal with a period of T _p , a signal equivalent to the lead can be obtained even in the form of a delay of T _p −T _D . Since the duty ratio D ₀ of the reference signal is 0.5, it rises by counting the time of (T _p /2)−T _D from the fall of the reference signal, and falls after T _p /2. An equivalent signal can also be obtained in the form of

FIGS. 6A and 6B show how the two control signals Sig _S2lead and Sig _S2lag thus obtained control the bidirectional switch elements S2Hi and S2Lo of the secondary circuit 20. . More specifically, when V1 has a maximum value of +V _DD /2 and a minimum value of -V _DD /2 (corresponding to m=1), the bidirectional switch element of the secondary circuit 20 is turned on. /OFF state and the waveform of the output voltage V2 of the secondary side circuit 20 are shown.

_{Here , the duty} ratio _D Define _V as follows.
_Dv = 1 - T _D /(T _p /2)
= 1-2 T _D /T _p
... formula (1)

At this time, regardless of which of the two control signals Sig _S2lead and Sig _S2lag is used as the control signal Sig _S2 for control, the output voltage V2 of the secondary side circuit 20 has an amplitude of ±V _DD /2 and 1/1 of the reference signal. It can be seen that a square wave having a period of 2 (=T _p /2, doubled in terms of frequency) and a duty ratio of _DV is obtained.

7A and 7B show the duty ratio D _V (from 0 to 1 to 1/1) of the voltage waveform of the output voltage V2 of the representative secondary circuit 20, corresponding to FIGS. 6A and 6B, respectively. 4 increments) and the output voltage V2 of the secondary circuit 20 when controlled by two control signals Sig _S2lead and Sig _S2lag .

FIG _{. 7A shows the case where Sig S2lead is used as the control signal Sig S2 ( when the} time difference of _TD is taken in the leading direction), and _FIG . when the time difference is taken in the delay direction).

7A and 7B, it can be confirmed that the duty ratio _DV of the voltage waveform of the output voltage V2 of the secondary circuit 20 is the same regardless of whether the time difference _TD is advanced or delayed.

In the case of FIG. 7A, the voltage waveform of the output voltage V2 of the secondary side circuit 20 is such that the rise and fall of the output voltage V1 of the primary side circuit 10 are always aligned with the rise of the output voltage V2 of the secondary side circuit 20. The duty ratio _DV is changing. On the other hand, in the case of FIG. 7B, the output voltage of the secondary circuit 20 is such that the rising edge and the falling edge of the output voltage V1 of the primary side circuit 10 always coincide with the falling edge of the output voltage V2 of the secondary side circuit 20. The difference is that the duty ratio _DV of the voltage waveform of the voltage V2 changes.

FIG. 8 shows how the output voltage V2 of the secondary side circuit 20 is output to the load 9 through the smoothing circuit 30 as the output voltage V3 of the smoothing circuit 30. As shown in FIG.

The relationship between the output voltage V2 of the secondary side circuit 20 and its duty ratio _DV , which are input to the smoothing circuit 30, and the output voltage V3 of the smoothing circuit 30 will be described below.

When the smoothing circuit 30 includes the capacitance element 32, the inductor element 31 having an inductance value of _Lout and the capacitance element 32 having a capacitance value of _Cout form a secondary low-pass filter. A load 9 is connected to the output of . The inductance value L _out and capacitance value C _out of the low-pass filter are selected so that the cutoff frequency is lower than the switching frequency (=1/T _p ). The second-order low-pass filter has an attenuation characteristic asymptotically with a slope of -40 dB/dec in the frequency range above the cutoff frequency. Therefore, by setting the switching frequency and its harmonic components as frequency components higher than the cutoff frequency, these can be attenuated.

It is preferable to set the cut-off frequency to a frequency sufficiently lower than the switching frequency, such as a frequency lower than a fraction of the switching frequency, within the range in which the response speed is acceptable.

Even if the smoothing circuit 30 does not include the capacitance element 32, the smoothing circuit 30 operates as a first-order low-pass filter formed by the inductor element 31 and the load 9 if the load 9 is a resistive load. In that case, in the frequency region above the cutoff frequency of the low-pass filter, the attenuation characteristic becomes asymptotic with a slope of -20 dB/dec, and frequency components higher than the cutoff frequency can be attenuated.

Note that the output voltage V3 of the smoothing circuit 30 has a ripple component depending on the frequency characteristic of the attenuation amount of the low-pass filter, and does not become a perfect DC voltage. However, in order to simplify the description, unless otherwise specified, the output voltage V3 and the output current i3 of the smoothing circuit 30 ignore time fluctuations in the order of the cycle _Tp of the reference signal, treated as behaving

In addition, the parasitic components of each element included in the power conversion device 100, such as the winding resistance component of inductor elements and transformers, the capacitance component between terminals, the nonlinearity of the magnetic permeability of the core material, and the dielectric loss of the capacitance element For example, a case is assumed in which the qualitative and quantitative effects are small in grasping the operation of the power conversion device 100 and the device can be approximated as an ideal device. Based on this assumption, an approximation assuming an ideal element is used for the sake of simplicity of explanation. In other words, except for the elements that need to be qualitatively or quantitatively related to the embodiment, the description assumes ideal elements. does not exist or should not exist.

When the output voltage V2 of the secondary side circuit 20 is input to the input of this smoothing circuit 30, the frequency components above the cutoff frequency are removed, and in a steady state, the DC component, which is the average value of V2, is reduced to the smoothing circuit 30. is output as the output voltage V3 of . Since the output voltage V2 of the secondary circuit 20 is a rectangular wave with an amplitude of ±V _DD /2 and a duty ratio of _DV , the voltage V3 _AVE , which is the average value in the steady state, is expressed by the following equation (2 ).

As a specific example, when D _V =1, V3 _AVE =+V _DD /2, when D _V =0.5, V3 _AVE =0, and when D _V =0, V3 _AVE =−V _DD /2.

Then, for example, if the load 9 is a resistive element with a resistance value R, the output current i3 of the smoothing circuit 30, which is the current flowing through the load 9, is V3 _AVE /R.

Note that the inductor element 31 of the smoothing circuit 30 may have a configuration including an inductor element 31a and an inductor element 31b as shown in FIG. 8B, instead of the configuration shown in FIG. If L _out1 + L _out2 = L _out , the configuration of FIG. 8A and the configuration of FIG. can be used.

FIG. 9 shows an example in which an inductive load such as a motor and a charging device for charging a storage battery connected as a DC power supply 8 from the load side are connected to the output side of the smoothing circuit 30 . Here, although not shown in the figure, since the inductive load itself contains an inductor component, in such a case, it is possible to use it as a power converter without including the smoothing circuit 30. .
Alternatively, a commercial AC power supply may be connected to the load 9, the potential of the output voltage V3 may be fed back to the input side of the output target value Vi, and the output target value may be controlled to follow the commercial AC power supply. . In this case, the DC power supply 8 can be charged from the commercial AC power supply via the inductor element 31 of the smoothing circuit 30 .

When a load that consumes power, such as a resistor, is connected as the load 9, the output current i3 flows in the same direction as the output voltage V3 (the direction in which the higher potential flows from V3Hi or V3Lo, or from the power converter 100). look and flow to power).

As the load 9, when an inductive load such as a motor, a charging device for charging the storage battery connected as the DC power supply 8 from the load side, a solar inverter, or the like is connected, the output current i3 is It is also possible to set the operation in the direction in which the output voltage V3 is supplied (the direction in which the higher potential of V3Hi and V3Lo is supplied, and the power conversion device 100 is regenerated).

Consider the case where a constant current source is connected to the output side of the smoothing circuit 30 .

In a steady state with a constant output voltage V3, this current source causes a constant current to flow through _Lout , and since the current is constant, there is no voltage drop across _Lout . Here, the duty ratio of Sig _S2 is represented by _D2 . This current is converted into a rectangular wave current with a duty ratio of _D2 by switch elements S2Hi and S2Lo corresponding to the secondary side of the transformer T of the secondary circuit 20, and is transmitted from the secondary side of the transformer T to the primary side.

The polarity of this current is switched at a duty ratio _D1 by the switch elements S1Hi and S1Lo on the primary side with a time difference _TD .

The current flowing from the constant current source to the DC power _supply 8 is also changed by the same process as the ON/OFF state of the switch element due to the time difference _TD between the primary circuit 10 and the secondary circuit 20 of the voltage VDD of the DC power supply 8. The current has a duty ratio of _DV . The flow of these stories is almost the same as the case of power, so the details are omitted.

The switch element used in the primary side circuit 10 of this configuration, the bidirectional switch element used in the secondary side circuit 20, the transformer T, the inductor element 31 of the smoothing circuit 30, and the like are elements that allow current to flow in both directions. , it can be operated regardless of the direction of the voltage. In this case, the power conversion device 100 outputs the output V3 of the smoothing circuit 30 as an output voltage based on the output target value Vi, and the current flowing through the circuit is regenerated from the output side of the smoothing circuit 30 as an average current, The DC power supply 8 of the primary side circuit 10 is charged.

FIG. 10 shows the relationship between the polarity of the current i2 flowing between the secondary circuit 20 and the smoothing circuit 30 and the direction of the current of the DC power supply 8. As shown in FIG. The direction of the current (discharge or charge) of the DC power supply 8 is hereinafter referred to as "power supply current state", and the power supply current state is expressed as discharging and the power supply current state as charging.

From the relationship between the primary winding and two secondary windings of the transformer T and the bidirectional switch elements S2Hi and S2Lo, the following can be said. That is, when Sig _S2 is + with respect to the polarity of the output current i2 of the secondary circuit 20, the polarity of the output current i1 of the primary circuit 10 is + as shown in FIG. 10(A). When Sig _S2 is negative with respect to the polarity of the output current i2 of the secondary circuit 20, the polarity of the current i1 of the primary circuit 10 is negative, as shown in FIG. 10(B). Specifically, the direction of each current is indicated by an arrow in FIG. 10(A) or FIG. 10(B).

FIG. 11 shows the power supply current state in the on/off state of switch elements S1Hi and S1Lo according to the control signal Sig _S1 of the primary side circuit 10 with respect to the polarity of the current i1 at the output of the primary side circuit 10. It is a thing.

It should be noted that the handling of the current i1=0 may be appropriately included in either the discharging or charging state as a design matter as necessary.

According to the transition of each of the two control signals Sig _S1 and Sig _S2 , there is a transition between the four states listed in FIG. 11, and the manner of transition is as follows.
・Due to the transition of Sig _S1 , there is a transition in the oblique direction in FIG. (While maintaining the polarity of i1, the polarity of the connection between the DC power supply 8 and the primary circuit output is reversed.)
・The transition of Sig _S2 causes a transition in the horizontal direction of FIG. 11 between (open-Hi) and (full-Hi) or between (open-Lo) and (full-Lo). (The state of the switch element of the primary circuit 10 does not change, and the polarity of the current i1 changes.)

For _example , when Sig S1 transitions from the (release -Hi) state (Sig _S1 : +) to - in FIG. and i1 change. That is, in this case, there is a transition from (release-Hi) to (charge-Lo).

Similarly, in the (release-Hi) state (Sig _S1 : +) in FIG. 11, when the polarity of SigS2 is reversed, the current i1 is reversed. That is, there is a transition from (release-Hi) to (charge-Hi).

12A, 12B, 12C, and 12D show current paths related to power supply current states before and after the control signal Sig _S1 of the primary circuit 10 transitions (from + to - or from - to +). , including the period of dead time described in FIG. 3 and its description.

It should be noted that the current indicated by the dotted line in these four figures indicates the current flowing through the diode element D1Hi or D1Lo.

FIG. 12A shows the state of transition from (full-Hi) to (release-Lo) in FIG. This is one form when the power supply current state transitions from charging to discharging due to the transition of Sig _S1 .

Before the transition, the power supply current state is charging, and the control signal Sig _S1 of the primary side circuit 10 is +, that is, the switch element S1Hi is on and S1Lo is off. In this state, the switch element S1Lo is off, and a reverse voltage is applied to the diode element D1Lo. A charge is accumulated with respect to the capacitance. In order to show the locations where charges are accumulated in this way, they are indicated by +Q and -Q in the drawing.

From this state, when the control signal Sig _S1 of the primary side circuit 10 transitions to -, there is a dead time mentioned in FIG. exist.

By turning off both switching elements S1Hi and S1Lo, the path of the current originally flowing through S1Hi is cut off. Due to the induced voltage of the transformer T generated at that time, i1 flows along the current path indicated by the dotted line (charge-Hi').

After the dead time has elapsed, the switch element S1Lo is turned on because the control signal Sig _S1 of the primary side circuit 10 is -. At this time, a recovery loss and accompanying large electromagnetic noise occur in the diode D1Hi due to the abrupt cutoff of the forward current of the diode D1Hi. Lo), the electrostatic energy is consumed by flowing through the closed circuit formed by the switch element S1Lo and the diode element D1Lo. Specifically, the current flowing through the switch element S1Lo and the on-resistance of the switch element S1Lo, the forward voltage of the diode element D1Lo, and the current flowing through them are consumed as heat energy. In addition, the rapidly changing current flows through a closed circuit composed of the ON switch element S1Lo and the diode element D1Lo, thereby generating a magnetic flux in the direction perpendicular to the closed curved surface of the closed circuit. That induces a change in the electric field, ie an electromagnetic wave is generated thereby. That is, it also becomes a source of switching noise in the power converter.

FIG. 12B shows the state of the transition from (full-Lo) to (release-Hi) in FIG.

The on/off state of the switch element and the polarity of the current i1 of the primary circuit 10 in the state before the transition and the state after the transition are opposite to those in FIG. 12A, but the process is similar to that in FIG. 12A. That is, when the switch element S1Hi is turned on, a recovery loss and accompanying large electromagnetic noise occur in the diode D1Lo due to the sudden interruption of the forward current of the diode D1Lo, and the switch element S1Lo and the diode element D1Lo The charge accumulated between the terminals provided in parallel flows through the closed circuit formed by the switch element S1Lo and the diode element D1Lo. As a result, the electrostatic energy is consumed as heat and switching noise.

FIG. 12C shows the transition from (release-Hi) to (charge-Lo) in FIG. This is one form in which the power supply current state transitions from discharging to charging due to the transition of Sig _S1 .

In this case, when the switch element S1Hi is on and the polarity of the current i1 of the primary circuit 10 is +, charge is accumulated in the capacitance between the terminals where the switch element S1Lo and the diode element D1Lo are provided in parallel. state.

In this state, when the control signal Sig _S1 of the primary side circuit 10 transitions from + to -, both switch elements S1Hi and S1Lo are turned off during the dead time period. At this time, due to the induced voltage of the transformer T, V1 changes from _VDD to approximately 0 V (a voltage lower than 0 V by the forward voltage of the diode element D1Lo), and the diode element D1Lo becomes conductive. When the diode element D1Lo becomes conductive, i1 flows along the current path indicated by the dotted line (Charging-Lo'), and the power supply current state changes to charging. Then, when S1Lo is turned on, the state becomes (full-Lo) and the transition is completed.

At this time, unlike the case of FIGS. 12A and 12B, no recovery loss occurs due to the diode D1Hi, no closed circuit occurs due to the switching element S1Lo and the diode element D1Lo, and the electrostatic energy due to the accumulated charge is transferred to the charging current. be part of Therefore, the electrostatic energy due to the accumulated charge is not consumed as heat or switching noise.

FIG. 12D shows the state of the transition from (release-Lo) to (charge-Hi) in FIG. This is one form in which the power supply current state transitions from discharging to charging due to the transition of Sig _S1 , which is the same as in FIG. 12C.

In the state before the transition and the state after the transition, the on/off of the switch elements S1Hi and S1Lo and the polarity of the current i1 of the primary side circuit 10 are opposite to those in FIG. 12C, but the process is the same as in FIG. 12C. That is, the electrostatic energy due to the charge accumulated in the electrostatic capacitance between the terminals of the switching element S1Hi and the diode element D1Hi provided in parallel becomes part of the charging current. In the closed circuit formed by the switch element S1Hi and the diode element D1Hi, no electrostatic energy is consumed and no recovery loss is generated by the diode D1Lo.

12A and 12B, when the control signal Sig _S1 of the primary circuit 10 transitions (from + to - or from - to +), the power supply current state transitions from charging to discharging, and the forward current is The recovery loss due to hard switching of the diode that was flowing and the electrostatic energy stored in the diode that was off are consumed as heat and noise. On the other hand, in FIGS. 12C and 12D, when the control signal Sig _S1 of the primary side circuit 10 transitions, the terminal voltage of the diode through which the forward current is flowing does not change suddenly, and recovery loss does not occur due to soft switching. Also, the power supply current state is transitioning from discharging to charging, and the electrostatic energy due to the charge accumulated between the terminals of the switching element S1Hi and the diode element D1Hi or the switching element S1Lo and the diode element D1Lo is part of the charging current. Become. That is, power loss can be reduced by causing the power supply current state to transition from discharging to charging when the control signal Sig _S1 of the primary side circuit 10 transitions.

(Power supply current state for Sig _S1 , Sig _S2lead , Sig _S2lag )
FIG. 13 shows whether the power supply current state is discharging or charging for Sig _S1 , Sig _S2lead , and Sig _S2lag in the four transition processes shown in FIGS. 12A, 12B, 12C, and 12D. showing. FIG. 13(1) Sig _S1 represents how the control signal Sig _S1 of the primary side circuit 10 alternately outputs + and - at the period _Tp of the reference signal.

Sig _S2lead in FIG. 13(2) is Sig _S1 in the case where the control signal Sig _S2lead , which leads the time _{difference TD} from the cycle Tp of the reference signal, is used as the control signal Sig _S2 for the secondary circuit 20. It shows the state of the output based on .

13(3), the primary side circuit 10 is controlled by Sig _S1 of FIG. 13(1), and the secondary side circuit 20 is controlled by Sig _S2lead of FIG. It shows whether the power supply current state with respect to the polarity of the output current i2 of the secondary circuit 20 when controlled is discharging or charging. 13(3-1) shows the case where the polarity of the output current i2 of the secondary circuit 20 is positive, and FIG. 13(3-2) shows the case where the polarity of the output current i2 of the secondary circuit 20 is negative. be.
In other words, FIGS. 13(2), 13(3), 13(3-1), and 13(3-2) show the period of the control signal Sig _S1 as the control signal Sig _S2 of the secondary circuit 20. It shows the state of the output based on Sig _S1 when using the control signal Sig _S2 lead of the secondary side circuit 20 whose time difference _TD with respect to T _p is leading.

As an example, FIG. 13 (3-1) shows the repetition as in (1) below.
・ Fig. 11 lower left: transition from (release - Lo) to Fig. 11 lower right: transition to (full - Lo) ・ Fig. 11 lower right: transition from (full - Lo) to Fig. 11 upper left: transition to (release - Hi), that is Figure 12B
・ Fig. 11 upper left: transition from (release - Hi) to Fig. 11 upper right: transition to (full - Hi) ・ Fig. 11 upper right: transition from (full - Hi) to Fig. 11 lower left: transition to (release - Lo), that is, Fig. 12A

13(2′), FIG. 13(3′), FIG. 13(3′-1) and _{FIG .} It shows the state of the output based on Sig _S1 when using the control signal Sig _S2lag of the secondary side circuit 20, which is delayed by the time difference _TD from the cycle _Tp .

In these FIGS. 13(3) and 13(3'), three + or - signs are written side by side in parentheses under the notation of "charging" and "discharging". These codes, in order from the left, represent the code of the control signal Sig _S1 for the primary circuit 10 in FIG. 13 represents the sign of signal Sig _S2lead , and the third represents the sign of the polarity of output current i2 of secondary side circuit 20 in FIG. 13(3). Among these three signs, if the number of negative signs (-) is an even number (0 or 2), the power current state is discharging, and if the number is odd (1 or 3), the power current state is charging. Become.

This table shows how the power supply current state transitions according to the control of the switch elements S1Hi, S1Lo, S2Hi and S2Lo and the state of the output current i2 of the secondary circuit 20. FIG. 13(3-2) and 13(3'-1), where the power supply current state changes from discharging to charging before and after the state of Sig _S1 changes. is applicable. These changes have the effect of reducing power loss and reducing switching noise.

FIG. 13(3-2) shows the case where the polarity of i2 is − and Sig _S2lead is used as the control signal Sig _S2 of the secondary circuit 20, and FIG. 13(3′-1) shows the case where the polarity of i2 is + and Sig _S2lag is used as the control signal Sig _S2 of the secondary circuit 20 .

As described above with reference to FIGS. 6, 7A and 7B, the output voltage V3 of the smoothing circuit 30 is the same regardless of whether the time difference _TD advances or lags.

_Then , according to the polarity of the control current _i2 of the secondary circuit 20, as shown in FIG. By doing so, the power supply current state before and after the transition of Sig _S1 can transition from discharging to charging. As a result, the effect of reducing power loss and reducing switching noise can be obtained.

In the above description, both the duty ratio _D1 of Sig _S1 and the duty ratio _D2 of Sig _S2 are assumed to be 0.5, and a brief description has been given so that the essence of the embodiment can be easily understood. However, as explained below, neither D ₁ nor D ₂ need necessarily be 0.5. I will add a brief explanation about it.

FIG. 14 shows an equivalent circuit of the primary side circuit including the exciting inductor on the primary side of the transformer T. As shown in FIG. If a DC voltage is continuously applied to the transformer T, the core will be magnetically saturated and will not function as a transformer.

One potential V1Hi of the output voltage V1 is a square wave that alternately outputs _VDD and 0 by alternately turning on/off the switch elements S1Hi and S1Lo, when the potential of the negative electrode of the DC power supply 8 is set to 0 as a reference potential. It can be expressed as a voltage output. Therefore, it is simply described as an element of the square wave generator in FIG. 14(A).

Here, as shown in FIG. 14B, when the excitation inductor on the primary side of the primary circuit 10 and the transformer T is considered in terms of DC, the excitation inductor is equivalent to a short circuit, and the capacitance element 16a and capacitance element 16b are equivalent to open. In addition, the rectangular wave generator is equivalent to a DC voltage source that outputs a DC component in terms of DC. When the DC voltage (time average of one cycle) of the output voltage V1 of the primary side circuit is represented by <V1>, V1Hi=V1Lo=<V1>. That is, V1Hi and V1Lo, which are the potentials of the two output terminals of the primary side circuit 10, are equal in terms of DC potentials.

Next, consider an AC equivalent circuit as shown in FIG. 14(C). At this time, the DC power supply 8 can be regarded as a short-circuit, and the rectangular wave generator can be regarded as a rectangular wave generator that outputs only the AC component of V1-<V1> after subtracting the DC voltage <V1>. At this time, if the capacitive reactance of the electrostatic capacitive element 16a and the electrostatic capacitive element 16b is a capacitance value that can be regarded as a short circuit at the frequency of the rectangular wave, the potential of the negative electrode of the DC power supply 8 is set to 0, and the AC equivalent circuit is considered, the potential of V1Lo is (almost) zero. Also, since V1Hi is at the same potential as the output of the rectangular wave generator, it is V1-<V1> in terms of alternating current. That is, if the primary side circuit 10 has at least one capacitance element, the output voltage of the primary side circuit 10 is a rectangular wave, considering both the DC equivalent circuit and the AC equivalent circuit. It can be said that only the AC component is applied to the primary side of the transformer T.

That is, since the primary circuit 10 includes the capacitance element, even if the duty ratio _D1 is not 0.5, the rectangular wave voltage output generated by turning on/off the switch elements S1Hi and S1Lo is Only the AC component is applied to the primary side of the transformer T.

From the consideration here, it can be seen that the ratio of the two capacitance values of the capacitance element 16a and the capacitance element 16b of the primary side circuit 10 does not play a role in determining the DC potential in the steady state. Recognize. Therefore, even if the duty ratio _D1 is 0.5, the capacitance values C1Hi and C1Lo of the two capacitance elements 16a and 16b of the primary circuit 10 need not be equal. This can be seen from the discussion here. Furthermore, as can be seen from the equivalent circuits of FIGS. 14B and 14C, the two electrostatic capacitive elements 16a and 16b are equivalent to being arranged in parallel, so either one may be used. I also understand.

If the duty ratios D ₁ and D ₂ are not 0.5, the DC voltage component is still 0, so the absolute values of the upper limit and the lower limit of the amplitude of the output voltage V1 of the primary circuit 10 are not equal, and , the relationship of the output voltage to the time difference changes.

This will be explained with an example.

15A and 15B show a case where D ₁ =D ₂ <0.5 as an example of D ₁ =D ₂ ≠0.5 and the time difference _TD is advanced in FIG. 15A, and the time difference _TD is delayed. is shown in FIG. 15B. 15A and 15B are diagrams corresponding to FIGS. 7A and 7B.

First, V1 will be explained. Since the duty ratio _D1 is not 0.5 and the DC component is 0, the absolute values of the upper limit value (positive side voltage value) and the lower limit value (negative side voltage value) of the voltage V1 are no longer equal.
The DC component is 0, which means that the average of one cycle is 0. Since the duty ratio is D ₁ , the positive voltage interval is D ₁ and the negative voltage interval is (1−D ₁ ). From these facts, the upper limit value (positive voltage value) of the voltage V1 can be expressed as (1−D ₁ )· _VDD , and the lower limit value (negative voltage value) can be expressed as −D1·V _DD . I understand.
In the control circuit, it is necessary to limit the upper limit of the time difference T _D to less than [the lesser of D ₁ ·T _p and (1−D ₁ )·T _p ]. This is because the correspondence between the charging/discharging state of the power supply current and the direction of i2 does not hold. 15A and 15B are examples in which D1<0.5, so it is necessary to limit D1 to less than _D1 ·Tp. Note that even if the upper limit of the time difference _TD is this value, the possible output range for a given _D1 can be covered. Assuming that the upper limit of the output range of the voltage V3 is V _MAX and the lower limit is V _min , they can be expressed as follows.
Upper limit V _MAX : 2·D ₁ ·(1−D ₁ )V _DD
Lower limit V _min : When D ₁ <0.5 −2·D ₁ ² ·V _DD
V _min : When D ₁ >0.5 −2·(1−D ₁ ) ² ·V _DD

16A and 16B show a case where D ₁ <D ₂ as an example of D ₁ ≠D ₂ , and the time difference _TD is advanced in FIG. 16A and the time difference _TD is delayed in FIG. 16B. . These are also diagrams corresponding to FIGS. 7A and 7B.
In FIGS. 16A and 16B as well, the upper limit of voltage V3, which is the average value of voltage V2, is represented by V _MAX and the lower limit by V _min .

When the duty ratios of _D1 and _D2 are different, even if the time difference _TD is the same, the output voltage varies depending on whether it is delayed or advanced. For example, looking at FIG. 16A, which is an example in which the time difference T _D is the leading time difference, as shown in (1) and (2), T _D is from 0 to |D ₂ −D ₁ |·T _p. , voltage V3, which is the average value of voltage V2, remains at the same voltage value _VMAX . By further increasing the time difference _TD , the value of the voltage V3 can be controlled within a range from _Vmin to _VMAX .

On the other hand, looking at FIG. 16B, which is an example in which the time difference _TD is the delay time difference, as shown in (1) to (4), when _TD is between 0 and _D1 · _Tp , , the value of the voltage V3 can be controlled in the range from V _min to V _MAX . Then, as shown by (4) and (5) in FIG. 16B, even if the time difference _TD is further increased, the voltage V3, which is the average value of the voltage V2, remains the same voltage value _VMAX .

Therefore, when switching the delay and advance of the time difference _TD so that the same voltage V3 is obtained, for example, the advance control is performed to change the time difference _TD when D ₁ =D ₂ to | _{D 1} −D ₂ | • Control should be performed in a form in which a correction value for _Tp is added.

Regarding the time difference _TD , although the usable range and design items are added as described above, the correspondence between charge/discharge and occurrence of loss remains the same.

Note that the relationship between Sig _S1 and Sig _S2 in FIG. 13 is determined only by the direction of i2 and the relationship before and after the transition, and there is no element that changes depending on the values of the duty ratios D ₁ and D ₂ . Therefore, the duty ratios D ₁ and D ₂ do not need to be 0.5, nor do they need to be equal.

When the duty ratios D ₁ and D ₂ are not 0.5, or when these two duty ratios are unequal, as an example, the control circuit 40 can be configured as follows. That is, at the rising edge of the reference signal, a triangular wave (triangular wave 1) similar to the conventional one is prepared which has an upper limit of its amplitude. In addition, another triangular wave (triangular wave 2) having a lower amplitude limit at the falling edge of the reference signal is also prepared. If the duty ratio _D0 is 0.5, the above two triangular waves overlap and only one is required. 1 and 5 exemplify a case in which only one such triangular wave is required.

(Conventional power converter)
A conventional power converter does not have a configuration in which the secondary side control signal is switched through a determination circuit according to the polarity of i2.

Therefore, in a conventional power conversion device, for example, with respect to the load of the resistive element, at least either positive output voltage or negative output voltage, power loss and switching noise are relatively larger than the other. It may work.
(Summary of effects)
As described above, in the power conversion device 100, by switching the secondary side control signal according to the polarity of i2 through the determination circuit 43, effects such as reduction of power loss and reduction of switching noise can be obtained.

This can be said to indicate the superiority of the present embodiment over the conventional power converter.

[Second Embodiment] (Modified Example of Control Circuit)
The second embodiment is another example of the control circuit 40. FIG.

17A, 17B, 17C, and 18 show control circuits according to the second embodiment.

17A, 17B, and 17C show modifications of the control circuit 40, and FIG. 18 shows the difference between the secondary side circuit 20 and the smoothing circuit 30 when the input Sign of the determination circuit 43 is positive and negative, respectively. shows the relationship between the polarity of the current i2 flowing through and the power supply current state.

17A, 17B, 17C, and 18, only different parts from the first embodiment will be explained, and explanations of parts common to the first embodiment will be omitted.

FIG. 17A shows a control circuit 40a as an example of modification of the control circuit 40. FIG.

Depending on whether the input signal Sign of the determination circuit 43 is positive or negative, the switching means changes the logic signal of the control signal Sig _S1 of the primary side circuit 10 as it is if it is positive, or to a signal obtained by inverting the logic of the reference signal by the inverting circuit INV2 if it is negative. SW2 is used to configure the primary side control circuit 41a. Further, the secondary side circuit control signal Sig _S2 is generated when the input signal Sign of the determination circuit 43 is positive by configuring the lead/lag signal generation circuit 103 and the switching means SW3 as in the secondary side control circuit 42a of FIG. A signal delayed by a time difference _TD with respect to the reference signal is selected, and if the input signal Sign of the determination circuit 43 is negative, a signal obtained by inverting the logic of the signal advanced by a time difference _TD with respect to the reference signal is selected. .

FIG. 18 shows the relationship between the polarity of i2 and the power supply current state when the input signal Sign to the determination circuit 43 is positive or negative. When the polarity of i2 is positive, Sign is selected to be positive, and when the polarity of i2 is negative, it is selected to be negative. It can be seen that there is a transition from discharging to charging instead of ). By selecting in this way, the effect of reducing power loss and reducing switching noise can be obtained.

In FIG. 17A, the example in which the output of the comparator COMP2 is inverted in the lead/lag signal generation circuit 103 and the switching means SW3 is given. However, as Sig _S2 , when the input signal Sign of the determination circuit 43 is positive, it suffices to obtain a delayed signal corresponding to the time difference _TD with respect to _the reference signal. Any configuration may be used as long as a signal that inverts the logic with respect to is obtained. For example, in the configuration in which the output of either the lead signal generation circuit 101 or the lag signal generation circuit 102 in the control circuit 40 is selected, the lead/lag signal generation circuit 104 can be used by minor design changes such as logic inversion. It is also possible to change the configuration to the one shown in FIG. 17B (control circuit 40b in FIG. 17B).

Further, after preparing the lead signal generation circuit 101 and the delay signal generation circuit 102 in the control circuit 40, the output of the delay signal generation circuit 102 may be inverted (the delay signal generation circuit 102a) (the control shown in FIG. 17C). Secondary side control circuit 42c) of circuit 40c. Examples of the first embodiment also include configurations such as these. 17B and the delay signal generation circuit 102a in FIG. 17C, an output signal in the form of a bar above X is used. It represents that the logic of the output of signal generation circuit 103 and the output of delay signal generation circuit 102 in FIG. 1 are inverted.

It is also possible to implement functions equivalent to at least part of the functions of the control circuit 40a, the control circuit 40b, and the control circuit 40c using a programmable device or processor such as a PLD, FPGA, or CPU. ) may be included in the components.

As in the modification described in the second embodiment, the determination result Sel of the determination circuit 43 is obtained not only for the control signal Sig _S2 for the secondary circuit 20 but also for the control signal Sig _S1 for the primary circuit 10 . It is also effective to reduce power loss and switching noise by applying switching by .

[Third Embodiment] (Modification of Determination Circuit)
The third embodiment is a modification of the configuration of the comparator COMP3 in the determination circuit 43. FIG.

FIGS. 19 to 22 show examples of the determination circuit 43a and the determination circuit 43b according to the third embodiment, and states of related signals.

FIG. 19 shows a determination circuit 43a as a modified example of the determination circuit 43. As shown in FIG. FIG. 20 shows voltages and currents at various parts centering on the smoothing circuit 30, and _FIG . 20 shows an example of waveforms of the output voltage V2 of the circuit 20 and the output current i2 of the secondary circuit 20. FIG. FIG. 22 shows an example in which a means for holding the output of comparator COMP3 is added.

Based on FIGS. 19 to 22, only parts different from other embodiments will be described, and descriptions of parts common to other embodiments will be omitted.

FIG. 19 shows a determination circuit 43a as a modified example of the determination circuit 43 according to the third embodiment. The determination circuit 43a is an example of a configuration of a comparator with hysteresis by positively feeding back the output of the comparator COMP3. If the ripple current crosses over the threshold value of the comparator COMP3, a state occurs in which the determination signal is switched at intervals shorter than the cycle of the reference signal. The output of the comparator COMP3 is divided by two resistance elements having resistance values R3 and R4, respectively, and positive feedback is applied to the input side of the comparator COMP3 to apply hysteresis to the input signal Sign. Since this figure is an example of replacing the portion of the comparator COMP3 in the determination circuit 43 of FIG. 1, a logic inverting circuit INV2 is provided in the output portion. The output of INV2 may be used as the determination result Sel. This can have the effect of reducing the influence of noise in the signal Sign from the current sensor CT on the result of signal comparison. A comparator with hysteresis is also commercially available as a component, and such a comparator may be used as it is as the comparator COMP3.

Here, an example is given in which the hysteresis width is determined by design. In order to obtain a suitable width of hysteresis provided by adding an external circuit to the comparator COMP3, the time dependence of the current i2 flowing through the inductor element 31 of the smoothing circuit 30 will be examined below. The direction of current flow from the secondary circuit 20 side to the load 9 side is defined as the positive direction of the current.

FIG. 20 shows voltages and currents at various parts, centering on the smoothing circuit 30. FIG.

The voltage _VL applied across the inductor element 31 can be represented by V2-V3 _AVE with one end of the inductor element 31 on the load 9 side as a reference.

The current i2 flowing through the inductor element 31 is proportional to the integral amount of the current i3 (=output current of the smoothing circuit 30) flowing through the load 9 and the voltage V2-V3 _AVE applied across the inductor element 31 in the steady state. It is the sum of the amount of fluctuation that When the current flowing through the inductor element 31 is in a steady state, and the duty ratio of the output waveform of the output voltage V2 of the secondary circuit 20 _{is DV} , the amplitude of this variation (the width between the upper limit and the lower limit) i _rpl is given below. can be expressed in the form of

ｉ３_Ｍは、Ｒの抵抗負荷に対する最大電流を表わす。また、ｉ_{Ｍ＿ｒｐｌ}は、式（３）で、Ｄ_Ｖ＝１／２としたときのリップル電流の振幅ｉ_ｒｐｌの値であり、最大のリップル電流の振幅を表わす。 FIG. 21 shows the output voltage V2 of the secondary circuit 20 and the inductor when the duty ratio _DV of the output voltage waveform of the secondary circuit 20 ranges from 0 to 1 at 1/4 intervals when the load 9 is a resistive load. The state of the current i2 flowing through the element 31 (the output current of the secondary circuit 20) is shown.
However, in FIG. 21, the following are used as constants.

i3 _M represents the maximum current for a resistive load of R. Also, i _{M_rpl} is the value of the ripple current amplitude i _rpl when D _V =1/2 in Equation (3), and represents the maximum ripple current amplitude.

The current i2 flowing through the inductor element 31 in the steady state has the average value (DC component) of the output current i3 of the smoothing circuit 30, and the ripple amplitude is D _V =1/2, which is the maximum amplitude ripple.

In order to suppress fluctuations in the determination result within a short time of T _p /2 or less due to this ripple, the hysteresis current width i _hys of COMP3 should be made larger than the maximum ripple value i _{M_rpl} . The hysteresis current width i _hys can be determined by selecting the resistance values R3 and R4 of the two resistors based on the gain (current-voltage conversion ratio) of the current sensor CT and the binary output voltage level of the comparator COMP3. . Details of specific constant design are omitted.

The maximum value _{iM_rpl} of this ripple is an approximation when all signals with frequencies equal to or higher than the cutoff frequency of the low-pass filter of the smoothing circuit 30 are removed. Based on this approximation, the ripple current amplitude is determined by the period T _p of the reference signal, the voltage V _DD of the DC power supply 8, and the inductance value Lout of the inductor element 31 of the smoothing circuit 30, and does not depend on the load. The property that the ripple current does not depend on the load 9 is an advantageous feature in designing the hysteresis width of the comparator.

In implementation, the hysteresis width may be adjusted by including the current value _{i_hys} , which is determined regardless of the load 9, in the adjustment range. As an example, if one of R3 and R4 in FIG. 19 is a series connection of a fixed resistor and a variable resistor, the hysteresis width can be adjusted. Then, in a state where the actual device is operated, the duty ratio _DV of the output voltage waveform of the secondary side circuit 20 is 0.5, and the output voltage V3 of the smoothing circuit 30 is 0, and the determination result Sel becomes i2. The hysteresis width is adjusted according to the included ripple current. With such adjustment, it is possible to operate the determination circuit 43a without inversion of the determination result Sel due to the ripple current in the actual device.

FIG. 22 shows an example in which a holding circuit, which is means for holding the output of the comparator COMP3, is added to the determination circuit 43b according to the third embodiment.

A signal Sign from the current sensor CT is connected to the input of the comparator COMP3. there is With such a configuration, the output of the comparator COMP3 is held until the next rising signal by the trigger signal CL, which is the output of the comparator COMP4, is input to the trigger input of the D-FF.

A trigger signal CL input to the trigger input of the D-FF is generated by a triangular wave and a comparator COMP4 in this configuration example. The output of the D-FF is the output Sel of the decision circuit 43b.

Since the upper and lower limits of the amplitude of the triangular wave correspond to the rising and falling portions of the reference signal, the trigger signal CL has rising and falling edges between the rising and falling edges of the reference signal. Since the ON/OFF transition of the switch element of the primary circuit 10 occurs at the rise and fall of the reference signal, the rise and fall of the trigger signal CL correspond to the ON/OFF transition of the switch element of the primary circuit 10. Between transitions.

The comparison potential of the comparator COMP4 does not need to be a common potential, and may be arbitrarily determined to be a potential between the upper limit and the lower limit of the triangular wave voltages to be compared.

According to the third embodiment, it is possible to avoid the influence of the ripple component of the output current i2 of the secondary circuit 20 on the determination result Sel generated based on the comparator COMP3.

As in this modification, providing hysteresis for the comparator COMP3 of the decision circuit 43 and providing the decision result with the trigger signal CL synchronized with the reference signal are intended to achieve stable operation under the ripple current. is effective.

[Fourth Embodiment] (Modified Example of Output Current Detection)
4th Embodiment is a modification of the power converter device regarding the current sensor CT.

FIGS. 23 to 29 show configuration examples of power converters 200, 210, 220, 230 and 240, and power converters 230 and 240 according to the fourth embodiment.

FIG. 23 shows an example of a configuration in which a low-pass filter LPF is arranged between the current sensor CT and the determination circuit 43. As shown in FIG. 24 shows an example in which the current sensor CT is configured to detect the output current i3 of the smoothing circuit 30, and FIG. 25 shows an example in which the output of the load 9 is configured with a voltage sensor VT. FIG. 26 shows an example in which the current sensor CT is arranged at the output terminal of the DC power supply 8, and FIG. ing. FIG. 28 shows a current sensor CT, which is configured between the primary side circuit 10 and the secondary side circuit 20. As shown in FIG. FIG. 29 shows the current sensor CT, and shows a configuration example of the determination circuit 43e in an example configured between the primary side circuit 10 and the secondary side circuit 20. As shown in FIG.

Based on FIGS. 23 to 29, only parts different from other embodiments will be described, and descriptions of parts common to other embodiments will be omitted.

FIG. 23 shows an example of a power conversion device 200 in which a low-pass filter LPF is arranged between the current sensor CT and the determination circuit 43. As shown in FIG.

Since the period of the ripple component is T _p /2, the cutoff frequency of the low-pass filter LPF is sufficiently lower than the frequency of the ripple component, for example, a fraction of the frequency. This makes it possible to attenuate the ripple component in the steady state to a desired amount or less. In addition, an effect of attenuating external noise other than the ripple component can be obtained.

The low-pass filter LPF may be, for example, a first-order low-pass filter composed of a resistive element and a capacitance element, an inductor element and a resistive element, or a high-order low-pass filter. Furthermore, the filter does not need to be composed only of passive elements, and may be an active filter. It is also possible to A/D convert the detection signal Sign of the current sensor CT and use a digital filter. Of course, after passing through a low-pass filter as an analog circuit, A/D conversion may be performed and a digital filter may be used. Especially in the case of a digital filter, if the integration interval in the moving average filter is set to T _p /2 or a positive integral multiple thereof, the sensitivity to the frequency component of the ripple component becomes 0, which is suitable for the purpose of removing the ripple component. configuration. Regarding filters, both analog filters and digital filters have a method of designing a filter for obtaining a desired cutoff frequency according to the attenuation amount, response speed, etc., and the details thereof are omitted.

FIG. 24 shows an example of a power conversion device 210 configured to detect the output current i3 of the smoothing circuit 30 instead of the output current i2 of the secondary circuit 20 with the current sensor CT.

In this example, the current sensor CT can detect a signal equivalent to when the smoothing circuit 30 is used as the low-pass filter LPF in FIG. This can be considered as follows. That is, for example, when considering a time longer than several times the period _Tp of the reference signal, the average of the ripple component is 0, so the average of the output current i2 of the secondary side circuit 20 is the output current i3 of the smoothing circuit 30. . Therefore, if the output current i3 of the smoothing circuit 30 is used as a detection signal by the current sensor CT, the output current i2 of the secondary side circuit 20 can be obtained without separately providing the low-pass filter LPF as described in FIG. Averaged signal detection is possible.

Note that the output voltage V3 of the smoothing circuit 30 is a voltage obtained by smoothing the square wave of the output voltage waveform of the secondary side circuit 20 with a duty ratio _DV by the smoothing circuit 30. not. Therefore, a holding circuit using a low-pass filter, COMP3 hysteresis, and D-FF may be used in combination with the signal obtained by detecting the output current i3 of the smoothing circuit 30 by the current sensor CT.

FIG. 25 shows an example of a power conversion device 220 in which a voltage sensor VT is configured at the output of the load 9. As shown in FIG.

If the load 9 is a resistance load such as a resistance element, the output current i3 of the smoothing circuit 30 is proportional to the output voltage V3 of the smoothing circuit 30. Therefore, even if the output voltage V3 of the smoothing circuit 30 is detected instead of the output current i3 of the smoothing circuit 30, an effect equivalent to that of detecting the output current i3 of the smoothing circuit 30 can be obtained.

Note that the output voltage V3 of the smoothing circuit 30 in the steady state is determined based on the setting of the output target value Vi. Therefore, if the load 9 is a resistive load, instead of obtaining the determination result Sel based on the detection signals of the current sensor CT and voltage sensor VT, the determination result Sel may be generated based on the positive/negative of the output target value Vi. good.

Further, a manual changeover switch may be provided as means for generating a further simplified determination result Sel. For example, in the case of a resistive load with a fixed output target value Vi, if it is possible to manually set whether to lead or lag, it can be used sufficiently as a means to achieve the purpose of obtaining operation with little loss.

FIG. 26 shows a configuration example in which the discharge or charge state of the DC power source 8 is directly detected by configuring the current sensor CT to detect the output current i0 of the DC power source 8. FIG.

FIG. 27A shows that the output state of JK-FF (JK-flip-flop) does not change if the output of the comparator COMP3 is + with respect to the falling transition of the control signal Sig _S1 of the primary side circuit 10. First, it is an example of a circuit in which the output state of JK-FF is inverted if the output of comparator COMP3 is -.

+ and - of the output of the comparator COMP3 indicate that the output current i0 of the DC power supply 8 in FIG. ing.

Immediately before the fall of the control signal Sig _S1 of the primary circuit 10, when the output of the comparator COMP3 is +, the DC power supply 8 is in a discharging state, and when the output of the comparator COMP3 is -, the DC power supply 8 is the state of charge.

When the DC power supply 8 is in a charging state when the control signal Sig _S1 falls, that is, when the output of the comparator COMP3 is -, the output Sel of the determination circuit is inverted. With this configuration, the power supply current state of the DC power supply 8 can transition from discharging to charging before and after the transition of the control signal Sig _S1 .

FIG. 27B shows that when the output of the comparator COMP3 is + with respect to the rising transition of the control signal Sig _S1 of the primary side circuit 10, the output state of JK-FF does not change, and the output of the comparator COMP3 is This is an example of a determination circuit in which the output state of JK-FF is inverted if +.

FIG. 27B shows an example in which the determination is made at the rising transition of the control signal Sig _S1 of the primary side circuit 10 in contrast to the example of FIG. 27A.

FIG. 27(C) shows a circuit configured so that the DC power supply 8 changes from discharging to charging at both rising and falling transitions of the control signal Sig _S1 of the primary side circuit 10 .

The two inputs of the exclusive OR Ex-OR1 are the control signal Sig _S1 of the primary side circuit 10 and the signal passed through the time constant circuit of the resistance element R _JK and the capacitance element C _JK . Input two signals. As a result, the output of the exclusive OR Ex-OR1 becomes a logic signal that rises at both the rising and falling edges of the control signal Sig _S1 of the primary side circuit .

The time constant of the resistance element R _JK and the capacitance element C _JK may be configured to be, for example, a fraction of the period of the control signal Sig _S1 of the primary circuit 10 or less.

With such a configuration, the determination circuit shown as an example in _FIG . A determination can be made as to whether or not

As shown in these examples, according to the polarity of the output current i0 of the DC power supply 8, when the control signal Sig _S1 of the primary circuit 10 transitions, the polarity of the output current i0 of the DC power supply 8 is determined to be -. By inverting the output Sel of the circuit, the DC power supply 8 can be controlled from discharging to charging.

FIG. 28 shows a current sensor CT configured to detect the output current i1 of the primary circuit 10. FIG.

FIG. 29A shows that when the output of the comparator COMP3 is + with respect to the falling transition of the control signal Sig _S1 of the primary side circuit 10, the output state of JK-FF does not change and the output of the comparator COMP3 does not change. This is an example of a circuit in which the output state of JK-FF is inverted if is -.

The + and - outputs of the comparator COMP3 indicate that the output current i1 of the primary side circuit 10 in FIG. represents Immediately before the fall of the control signal Sig _S1 of the primary circuit 10, that is, when the control signal Sig _S1 is +, the positive electrode side of the DC power supply 8 is switched to the primary side by turning on the switch element S1Hi of the primary circuit 10. This is the state where the potential of the output of the circuit 10 is connected to the side indicated by V1Hi.

Immediately before the fall of the control signal Sig _S1 of the primary circuit 10, when the output of the comparator COMP3 is +, the DC power supply 8 is in a discharging state, and when the output of the comparator COMP3 is -, the DC power supply 8 is the state of charge.

When the DC power supply 8 is in a charging state when the control signal Sig _S1 falls, that is, when the output of the comparator COMP3 is -, the determination circuit is configured to invert the output Sel of the determination circuit. By doing so, it is possible to change the power supply current state of the DC power supply 8 from discharging to charging before and after the transition of the control signal Sig _S1 .

FIG. 29B shows that when the output of the comparator COMP3 is negative with respect to the rising transition of the control signal Sig _S1 of the primary side circuit 10, the output state of JK-FF does not change, and the output of the comparator COMP3 is This is an example of a determination circuit in which the output state of JK-FF is inverted if +. Immediately before the rise of the control signal Sig _S1 of the primary circuit 10, that is, when the control signal Sig _S1 is -, the negative electrode side of the DC power supply 8 turns on the switch element S1Lo of the primary circuit 10 to turn on the primary circuit. 10 is connected to the side indicated by V1Hi. Immediately before the rise of the control signal Sig _S1 of the primary circuit 10, when the output of the comparator COMP3 is -, the DC power supply 8 is in a discharging state, and when the output of the comparator COMP3 is +, the DC power supply 8 is It is in charging state.

The determination circuit is configured to invert the output Sel of the determination circuit when the DC power supply 8 is in a charging state when the control signal Sig _S1 rises, that is, when the output of the comparator COMP3 is +. By doing so, it is possible to change the power supply current state of the DC power supply 8 from discharging to charging before and after the transition of the control signal Sig _S1 .

FIG. 29(C) is a circuit configured so that the DC power supply 8 changes from discharging to charging at both rising and falling transitions of the control signal Sig _S1 of the primary side circuit 10 .

Whether or not to invert the output Sel of the determination circuit can be determined at each transition timing of the rising edge and the falling edge.

27A, 27B, 29C and 29A, 29B, and 29C, the signal from the current sensor CT is passed through the low-pass filter LPF to the comparator COMP3. It may be the + side input signal Sign. A hysteresis comparator may be used as the comparator COMP3 in these determination circuits.

[Fifth Embodiment] (Modification of Primary Side Circuit)
5th Embodiment is an example regarding the structure of the various primary side circuits 10 of the power converter device 100. FIG.

In either configuration, a rectangular wave-shaped output voltage is obtained as the output voltage V1 of the primary side circuit 10 .

30, 31A and 31B show a power converter according to the fifth embodiment.

30A, 30B, 30C, and 30D show an example of a half-bridge type embodiment as the configuration of the primary side circuit 10. FIG. 31A and 31B show an example of a full-bridge type embodiment as the configuration of the primary side circuit 10. FIG.

30, 31A, and 31B, only different parts from other embodiments will be explained, and explanations of parts common to other embodiments will be omitted.

The primary circuit 10a in FIG. 30A and the primary circuit 10b in FIG. 30B are arranged in the primary circuit 10 in the first embodiment shown in FIG. This is a modified example in which only one of the electrostatic capacitive elements is used.

As described in the description of the first embodiment, the capacitance value of the capacitance element can be C1Hi.noteq.C1Lo. FIGS. 30A and 30B can also be considered as examples in which one of the two capacitance elements has a capacitance value of zero.

A primary circuit 10c of FIG. 30C and a primary circuit 10d of FIG. As shown in FIGS. 30(C) and 30(D), DC power supply 8' includes DC power supply 1 and DC power supply 2 connected in series. Let V _DD1 be the voltage value across the DC power supply 1 and V _DD2 be the voltage value across the DC power supply 2 . At this time, when only the switch element S1Hi is on, the potential of the negative electrode side of the DC power supply 1 (=the potential of the positive electrode side of the DC power supply 2), that is, V1Lo is set to 0, and the potential of the output voltage V1Hi becomes _VDD1 . Also, when only the switch element S1Lo is on, the potential of the output voltage V1Hi becomes -V _DD2 . Also in these forms, a square wave is output as V1. However, in the form of FIG. 30(D), since the voltage is output without passing through the capacitance element, the DC voltage is continuously applied to the primary side of the transformer T connected to the subsequent stage of this output. It is necessary to control the switch element of the primary side circuit 10d so that the voltage is not applied.

Specifically, regarding the voltages V _DD1 and V _DD2 of the two DC power sources 1 and 2 constituting the DC power source 8′ and the duty ratio D ₁ of the control signal of the primary side circuit 10d, the following equation is established. must be controlled to
D ₁ : 1-D ₁ = V _DD2 : V _DD1
... (Formula 6)

The process of accumulating and discharging electric charges when the switching element of the primary circuit 10d is turned on/off is the same as the example in the first embodiment, and thus the details thereof will be omitted.

A primary side circuit 10e in FIG. 31A and a primary side circuit 10f in FIG. 31B show an example of a full-bridge type embodiment as the configuration of the primary side circuit.

In contrast to the half-bridge type including two electrostatic capacitive elements 16a and 16b, the full-bridge type consists of switch elements (S1'Hi, S1'Lo) and diode elements ( D1'Hi, D1'Lo) are arranged in parallel. It should be noted that the capacitance element having the capacitance value C1M provided in FIG. Alternatively, two may be provided at both ends of the primary side output. The on/off of the switch element S1'Lo is controlled according to the on/off of the switch element S1Hi, and the on/off of the switch element S1'Hi is controlled according to the on/off of the switch element S1Lo. . The same applies to dead time. In FIG. 31A, the output of the primary side driver Drv _S1 surrounded by a square with a dotted line represents a set of control signals in which the ON/OFF states of the switch elements are the same.

The on/off control of the switch elements S1Hi and S1Lo with respect to the control signal Sig _S1 of the primary side circuits 10e and 10f is the same as in the other embodiments, and thus description thereof is omitted.

The full-bridge type has a configuration in which both poles of the DC power supply 8 become the output of the primary side circuit 10e in either positive or negative direction via switching elements that are turned on. Therefore, in the example of the fifth embodiment, V1, which is the potential of V1Hi based on V1Lo, is twice the output voltage of the half-bridge type.

31B, as in the case of FIG. 30(D), since the voltage is output without passing through the capacitance element, the primary side of the transformer T connected to the subsequent stage of this output It is necessary to control the switch element of the primary side circuit 10f so that the DC voltage component is not continuously applied.

Specifically, it is necessary to set the duty ratio _D1 of the control signal for the primary circuit 10f to 0.5.

In this way, there is a configuration example other than the configuration of the first embodiment as a configuration example in which a rectangular wave-shaped output voltage can be obtained as the output voltage V1 of the primary side circuit 10 . However, it is not limited to these.

[Sixth Embodiment] (Specific example of bidirectional switch element)
The sixth embodiment is an example relating to the configuration of the switch element of the power conversion device 100. FIG.

Based on FIG. 32, only parts different from other embodiments will be explained, and explanations of parts common to other embodiments will be omitted.

FIG. 32 shows a specific configuration example of a bidirectional switch element.

A bidirectional switch element needs to be able to pass current in both directions when it is on, and to block current in either direction when it is off.

The switch elements S1Hi and S1Lo used in the example of the configuration of the primary side circuit 10 must be able to flow in the desired direction of current when turned on. It is necessary to be able to block forward current, but it is not necessary to be able to block reverse current.

In parallel with the switch elements S1Hi and S1Lo, diode elements D1Hi and D1Lo are attached to the outside in the direction opposite to the forward current, or equivalently, an element in which such a diode element is inherent is used as the switch element. It is used as an element in which diode elements are provided in parallel. A typical example of such an element is a MOS-FET.

FIGS. 32A to 32E exemplify MOS-FETs as the switching elements inherent in the bidirectional switching elements, but J-FETs, bipolar transistors, IGBTs, SiC devices, GaN transistors, etc., can also be used as MOS transistors. - switch elements other than FETs can be used; Also, P-channel elements and PNP elements can be used instead of N-channel elements and NPN elements, or they can be used together. Furthermore, when a diode element is provided in parallel with the switch element, a built-in diode (including a parasitic diode) of the switch element can be used instead, thereby reducing mounting area and cost.

FIGS. 32(A) and 32(B) show an example of a bidirectional switch element configured by connecting two switch elements S5 and S5' in series in opposite directions. The switch elements S5 and S5' may include a parasitic diode or a built-in diode so that a reverse current can flow when a reverse voltage is applied. Further, if necessary, diode elements D5 and D5' marked with [ ] in FIGS. 32A and 32B may be separately connected in parallel with switch elements S5 and S5'.

As an example, in FIG. 32A, consider a case where the switch element S5 on the left side is turned on and current flows from left to right. A so-called forward current flows from the drain to the source in the switch element S5 on the left side. Even if the right switch element S5' remains off, the current flows from left to right via the diode element built in the right switch element S5' or via the diode element D5' provided in parallel. flow. However, if the switch element S5' on the right side is also turned on, the current also flows from the left to the right in the switch element S5' on the right side, so that the switching loss can be reduced.

In this case, a so-called reverse current flows from the source to the drain in the right switch element S5'. The electrical characteristics of the switch element S5' are often not specified for the reverse current. However, it is known that the illustrated MOS-FET and J-FET are capable of bidirectional current flow when turned on. It is also known that bipolar transistors, IGBTs, and the like generally have a low reverse withstand voltage, but are capable of bidirectional current flow when turned on. In this embodiment, if necessary, diodes D5 and D5' are connected in parallel to the switch elements S5 and S5' so that the reverse voltage applied to the switch elements S5 and S5' can be controlled by the diode elements D5 and S5'. It can be suppressed to the forward voltage of D5'. Therefore, the problem of damage to the switch elements due to the low reverse withstand voltage is solved by connecting diode elements having a forward voltage lower than the reverse withstand voltage of the switch elements S5 and S5' in parallel with the switch elements S5 and S5'. It is possible to avoid this by making a separate connection.

When both switch elements S5 and S5' are turned off in FIGS. 32A and 32B, no current flows through diode elements D5 and D5' provided in parallel with switch elements S5 and S5'. Therefore, the current on both the left and right sides of the figure can be turned off.

FIGS. 32(C) and 32(D) show modifications of the bidirectional switch element, respectively, in which the connection order of the switch elements S5 and S5' and the diode elements D5 and D5' are reversed. .

It is also possible to change the connection order of only one of the sets of the switch elements S5 and S5' and the diode elements S5 and S5' connected in series. That is, it is also possible to connect the upper group in FIG. 32(C) and the lower group in FIG. 32(D) in parallel. A parallel connection is also possible.

FIG. 32(E) is a further configuration example of the bidirectional switch element. There is an advantage that only one switch element S5 is required, and current can flow in both directions simply by turning on this switch element S5.

FIG. 32(F) shows a modification possible in the case of switch elements S6 and S6' having sufficient reverse withstand voltage. For example, there is an IGBT (reverse blocking IGBT) having a reverse withstand voltage, which can be used in the configuration shown in FIG. 32(F).

FIG. 32(G) is a configuration example using the bidirectional switch element S7. In FIG. 32(G), a symbol such as a bidirectional IGBT is used, but a bidirectional switch element configured by a combination of bipolar transistors or the like may also be used.

Note that the bidirectional switch element is not limited to the above examples, and any element or configuration may be used as long as it can turn on/off a bidirectional current, and is limited to the examples described above. isn't it.

In this manner, a bidirectional switch element can be configured using various switch elements that are given as specific examples of the switch element, and effects such as reduction of power loss can be obtained.

[Seventh embodiment] (Modification of secondary circuit)
The seventh embodiment is a modification of the configuration of the secondary side circuit of the power converter.

Figures 33, 34A, 34B, 34C, 34D, and 35 illustrate example embodiments of secondary side circuitry.

FIG. 33 shows an example of an embodiment of a secondary side circuit composed of bidirectional switch elements. Figures 34A, 34B, 34C, and 34D show examples of embodiments of secondary side circuits that are composed of switch elements. FIG. 35 shows an embodiment of a secondary circuit in which the winding w2 on the secondary side of the transformer T is composed of a single turn.

33, 34A, 34B, 34C, 34D, and 35, only parts different from other embodiments will be described, and descriptions of parts common to other embodiments will be omitted. .

FIG. 33 shows an example of an embodiment of a secondary side circuit composed of bidirectional switch elements. In addition to the arrangements shown in FIGS. 1 and 4, secondary circuit 20a in FIG. 33A and secondary circuit 20b in FIG. , and the arrangement shown by the secondary side circuit 20c in FIG.

Figures 34A, 34B, 34C, and 34D show examples of embodiments of secondary side circuits that are composed of switch elements. As illustrated in the sixth embodiment, the bidirectional switch element must be a combination of two switch elements or a combination of diode elements. In some cases, there are cost advantages to using switch elements. 34A to 34D show an example of a configuration of a secondary side circuit using a switch element as another embodiment including such a case. 34A, the secondary circuit 20e in FIG. 34B, the secondary circuit 20f in FIG. 34C, and the secondary circuit 20g in FIG. By connecting the switch elements to both poles of the same winding on the secondary side of the transformer T in the form of a shape, an example of a configuration in which the two together have the same function as a bidirectional switch element is shown.

A secondary circuit 20h in FIG. 35 is an example in which the winding w2 on the secondary side of the transformer T is composed of a single winding. The polarity of the output voltage V2 of the secondary side circuit with respect to the output voltage V1 of the primary side circuit 10 is switched as it is or in the opposite direction by turning on/off the bidirectional switch element of the secondary side circuit 20h. be able to.

In this way, as examples of the configuration of the secondary circuit, it is possible to change the arrangement of the switch elements, to use not only bidirectional switch elements but also switch elements. It is also possible to configure the winding on the side with a single turn.

[Eighth embodiment] (Example of obtaining an insulated DC voltage source)
The eighth embodiment is a modified example of obtaining an insulated DC voltage source by the transformer T of the power converter, and FIG. 36 shows an example of an embodiment in which an insulated DC voltage source is added. .

Based on FIG. 36, only parts different from other embodiments will be described, and descriptions of parts common to other embodiments will be omitted.

The secondary side circuit 20m of FIG. 36 has additional windings w23-1 to w23-n (n: a positive integer) as tertiary windings to the transformer T, and rectifier circuits 24-1 to 24- A configuration example is shown in which DC voltage sources 25-1 to 25-n, each of which is insulated, are obtained by providing .n. A smoothing circuit, a voltage stabilizing function, etc. can be added to the rectifying circuits 24-1 to 24-n.

The insulated DC voltage sources 25-1 to 25-n obtained in this way are, for example, power sources for driving circuits of switch element drivers, current sensors CT and voltage sensors VT of a type requiring power sources for driving. It can be widely used for a plurality of circuits having different operational reference potentials, such as a power source for a sensor when using . After the control circuit 40 is driven by an external DC voltage source for starting the circuit, the DC voltage source for driving the control circuit 40 is obtained from any one of the DC voltage sources 25-1 to 25-n. It is also possible to configure to switch to . With such a configuration, the power supply for driving the control circuit 40 and the switch element drivers of the primary side circuit 10 can be obtained from any one of the DC voltage sources 25-1 to 25-n.

Needless to say, the DC voltage sources 25-1 to 25-n can be used without being insulated for driving circuits having the same reference potential.

[Other Embodiments] (Feedback)
In the above-described embodiment, the output voltage V3 of the smoothing circuit 30 is monitored and fed back to the control circuit 40 via the insulated transmission section, so that the actual output voltage V3 of the smoothing circuit 30 is different from the output target value Vi. It can also be operated to be more accurate. However, it is possible to operate without these.

In addition, depending on the configuration and operation of the control circuit 40, constant current output, constant impedance output, constant power output, etc., or overcurrent protection (foldback type, vertical type, etc.) may be optionally added. be.

In the above embodiments, the isolated bipolar bidirectional DC/DC converter operation and the isolated bidirectional DC/AC inverter operation are exemplified. It can be applied not only to the sexual movement form but also to the case where it is configured for the purpose of one movement form.

As described above, the embodiments and the like have been described, but the present invention is not limited to the above description, and the invention described in the claims or disclosed in the mode for carrying out the invention It goes without saying that various modifications and changes can be made by those skilled in the art based on the gist of the above, and such modifications and changes are included in the scope of the present invention.

[Comparison example of power loss in actual equipment]
37A and 37B are the results of measuring the output current i1 of the primary circuit 10 and the output voltage V1 of the primary circuit 10 with an oscilloscope. The output current i1 of the primary circuit 10 flows in the positive direction from the V1Hi side of the primary circuit 10 to the secondary circuit 20 .

In the measurement circuits of FIGS. 37A and 37B, in the power conversion device 100 corresponding to the example of the first embodiment, positive and negative signals are input from the outside instead of the signal from the current sensor CT as the determination result Sel. As a result, it is made possible to arbitrarily select from the outside whether the direction of the time difference _TD is advanced or delayed. Further, the time difference _TD is set to _TD = _Tp /8 with respect to the period _Tp of the reference signal, the output voltage of the smoothing circuit 30 is set to 1/2 of the maximum output voltage, and the direction of the time difference _TD is set to advance. A comparison was made between the case of taking it and the case of taking it late.

37A and 37B also, if the output current i1 of the primary circuit 10 and the output voltage V1 of the primary circuit 10 are both of the same sign, discharge occurs to the DC power supply 8, and the direction is opposite. If so, it will be charged. In FIGS. 37A and 37B, the waveform portion of the output voltage V1 of the primary side circuit 10 indicates in which direction the power supply current is discharged or charged.

The rise and rise of the waveform of the output current i1 are caused by the transition of the control signal Sig _S2 of the secondary circuit 20. FIG. Also, the rise and rise of the waveform of the output voltage V1 are caused by the transition of the control signal Sig _S1 of the primary side circuit 10 .

The time difference _TD is a waveform when it is advanced in FIG. 37A and delayed in FIG. 37B.

In each case, the power loss of the power conversion device 100 is obtained by measuring the power consumption of the entire device and subtracting the power consumption by the load 9. FIG. The load 9 is a resistive load of 1Ω, the applied voltage to the load 9 is a DC voltage of 10V, the current of 10A flows, and the power consumption by the load 9 is 100W.

The advance of FIG. 37A resulted in a power loss of 72.0 W and the lag of FIG. 37B resulted in 20.8 W.

That is, when the output voltage V1 of the primary circuit 10 transitions, if the direction of the current to the DC power supply 8 is controlled to change from discharging to charging, the direction of the current to the DC power supply 8 changes from charging to discharging. A power loss reduction of 51.2 W was observed compared to the variable control case.

[Summary]
In the power conversion device of the embodiment described above, by adopting a configuration in which the secondary side control signal is switched through the determination circuit according to the polarity of i2, effects such as reduction in power loss and reduction in switching noise can be obtained. .

It can be said that this indicates the superiority of the power conversion device of the embodiment described above over the conventional technology.

The power converters of the embodiments described above can be used for a wide range of applications described below and are useful.

(1) It can be used as an insulated DC/DC converter and used as a direct-current power supply capable of outputting both positive and negative polarities. For example, in a direct current motor, the positive and negative polarities of the output voltage can be used to control the direction of rotation, and the absolute value of the output voltage can be used to control the number of rotations. In this case, it is possible to regenerate surplus power due to the inertia of the DC motor, or to regenerate power using the DC motor as a generator during regenerative braking.

(2) It can be used as an isolated bi-directional DC/AC inverter to generate AC power from a battery or to charge a battery from AC power through bi-directionality.
(2-1) Using a power storage device such as a large-capacity battery or a flywheel, it can be used as a power consumption leveling device for a commercial power supply. (When the commercial power supply is insufficient, it operates as a DC/AC inverter and is used as a grid-connected inverter that supplies power from the battery to the commercial power supply. When the commercial power supply is somewhat surplus, it operates as an AC/DC converter. , used to charge the storage device from the commercial power supply.)
(2-2) Using a stationary battery, it can be used both as an emergency power supply and as a battery charger in the event of a power outage. (During a power outage, it is used as a DC/AC inverter and as an emergency power supply. Normally, it is used as an AC/DC converter to charge the battery from commercial power.)
(2-3) A plug-in hybrid vehicle, an electric vehicle, or the like can be used as a battery, and can be used as an emergency power source (system-connected inverter) in the event of a commercial power failure, and also as a battery charger using a commercial power source. The power conversion device can be installed either on the stationary side or on the vehicle side, but it is particularly advantageous when it is installed on the vehicle because it is an insulated type and the circuit configuration is simple and suitable for miniaturization and weight reduction. is large.
(2-4) In hybrid vehicles, electric vehicles, etc., supplying AC power of any voltage and frequency from the battery to the AC motor during acceleration, and using the AC motor as a generator during regenerative braking to supply DC power to the battery. can be regenerated and used to charge the battery.
(3) It can be used as an insulated DC/AC inverter, and can be used as a grid-connected inverter or power conditioner in wind power generation, fuel cell power generation, and solar cell power generation. (Furthermore, it is also possible to utilize the power regeneration function by using a battery, etc., to charge the battery when there is a surplus of generated power, and to supply power from the battery when there is a shortage of power.)

8 DC power supply 10 primary side circuits 16a, 16b capacitance element 20 secondary side circuit 30 smoothing circuit 31 inductor element 32 capacitance element 40 control circuit 41 primary side control circuit 42 secondary side control circuit , 43 decision circuits 100, 200, 210, 220 power converter 101 lead signal generation circuit 102, 102a lag signal generation circuit 103 lead/lag signal generation circuit 104 lead/lag signal generation circuit 108 triangular wave generation means 110 dead time generation circuit; 111 resistive element, 112 electrostatic capacitive element, 113 resistive element, 114 electrostatic capacitive element

Claims (35)

A power converter,
a primary side circuit that outputs an alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a DC power supply;
A voltage based on the output of the primary side circuit is input, and by switching the conduction state of the secondary side switch element, a voltage having the same polarity as the input voltage or having a polarity opposite to the input voltage is output. a secondary circuit for switching whether to output a voltage of
a judgment circuit for judging the polarity of the current at the output terminal of the DC power supply, the output terminal of the primary side circuit, the output terminal of the secondary side circuit, or the output terminal of the power converter, wherein the judgment of the judgment circuit and a control circuit for switching whether to advance or delay switching timing of the conduction state of the secondary side switch element with respect to switching timing of the conduction state of the primary side switch element according to a result . A power converter,
a primary side circuit that outputs an alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a DC power supply;
A voltage based on the output of the primary side circuit is input, and by switching the conduction state of the secondary side switch element, a voltage having the same polarity as the input voltage or having a polarity opposite to the input voltage is output. a secondary circuit for switching whether to output a voltage of
a holding circuit for holding a determination result of current polarity at the output terminal of the DC power supply, the output terminal of the primary circuit, the output terminal of the secondary circuit, or the output terminal of the power conversion device; a control circuit for switching, according to the output of the holding circuit, whether to advance or delay the switching timing of the conduction state of the secondary side switch element with respect to the switching timing of the conduction state of the secondary side switching element;
A power conversion device comprising: A power converter,
a primary side circuit that outputs an alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a DC power supply;
A voltage based on the output of the primary side circuit is input, and by switching the conduction state of the secondary side switch element, a voltage having the same polarity as the input voltage or having a polarity opposite to the input voltage is output. a secondary circuit for switching whether to output a voltage of
The output terminal of the DC power supply, the output terminal of the primary side circuit, and the timing of switching the conduction state of the secondary side switching element to advance or delay the switching timing of the conduction state of the primary side switching element are determined by the output terminal of the DC power supply, the output terminal of the primary side circuit, a control circuit that switches according to the polarity of the current at the output terminal of the secondary circuit or at the output terminal of the power converter;
with
The control circuit controls the primary circuit according to the determination result of the polarity of the current at the output terminal of the DC power supply, the output terminal of the primary circuit, the output terminal of the secondary circuit, or the output terminal of the power converter. A signal having an advance time difference or a delay time difference with respect to a primary side control signal indicating switching timing of the conductive state of the secondary side switch element is used as a signal indicating switching timing of the conductive state of the secondary side switch element. Output to circuit
Power converter. The primary side circuit includes a first switching element and a second switching element connected in series between both ends of the DC power supply, and a first capacitance element and a first switching element connected in series between both ends of the DC power supply. 2 capacitive elements, and a half bridge circuit,
one end of the output of the primary circuit is connected to a connection point between the first switch element and the second switch element;
The power conversion according to any one of claims 1 to 3 , wherein the other end of the output of the primary side circuit is connected to a connection point between the first capacitance element and the second capacitance element. Device. 5. The power converter according to claim 4 , wherein the capacitance value of said first capacitance element is substantially equal to the capacitance value of said second capacitance element. The primary side circuit includes a first switch element and a second switch element connected in series between both ends of the DC power supply, and a static switch connected to one end of the DC power supply and one end of the output of the primary side circuit. A half bridge circuit comprising a capacitive element,
The power converter according to any one of claims 1 to 3 , wherein the other end of the output of the primary side circuit is connected to a connection point between the first switch element and the second switch element. The DC power supply comprises a first DC power supply and a second DC power supply connected in series,
The primary side circuit is a half bridge circuit comprising a first switch element and a second switch element connected in series between both ends of the first DC power supply and the second DC power supply connected in series,
one end of the output of the primary circuit is connected to a connection point between the first switch element and the second switch element;
The power converter according to any one of claims 1 to 3 , wherein the other end of the output of the primary circuit is connected to a connection point between the first DC power supply and the second DC power supply. The DC power supply comprises a first DC power supply and a second DC power supply connected in series,
The primary side circuit includes a first switch element and a second switch element connected in series between both ends of the first DC power supply and the second DC power supply connected in series, and the first DC power supply and the A half bridge circuit comprising a capacitance element connected between a connection point with a second DC power supply and one end of the output of the primary circuit,
The power converter according to any one of claims 1 to 3 , wherein the other end of the output of the primary side circuit is connected to a connection point between the first switch element and the second switch element. The primary side circuit includes a first switch element and a second switch element connected in series between both ends of the DC power supply, and a third switch element and a fourth switch element connected in series between both ends of the DC power supply. A full bridge circuit comprising:
one end of the output of the primary circuit is connected to a connection point between the first switch element and the second switch element;
The power converter according to any one of claims 1 to 3 , wherein the other end of the output of the primary side circuit is connected to a connection point between the third switch element and the fourth switch element. Between a connection point between the first switch element and the second switch element and one output end of the primary side circuit, and between a connection point between the third switch element and the fourth switch element and the primary side circuit. 10. The power converter according to claim 9 , comprising a capacitance element provided between at least one of the other end of the output of the side circuit and the other end of the output of the side circuit. The secondary circuit is
A transformer comprising a primary winding to which the output of the primary side circuit is input, and a first secondary winding and a second secondary winding,
The secondary side switch element is
Between one end of the output of the secondary circuit and one end of the first secondary winding, or between the other end of the output of the secondary circuit and the other end of the first secondary winding a first bidirectional switch element provided therebetween;
Between one end of the output of the secondary circuit and one end of the second secondary winding, or between the other end of the output of the secondary circuit and the other end of the second secondary winding a second bidirectional switch element provided therebetween;
The secondary circuit switches one end and the other end of the first secondary winding to the secondary side by switching the conductive state of the first bidirectional switch element and the second bidirectional switch element. from claim 1, switching between connecting one end and the other end of the output of the circuit and connecting one end and the other end of the second secondary winding to the one end and the other end of the output of the secondary circuit, respectively; 11. The power converter according to any one of 10 . The secondary circuit is
A transformer comprising a primary winding to which the output of the primary side circuit is input, and a first secondary winding and a second secondary winding,
The secondary side switch element is
a first unidirectional switch element provided between one end of the output of the secondary circuit and one end of the first secondary winding; and a reverse polarity parallel parallel to the first unidirectional switch element. a switch element comprising a connected first diode element;
a second unidirectional switch element provided between the other end of the output of the secondary circuit and the other end of the first secondary winding; and a polarity opposite to the second unidirectional switch element. a switch element comprising a second diode element connected in parallel with
a third unidirectional switch element provided between one end of the output of the secondary circuit and one end of the second secondary winding; a switch element comprising a connected third diode element;
a fourth unidirectional switch element provided between the other end of the output of the secondary circuit and the other end of the second secondary winding; and a polarity opposite to the fourth unidirectional switch element. A switch element comprising a fourth diode element connected in parallel with
the first unidirectional switch element and the second unidirectional switch element are connected in series with opposite polarities across the first secondary winding;
11. The third unidirectional switch element and the fourth unidirectional switch element are connected in series with opposite polarities across the second secondary winding. The power conversion device according to . The secondary circuit is
A transformer comprising a primary winding to which the output of the primary side circuit is input, and a secondary winding,
The secondary side switch element is
a first bidirectional switch element provided between one end of the output of the secondary circuit and one end of the secondary winding;
a second bidirectional switch element provided between the other end of the output of the secondary circuit and the other end of the secondary winding;
a third bidirectional switch element provided between one end of the output of the secondary circuit and the other end of the secondary winding;
The power converter according to any one of claims 1 to 10 , further comprising a fourth bidirectional switch element provided between the other end of the output of the secondary side circuit and one end of the secondary winding. Device. A smoothing circuit connected to the output side of the secondary circuit,
14. The power converter according to any one of claims 1 to 13 , wherein the smoothing circuit comprises an inductor element connected between one end of the input of the smoothing circuit and one end of the output of the smoothing circuit. 15. The power conversion device according to claim 14 , wherein said smoothing circuit comprises a capacitance element connected in parallel to the output of said smoothing circuit. The power converter according to any one of claims 1 to 15 , wherein the load connected to the output of said power converter is a resistive load. The power converter according to any one of claims 1 to 15 , wherein the load connected to the output of the power converter is a load that supplies power to the power converter. 16. The load connected to the output of the power conversion device is a load that consumes power output from the power conversion device and supplies power to the power conversion device. power converter. A power converter,
a primary side circuit that outputs an alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a DC power supply;
A voltage based on the output of the primary side circuit is input, and by switching the conduction state of the secondary side switch element, a voltage having the same polarity as the input voltage or having a polarity opposite to the input voltage is output. a secondary circuit for switching whether to output a voltage of
The output terminal of the DC power supply, the output terminal of the primary side circuit, and the timing of switching the conduction state of the secondary side switching element to advance or delay the switching timing of the conduction state of the primary side switching element are determined by the output terminal of the DC power supply, the output terminal of the primary side circuit, a control circuit that switches according to the polarity of the current at the output terminal of the secondary circuit or at the output terminal of the power converter;
with
The control circuit is
a primary side control circuit that outputs a reference signal indicating switching timing of the conduction state of the primary side switch element to the primary side circuit;
a determination circuit that determines the polarity of the current at the output terminal of the DC power supply, the output terminal of the primary circuit, the output terminal of the secondary circuit, or the output terminal of the power converter;
A signal having an advance time difference or a delay time difference with respect to the reference signal is output to the secondary circuit as a signal indicating switching timing of the conduction state of the secondary switch element according to the decision result of the decision circuit. A secondary side control circuit
Power converter. A power converter,
a primary side circuit that outputs an alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a DC power supply;
A voltage based on the output of the primary side circuit is input, and by switching the conduction state of the secondary side switch element, a voltage having the same polarity as the input voltage or having a polarity opposite to the input voltage is output. a secondary circuit for switching whether to output a voltage of
The output terminal of the DC power supply, the output terminal of the primary side circuit, and the timing of switching the conduction state of the secondary side switching element to advance or delay the switching timing of the conduction state of the primary side switching element are determined by the output terminal of the DC power supply, the output terminal of the primary side circuit, a control circuit that switches according to the polarity of the current at the output terminal of the secondary circuit or at the output terminal of the power converter;
with
The control circuit is
a determination circuit that determines the polarity of the current at the output terminal of the DC power supply, the output terminal of the primary circuit, the output terminal of the secondary circuit, or the output terminal of the power converter;
a primary side control circuit for outputting a reference signal or a signal obtained by inverting the reference signal to the primary side circuit as a primary side control signal for controlling the primary side switch element according to the determination result of the determination circuit; ,
A signal obtained by inverting a signal having a lead time difference with respect to the reference signal or a signal having a lag time difference with respect to the reference signal according to the decision result of the decision circuit is used as timing for switching the conductive state of the secondary side switch element. and a secondary side control circuit that outputs to the secondary side circuit as a signal indicating
Power converter. 21. A power converter according to any one of claims 1 to 20 , wherein at least part of said control circuit is implemented by a programmable device or processor. a current sensor connected to the output side of the secondary circuit and detecting a current at the output end of the secondary circuit;
The control circuit advances or delays switching timing of the conduction state of the secondary side switch element with respect to switching timing of the conduction state of the primary side switch element according to the polarity of the current detected by the current sensor. 22. The power converter according to any one of claims 1 to 21 , wherein switching is performed. a smoothing circuit connected to the output side of the secondary circuit for smoothing the output voltage of the secondary circuit;
A current sensor that detects the current at the output end of the smoothing circuit,
The control circuit advances or delays switching timing of the conduction state of the secondary side switch element with respect to switching timing of the conduction state of the primary side switch element according to the polarity of the current detected by the current sensor. 22. The power converter according to any one of claims 1 to 21 , wherein switching is performed. a smoothing circuit connected to the output side of the secondary circuit for smoothing the output voltage of the secondary circuit;
A voltage sensor that detects the voltage at the output end of the smoothing circuit,
The control circuit advances or delays switching timing of the conduction state of the secondary side switch element with respect to switching timing of the conduction state of the primary side switch element according to the polarity of the voltage detected by the voltage sensor. 22. A power converter as claimed in claim 16 , or claim 21 depending on claim 16 , which switches. A current sensor that detects a current between the output terminal of the DC power supply or the output terminal of the primary circuit and the secondary circuit,
The control circuit advances or delays switching timing of the conduction state of the secondary side switch element with respect to switching timing of the conduction state of the primary side switch element according to the polarity of the current detected by the current sensor. 22. The power converter according to any one of claims 1 to 21 , wherein switching is performed. A power converter,
a primary side circuit that outputs an alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a DC power supply;
A voltage based on the output of the primary side circuit is input, and by switching the conduction state of the secondary side switch element, a voltage having the same polarity as the input voltage or having a polarity opposite to the input voltage is output. a secondary circuit for switching whether to output a voltage of
The output terminal of the DC power supply, the output terminal of the primary side circuit, and the timing of switching the conduction state of the secondary side switching element to advance or delay the switching timing of the conduction state of the primary side switching element are determined by the output terminal of the DC power supply, the output terminal of the primary side circuit, a control circuit that switches according to the polarity of the current at the output terminal of the secondary circuit or at the output terminal of the power converter;
with
The load connected to the output of the power conversion device is a resistive load,
The control circuit is
a primary side control circuit that generates a primary side control signal indicating switching timing of the conduction state of the primary side switch element based on a reference signal;
a determination circuit that determines the polarity of an external signal indicating the output target value of the power conversion device;
A signal having an advance time difference or a delay time difference with respect to the reference signal is output to the secondary circuit as a signal indicating switching timing of the conduction state of the secondary switch element according to the decision result of the decision circuit. A power conversion device comprising a secondary side control circuit. A low-pass filter that attenuates high-frequency components of the output signal of the current sensor,
26. A power converter according to any one of claims 22 , 23 , or 25 , wherein the control circuit determines the polarity of the current based on the output of the low pass filter. A low-pass filter that attenuates high-frequency components of the output signal of the voltage sensor,
25. The power converter of claim 24 , wherein the control circuit determines the polarity of the current based on the output of the low pass filter. The control circuit is
29. Any one of claims 1 to 28 , comprising a hysteresis comparator that determines the polarity of the current at the output terminal of the DC power supply, the output terminal of the primary circuit, the output terminal of the secondary circuit, or the output terminal of the power converter. or the power converter according to claim 1. A power converter,
a primary side circuit that outputs an alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a DC power supply;
A voltage based on the output of the primary side circuit is input, and by switching the conduction state of the secondary side switch element, a voltage having the same polarity as the input voltage or having a polarity opposite to the input voltage is output. a secondary circuit for switching whether to output a voltage of
The output terminal of the DC power supply, the output terminal of the primary side circuit, and the timing of switching the conduction state of the secondary side switching element to advance or delay the switching timing of the conduction state of the primary side switching element are determined by the output terminal of the DC power supply, the output terminal of the primary side circuit, a control circuit that switches according to the polarity of the current at the output terminal of the secondary circuit or at the output terminal of the power converter;
with
The control circuit is
a comparator that determines the polarity of the current at the output end of the secondary circuit or the output end of the power conversion device;
a holding circuit that holds the output of the comparator or the inversion of the output of the comparator according to the output of the comparator at the switching timing of the conduction state of the primary side switch element;
Based on the output of the holding circuit, whether to advance or delay the switching timing of the conduction state of the secondary side switch element with respect to the switching timing of the conduction state of the primary side switching element is determined according to the output of the holding circuit. A power conversion device comprising a secondary side control circuit that switches by. A power converter,
a primary side circuit that outputs an alternating current by switching the conduction state of a primary side switch element connected in series between both ends of a DC power supply;
A voltage based on the output of the primary side circuit is input, and by switching the conduction state of the secondary side switch element, a voltage having the same polarity as the input voltage or having a polarity opposite to the input voltage is output. a secondary circuit for switching whether to output a voltage of
The output terminal of the DC power supply, the output terminal of the primary side circuit, and the timing of switching the conduction state of the secondary side switching element to advance or delay the switching timing of the conduction state of the primary side switching element are determined by the output terminal of the DC power supply, the output terminal of the primary side circuit, a control circuit that switches according to the polarity of the current at the output terminal of the secondary circuit or at the output terminal of the power converter;
with
The control circuit is
a comparator that determines the polarity of the current at the output terminal of the DC power supply or the output terminal of the primary circuit;
a holding circuit that holds the output of the comparator;
Based on the output of the holding circuit, whether to advance or delay the switching timing of the conduction state of the secondary side switch element with respect to the switching timing of the conduction state of the primary side switching element is determined according to the output of the holding circuit. A power conversion device comprising a secondary side control circuit that switches by. The secondary circuit is
a transformer including a primary winding to which the output of the primary circuit is input, a secondary winding connected to the secondary switching element, and one or more tertiary windings;
One or more power supply circuits formed by the one or more tertiary windings and one or more rectifier circuits connected to the one or more tertiary windings,
32. A power converter as claimed in any one of claims 1 to 31 . A driver that drives at least one of the control circuit that operates with power supplied from at least one of the one or more power supply circuits, or the primary side switch element and the secondary side switch element. 33. A power converter according to Item 32 . 33. A power converter as claimed in claim 32 when dependent on claim 22 , 23 or 25 , wherein the current sensor operates on power supplied from at least one of the one or more power supply circuits. 33. A power converter as claimed in claim 32 when dependent on claim 24 , wherein the voltage sensor operates on power supplied from at least one of the one or more power supply circuits.

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