JP7305114B2 - Power conversion device and its control device - Google Patents

Description

The invention disclosed in this specification relates to a power conversion device and its control device.

FIG. 11 is a diagram showing a conventional example of a PFC (power factor correction) circuit that converts AC power into DC power while improving the power factor. As shown in this figure, as the control body of the PFC circuit, a PI [proportional-integral] control type analog control device X that applies negative feedback according to each of the output voltage, the output current, and the input voltage is common. be.

JP 2005-218252 A

However, since the conventional analog control device X employs the PI control method, the vibrations until reaching the steady state become large, making control difficult. Also, since there are a voltage loop and a current loop, two compensators (amplifiers X1 and X2) are required, resulting in a large circuit scale. Furthermore, in a diode-bridgeless high-power PFC circuit, it is difficult to generate a control signal corresponding to an input voltage from a simple resistive voltage division. Therefore, it is necessary to provide a commercial frequency transformer between the input voltage application end and the analog control device X, which is not suitable for miniaturization and cost reduction. As long as the PI control system is employed, the above problem cannot be solved by simply replacing the analog control device X with a digital control device.

On the other hand, as a conventional technique for solving the above problem, an analog control device that detects input current instead of input voltage and applies negative feedback has been proposed (see, for example, Patent Document 1). Certainly, according to this prior art, it is possible to omit a compensator for current control, and a control signal corresponding to the input voltage is also unnecessary, so it is advantageous in terms of feedback loop adjustment and circuit scale. .

However, the analog control device of Patent Document 1 is only for the PFC circuit. /AC conversion operation (circuit for realizing both INV [inverter] operation)), it could not be applied as it is.

In view of the above-described problems found by the inventors of the present application, the invention disclosed in the present specification controls both an AC/DC conversion circuit (PFC circuit) and a DC/AC conversion circuit (inverter). It is an object of the present invention to provide a control device and a power conversion device using the same.

The control device disclosed in the present specification is used as a control subject of a power conversion device including a switch circuit including a transistor, and a reference signal set according to the operation method of the power conversion device A multiplied signal obtained by multiplying the controlled current of the switch circuit by a predetermined coefficient is subtracted from the current to generate the control signal for the transistor based on the result of the calculation (first configuration).

In the control device having the first configuration, the switch circuit may be a PFC circuit, the reference signal may be a constant value, and the controlled current may be an input current (second configuration).

Further, the control device having the first configuration may be configured so that the switch circuit is an inverter, the reference signal is a sinusoidal signal, and the current to be controlled is an output current (third configuration). .

In the control device having the first configuration, the switch circuit is a bi-directional inverter, and the reference signal has a constant value during AC/DC conversion, and the reference signal becomes a sinusoidal signal during DC/AC conversion. Further, the waveform of the reference signal is changed when the direction of current flow is switched, and the current flowing through the same node of the switch circuit is monitored as the current to be controlled during both AC/DC conversion and DC/AC conversion. The current to be controlled may be the input current during /DC conversion, and the current to be controlled may be the output current during DC/AC conversion (fourth configuration).

Further, the control device having the second or fourth configuration may be configured to superimpose a modulation signal for distorting the input current on the reference signal (fifth configuration).

Further, the control device having the third or fourth configuration may be configured to superimpose a modulation signal on the reference signal for canceling the distortion of the load current flowing from the commercial power supply to another load (sixth configuration). .

Further, in the control device having the third or fourth configuration, the frequency of the reference signal is set to the frequency required for the output current, and the coefficient is such that the multiplied signal can be ignored compared to the reference signal. A configuration (seventh configuration) that is set to be as small as possible may be used.

Also disclosed herein is a bi-directional inverter comprising: a capacitor bridge including at least two capacitors connected in series between a first DC node and a second DC node; said first DC node and said second DC node; a first transistor bridge and a second transistor bridge each comprising two transistors connected in series between; a midpoint node of said capacitor bridge and an output node of each of said first transistor bridge and said second transistor bridge; a first bidirectional switch and a second bidirectional switch respectively connected between; a transformer connected to an output node of each of said first transistor bridge and said second transistor bridge; a transformer connected to an AC node; a reactor connected between; a capacitor connected between the AC node and a midpoint node of the capacitor bridge; is configured to operate with (eighth configuration).

In the bidirectional inverter having the eighth configuration, the first transistor bridge and the second transistor bridge, the first bidirectional switch and the second bidirectional switch, the transformer, the reactor, and the capacitor should be a three-phase configuration (ninth configuration).

Further, in the bidirectional inverter having the eighth or ninth configuration, the transformer and the reactor may be configured as a transformer coupling reactor having both functions (tenth configuration).

Further, in the bidirectional inverter having any one of the eighth to tenth configurations, the transistors, the first bidirectional switch, and the second bidirectional switch are each made of a wide bandgap semiconductor (eleventh configuration).

Further, the power converter disclosed in this specification includes a bidirectional inverter having any one of the eighth to eleventh configurations, and a bidirectional inverter having any one of the first to eighth configurations. and a control device serving as a control subject (a twelfth configuration).

According to the invention disclosed in this specification, a control device capable of controlling both an AC/DC conversion circuit (PFC circuit) and a DC/AC conversion circuit (inverter) and a power converter using the same can be provided.

The figure which shows an example of 1st Embodiment of a power converter device. Diagram showing an example of PFC operation (AC→DC) Diagram showing input/output waveforms during PFC operation Diagram showing an example of INV operation (DC→AC) Diagram showing input/output waveforms during INV operation FIG. 4 is a diagram showing a first modulation example of the reference signal; FIG. 10 is a diagram showing a second modulation example of the reference signal; The figure which shows 2nd Embodiment of a power converter device The figure which shows the example of a changed completely type of 2nd Embodiment. The figure which shows one structural example of the control apparatus used by 2nd Embodiment. A diagram showing a conventional example of a PFC circuit

<Power converter (first embodiment)>
FIG. 1 is a diagram showing an example of a first embodiment of a power converter. The power conversion device 1 of the first embodiment has a switch circuit 10 , a drive circuit 20 and a digital control device 100 .

The switch circuit 10 is a bidirectional inverter that realizes both AC/DC conversion operation (PFC operation) and DC/AC conversion operation (INV operation) by switching its input/output (=conducting direction). . . . M4 (both N-channel in this figure), capacitors C1 and C2, and inductor L1.

The drains of transistors M1 and M3 are connected to node DC1. The source of transistor M1 and the drain of transistor M2 are connected to the first end of inductor L1. A second end of inductor L1 is connected to node AC1. The source of transistor M3 and the drain of transistor M4 are connected to node AC2. The sources of transistors M2 and M4 are connected to node DC2. Capacitor C1 is connected between node AC1 and node AC2. Capacitor C2 is connected between node DC1 and node DC2.

Gate signals G1 to G4 are input to the gates of the transistors M1 to M4, respectively. The transistors M1 to M4 are turned on when the gate signals G1 to G4 are at high level, and turned off when the gate signals G1 to G4 are at low level. The transistors M1 and M2 are complementarily turned on/off according to gate signals G1 and G2, respectively. Also, the transistors M3 and M4 are complementarily turned on/off according to the gate signals G3 and G4, respectively. The term “complementary” in this specification is used not only when the on/off states of the transistors are completely reversed, but also when the simultaneous off period of each transistor (so-called dead time) is used from the viewpoint of preventing through current. ) is provided.

The drive circuit 20 is a circuit block that generates gate signals G1-G4 according to the control signals S1-S4, and includes gate drivers 21-24. The gate drivers 21-24 increase the current capability of the control signals S1-S4 to generate the gate signals G1-G4, respectively.

The digital control device 100 is the control body of the switch circuit 10 (and the power conversion device 1 as a whole), and its various functional units include a reference signal setting unit 101, a coefficient setting unit 102, a multiplication unit 103, and an addition unit. It includes a section 104 , a pulse width modulation section 105 , a zero cross detection section 106 and a signal switching section 107 . The functional blocks described above are implemented in software by executing a control program in the digital control device 100 . Alternatively, the digital controller 100 may be replaced with an analog controller having equivalent functionality.

The reference signal setting unit 101 sets the reference signal REF according to the operation mode setting signal MODE (=control signal for switching the operation method of the power conversion device 1 to either the PFC operation or the INV operation).

Coefficient setting section 102 sets coefficient K and outputs it to multiplication section 103 .

Multiplication section 103 multiplies current I to be controlled flowing through switch circuit 10 (node AC2 in the figure) by coefficient K and outputs a multiplication signal (=K×I). Note that the digital control device 100 uses the same node (node AC2 in this figure) of the switch circuit 10 as the current I to be controlled, both during the PFC operation (during AC/DC conversion) and during the INV operation (during DC/AC conversion). ) is monitored. Although details will be described later, during PFC operation (AC/DC conversion), the controlled current I becomes the input current Iin, and during INV operation (DC/AC conversion), the controlled current I becomes the output current Iout. .

The adder 104 (subtracter in the example of this drawing) subtracts the multiplied signal (=K×I) from the reference signal REF and outputs a difference signal (=REF−K×I).

The pulse width modulation unit 105 compares the differential signal (=REF−K×I) with a triangular or sawtooth slope signal (not shown) to generate a pulse width modulated signal PWM (and its logic level is inverted). output an inverted pulse width modulated signal PWMB).

The zero-cross detection unit 106 detects polarity reversal timing (=zero-cross timing) of an alternating voltage (=input voltage Vin during PFC operation, output voltage Vout during INV operation) applied between nodes AC1 and AC2, and synchronizes with this. zero-cross signal ZX (and an inverted zero-cross signal ZXB whose logic level is inverted). More specifically, the zero-cross signal ZX, for example, becomes high level when the AC voltage is positive, and becomes low level when the AC voltage is negative. Therefore, if the AC voltage is a commercial AC voltage in Japan, the zero-cross signal ZX will be a pulse signal of 50 Hz or 60 Hz. One of the zero-cross signal ZX and the inverted zero-cross signal ZXB is output as the control signal S3, and the other is output as the control signal S4. The polarity reversal timing of the AC voltage may be detected using a photocoupler or the like.

The signal switching unit 107 outputs one of the pulse width modulated signal PWM and the inverted pulse width modulated signal PWMB as the control signal S1 according to the zero-crossing signal ZX. The signal switching unit 107 also outputs a control signal S2 (=S1B) obtained by inverting the logic level of the control signal S1. Thus, for example, when ZX=L, S1=PWM and S2=PWMB. On the other hand, when ZX=H, S1=PWMB and S2=PWM.

In this way, the digital control device 100 multiplies the controlled current I of the switch circuit 10 by the predetermined coefficient K from the reference signal REF set according to the operation mode (operation mode setting signal MODE) of the power conversion device 1. The multiplied signal (=K×I) is subtracted, and the control signals S1 and S2 for the transistors M1 and M2 are generated based on the operation result (=REF−K×I). Below, the PFC operation and the INV operation in the power conversion device 1 will be individually and specifically described.

<PFC operation (AC→DC)>
FIG. 2 is a diagram showing an example of PFC operation (AC→DC) in the power converter 1. In FIG. In the figure, an AC power supply E1 that supplies AC power (input voltage Vin, input current Iin) is connected between a node AC1 and a node AC2. A DC load Z1 that receives DC power (output voltage Vout, output current Iout) is connected between the node DC1 and the node DC2. At this time, the switch circuit 10 functions as a PFC circuit (boost converter) that converts AC power into DC power.

As shown in the figure, during the PFC operation, the reference signal REF is set to a constant value, and the input current |Iin| (=absolute value of the input current Iin) is input as the current to be controlled. Also, the coefficient K is variably controlled so that the output voltage Vout has a constant value. More specifically, the coefficient K may be variably controlled according to, for example, the difference value between the resistor voltage dividing value of the output voltage Vout and a predetermined output command value.

Also, during PFC operation, only the transistors M1 and M2 are PWM-operated, and the transistors M3 and M4 are turned off and used as diodes, so that the current direction can be made constant and reverse current can be prevented. For improved efficiency, transistors M3 and M4 are operated at a lower frequency. In that case, the zero-cross signal ZX is output as the control signal S4, and the inverted zero-cross signal ZXB is output as the control signal S3. As a result, one of the transistors M3 and M4 (only the switch for storing power in the reactor L1) is activated.

Next, the principle of PFC operation will be described. Let Vin be the input voltage, Vout be the output voltage, T be the switching period, and Ton and Toff be the on-time and off-time of the transistor, respectively. off), the following equations (1) to (3) hold.

Vin×Ton=(Vout−Vin)×Toff … (1)
Vin×(T−Toff)=(Vout−Vin)×Toff … (2)
Vin×T=Vout×Toff (3)

From the above equation (3), it can be seen that if the switching period T and the output voltage Vout are constant, the input voltage Vin is proportional to the OFF time Toff. Here, if the input current Iin has the same waveform as the input voltage Vin, the input current Iin should also be proportional to the OFF time Toff. Therefore, by controlling the input current Iin in proportion to the OFF time Toff, it is possible to obtain the input current Iin having the same waveform as the input voltage Vin. Further, by setting the coefficient K so that the output voltage Vout is a constant value, the PFC operation can be realized.

FIG. 3 is a diagram showing input/output waveforms during PFC operation, in which the input voltage Vin, the input current Iin, and the output voltage Vout are depicted in order from the top. As shown in the figure, it can be seen that the PFC operation of the power converter 1 converts AC power into DC power.

<INV operation (DC→AC)>
FIG. 4 is a diagram showing an example of the INV operation (DC→AC) in the power converter 1. In FIG. In the figure, a DC power supply E2 that supplies DC power (input voltage Vin, input current Iin) is connected between nodes DC1 and DC2. An AC load Z2 that receives supply of AC power (output voltage Vout, output current Iout) is connected between the node AC1 and the node AC2. That is, the inputs and outputs of the switch circuit 10 are reversed from those in FIG. At this time, the switch circuit 10 functions as an inverter that converts DC power into AC power.

As shown in this figure, during the INV operation, for example, the reference signal REF is set to a sinusoidal signal (which may be a sinusoidal signal obtained by full-wave rectification), and the output current |Iout| (=output current absolute value of Iout) is input. In the case of a regenerative inverter that regenerates power to an AC power supply, the reference signal REF may be set in synchronization with the zero-cross signal ZX. On the other hand, in the case of a stand-alone inverter, a sinusoidal signal of a desired frequency should be generated. Further, the coefficient K is variably controlled so that the output voltage |Vout|_ave (=absolute average value of the output voltage Vout) is constant. More specifically, the coefficient K may be variably controlled according to, for example, the difference between the average value of the full-wave rectified output voltage Vout and a predetermined output command value. During the INV operation, the inverted zero-cross signal ZXB is output as the control signal S3, and the zero-cross signal ZX is output as the control signal S4. This is an example of operation, and is not limited to this method as long as the control is such that the output becomes a sine wave.

Next, the principle of INV operation will be described. Let Vin be the input voltage, Vout be the output voltage, T be the switching period, and Ton and Toff be the on-time and off-time of the transistor, respectively. off), the following equations (4) to (6) hold.

(Vin−Vout)×Ton=Vout×Toff … (4)
(Vin−Vout)×Ton=Vout×(T−Ton) … (5)
Vin×Ton=Vout×T (6)

From the above equation (6), it can be seen that if the switching period T and the input voltage Vin are constant, the output voltage Vout is proportional to the ON time Ton. Therefore, in the case of an independent I inverter, if the output current Iout has the same waveform as the output voltage Vout, the output current Iout should also be proportional to the ON time Ton. Therefore, if the output current Iout is controlled in proportion to the ON time Ton, it seems that the output current Iout having the same waveform as the output voltage Vout can be obtained. However, in such control, when the output current Iout increases, the on-time Ton also increases in proportion, resulting in positive feedback control, making the INV operation impossible.

Therefore, when the output current Iout increases, it is necessary to shorten the on-time Ton. In order to realize such negative feedback control, a reference signal REF having the same waveform as the output voltage Vout (a sine wave if the output voltage Vout is a sine wave) is set in advance, and the output current is calculated from the reference signal REF. The on-time Ton can be controlled based on the difference signal (=REF-K×Iout) obtained by subtracting the multiplication signal (=K×Iout) proportional to Iout.

The reference signal REF is a sine wave. Also, if the output current Iout is a sine wave, the multiplication signal (=K×Iout) proportional thereto is also a sine wave. Therefore, the difference signal (=REF−K×Iout) obtained by subtracting the multiplication signal (=K×Iout) from the reference signal REF also becomes a sine wave, so the output current Iout controlled based on this becomes a sine wave. . In this case, the ON time Ton is shortened when the output current Iout increases. Therefore, since negative feedback control is realized, stable INV operation becomes possible. Also, by changing the coefficient K, the output current Iout can be adjusted.

When the power converter 1 is used as a regenerative inverter (=system-connected inverter connected to a commercial AC power supply system), zero and polarity of AC voltage are detected to generate a reference signal REF for operation. Also, when soft-starting the output voltage of the regenerative inverter, by operating only the switch that sends out the power and keeping the other switches off, the current can be safely started without reverse flow. Loss can be reduced by synchronously rectifying other switches after the output voltage reaches a predetermined voltage (=connected AC voltage value).

On the other hand, when the power converter 1 is used as a stand-alone inverter (=a general inverter connected to the AC load Z2), the frequency of the reference signal REF is required for the output current Iout according to the specifications of the AC load Z2. It should be set to a frequency that Also, the coefficient K may be set to a value (for example, K=0) at which the multiplied signal (=K×Iout) becomes negligibly small compared to the reference signal REF.

FIG. 5 is a diagram showing input/output waveforms during the INV operation, and depicts the input voltage Vin, the output voltage Vout, and the output current Iout in order from the top. As shown in the figure, it can be seen that the INV operation of the power converter 1 converts DC power into AC power.

<Operation mode switching (PFC/INV)>
As described above, according to the operation mode setting signal MODE, the digital control device 100 keeps the reference signal REF at a constant value during the PFC operation of the switch circuit 10 (during AC/DC conversion), and during the INV operation ( It has a function of dynamically changing the waveform of the reference signal REF when switching the energization direction so that the reference signal REF becomes a sinusoidal signal during DC/AC conversion).

By providing such a function, the switch circuit 10 can be operated as a bi-directional inverter. On the contrary, it is possible to convert the DC power generated by the solar cell into AC power and regenerate it to the commercial AC power supply. It also supports parallel operation and hot swapping.

However, the operation mode does not necessarily have to be dynamically switched. For example, when the switch circuit 10 is used only as a PFC circuit, the reference signal REF may be fixed at a constant value. It should be fixed to a wave signal.

<Modulation processing of reference signal>
FIG. 6 is a diagram showing a first modulation example of the reference signal REF during PFC operation (during AC/DC conversion), in which the reference signal REF and the input current Iin are depicted in order from the top. The left side of the drawing depicts the waveform when the reference signal REF is not modulated, and the right side of the drawing depicts the waveform when the reference signal REF is modulated.

When it is desired to cause an intentional distortion component in the input current Iin, a modulation signal (=modulation component corresponding to the difference between the sinusoidal current waveform and the desired current waveform) for giving the distortion component to the input current Iin is applied. It may be superimposed on the reference signal REF. According to such modulation processing, it is possible to use the power conversion device 1 as an AC electronic load device used in, for example, a pseudo-current load test.

FIG. 7 is a diagram showing a second modulation example of the reference signal REF during the INV operation (especially when the regenerative inverter is used as the distortion correction device). The input current Iin applied, the load current Iload flowing to other loads, and the total supply current Isup from the mains supply are depicted. The left side of the drawing depicts the waveform when the reference signal REF is not modulated, and the right side of the drawing depicts the waveform when the reference signal REF is modulated.

If an unintended distortion component occurs in the load current Iload, a modulation signal for canceling the distortion component (=a modulation component for causing a current opposite to the above distortion component) is superimposed on the reference signal REF and input. If the current Iin is also intentionally distorted and the input current Iin and the load current Iload are added, the total supply current Isup is a sinusoidal current. According to such modulation processing, for example, it is possible to suppress harmonic components of the load current Iload so as not to output a distorted current to the outside.

<Power Converter (Second Embodiment)>
FIG. 8 is a diagram showing a second embodiment of the power converter. In the power converter 11 of the second embodiment, a three-phase TL-NPC [Trans-Linked Neutral-Point-Clamped] type bidirectional inverter is used as the switch circuit 10 .

Specifically, the switch circuit 10 includes, for example, capacitors C11 and C12, three-phase switch circuit units 11 to 13, and capacitors C21 to C23.

The switch circuit unit 11 includes transistors M11 and M12 (N-channel type), transistors M21 and M22 (N-channel type), bidirectional switches SW1 and SW2 (N-channel type), a transformer TR11, and a reactor L11. and including.

Note that the circuit configurations of the switch circuit units 12 and 13 are the same as that of the switch circuit unit 11, so redundant description will be omitted. For convenience of illustration, the drawing omits the drive circuit 20 and the digital control device 100, but these will be described in detail later.

The capacitors C11 and C12 are connected in series between the node DC11 and the node D12, and function as a capacitor bridge in which the connection node between them serves as an AC neutral point (=applying end of the neutral point voltage VC). If the capacitance values of the capacitors C11 and C12 are equal, VC=(DC11-DC12)/2.

The drain of transistor M11 is connected to node DC11. The source of transistor M11 is connected to the drain of transistor M12. The source of transistor M12 is connected to node DC12. Gate signals G11 and G12 are input to the gates of the transistors M11 and M12, respectively. The transistors M11 and M12 are turned on when the gate signals G11 and G12 are at high level, and turned off when the gate signals G11 and G12 are at low level. Thus, transistors M11 and M12 are connected in series between nodes DC11 and DC12 and function as a first transistor bridge.

The drain of transistor M21 is connected to node DC11. The source of transistor M21 is connected to the drain of transistor M22. The source of transistor M22 is connected to node DC12. Gate signals G21 and G22 are input to the gates of the transistors M21 and M22, respectively. The transistors M21 and M22 are turned on when the gate signals G21 and G22 are at high level, and turned off when the gate signals G21 and G22 are at low level. Thus, transistors M21 and M22 are connected in series between nodes DC11 and DC12 and function as a second transistor bridge.

The first transistor bridge (=transistors M11 and M12) and the second transistor bridge (=transistors M21 and M22) have a predetermined phase difference .theta. π(180°)).

The bidirectional switch SW1 is connected between the midpoint node of the capacitor bridge (=connection node between the capacitors C11 and C12) and the output node of the first transistor bridge (=connection node between the transistors M11 and M12). It is A gate signal G13 is input to the gate of the bidirectional switch SW1. The bidirectional switch SW1 is turned on when the gate signal G13 is at high level, and turned off when the gate signal G13 is at low level.

The bidirectional switch SW2 is connected between the midpoint node of the capacitor bridge (=connection node between the capacitors C11 and C12) and the output node of the second transistor bridge (=connection node between the transistors M21 and M22). It is A gate signal G23 is input to the gate of the bidirectional switch SW2. The bidirectional switch SW2 is turned on when the gate signal G23 is at high level, and turned off when the gate signal G23 is at low level.

The transformer TR11 is connected between the output node of each of the first transistor bridge and the second transistor bridge and the first end of the reactor L11. A second end of reactor L11 is connected to node AC11.

If the exciting inductances Lm1 and Lm2 of the transformer TR11 are sufficiently large with respect to the reactor L11, the transformer TR11 and the reactor L11 can be configured as a transformer coupling reactor TCR11 (FIG. 9) having both functions. In this case, reactor L11 is formed by leakage inductances Ls1 and Ls2 of transformer coupled reactor TCR11. Also, the node voltages VN1 and VN2 appearing between the coupling portion of the transformer coupling reactor TCR11 and the leakage inductances Ls1 and Ls2 are substantially the same potential as the node voltage VN appearing at the midpoint node of the transformer TR11.

Capacitors C21 to C23 are respectively connected between nodes AC11 to AC13 and the midpoint node of the capacitor bridge (=connection node between capacitors C11 and C12).

In the power conversion device 1 of this embodiment, the switching output level of the transistor bridge is set to three values (+E, 0, -E) or more, instead of two values (+E and -E) of H/L. It can be changed in multiple stages so that it has a value. Therefore, since the voltage applied to the reactor L11 can be reduced, it is possible to reduce the size, loss, or noise of the reactor L11. In addition, since the voltage applied to each transistor can be reduced, it is possible to adopt low-voltage elements that are widely available in the market and to reduce switching loss.

In particular, power converters are currently being applied to a very wide range of fields such as in-vehicle equipment as well as consumer equipment and industrial equipment. Small size, light weight, and high efficiency are emphasized in the power conversion device used for these applications, and the demand is greater for devices with higher power. Therefore, the power conversion device 1 of the present embodiment is suitable. I can say

FIG. 10 is a diagram showing a configuration example of a digital control device 100 used in the power conversion device 1 of the second embodiment. The digital control device 100 of this configuration example is based on the first embodiment (FIG. 1), but the post-stage of the pulse width modulation section 105 is modified. More specifically, the digital control device 100 of this configuration example includes a phase shift section 108 and signal switching sections 109 and 110 instead of the signal switching section 107 described above.

Phase shift section 108 generates pulse width modulated signal PWM2 (and its It outputs an inverted pulse width modulated signal PWM2B) whose logic level is inverted.

The signal switching unit 109 switches output destinations of the pulse width modulated signal PWM and the inverted pulse width modulated signal PWMB according to the zero-crossing signal ZX.

For example, when ZX=H, S11=PWM, S12=L fixed, and S13=PWMB. By such signal switching, when the AC voltage is positive (ZX=H), the transistor M11 and the bidirectional switch SW1 are complementarily turned on/off, and the transistor M12 is always turned off.

On the other hand, when ZX=L, S11=L fixed, S12=PWMB, and S13=PWM. By such signal switching, when the AC voltage is negative (ZX=L), the transistor M12 and the bidirectional switch SW1 are complementarily turned on/off, and the transistor M11 is always turned off.

The signal switching unit 110 switches the output destinations of the pulse width modulated signal PWM2 and the inverted pulse width modulated signal PWM2B according to the zero-crossing signal ZX.

For example, when ZX=H, S21=PWM, S22=L fixed, and S23=PWMB. By such signal switching, when the AC voltage is positive (ZX=H), the transistor M21 and the bidirectional switch SW2 are complementarily turned on/off, and the transistor M22 is always turned off.

On the other hand, when ZX=L, S21=L fixed, S22=PWMB, and S23=PWM. By such signal switching, when the AC voltage is negative (ZX=L), the transistor M22 and the bidirectional switch SW2 are complementarily turned on/off, and the transistor M21 is always turned off.

<Application of Wide Bandgap Semiconductors (SiC, GaN)>
Various switch elements used in the power converter 1 (transistors M1 to M4 in FIG. 1, or transistors M11 to M12 and M21 to M22 in FIG. 8 (and FIG. 9), and bidirectional switches SW1 and SW2) , at least one of them is preferably formed of a wide bandgap semiconductor (SiC-based semiconductor, GaN-based semiconductor, or the like).

Thus, a switch element made of a SiC-based semiconductor (SiC-MOSFET, etc.) or a GaN-based semiconductor (GaN-HEMT [high electron mobility transistor], etc.) has a higher output than a switch element made of a Si-based semiconductor. Since parasitic capacitance such as capacitance and feedback capacitance can be reduced, an increase in switching loss during high-frequency driving can be suppressed.

Also, if SiC-MOSFETs are used as the switch elements, low on-resistance and high thermal conductivity due to the vertical structure can be obtained. Therefore, it is possible to realize the power conversion device 1 with large current and large power.

In addition, the SiC-MOSFET has a small reverse recovery current of the body diode and a small parasitic capacitance, so the effective value of the current can be kept low, and the conduction loss of the switch element and pattern, and the copper of the transformer coupling reactor. Loss can be reduced.

As described above, a switching element made of a wide bandgap semiconductor has a high breakdown voltage, a low on-resistance and a low switching loss, and these tendencies are relatively maintained even at high temperatures. Therefore, even when the input voltage and the voltage directly applied to the switch element are high, sufficiently thermally permissible operation is possible.

<Other Modifications>
In addition to the above embodiments, the various technical features disclosed in this specification can be modified in various ways without departing from the gist of the technical creation. For example, the mutual replacement of bipolar transistors with MOS field effect transistors and the logic level inversion of various signals are optional. That is, the above-described embodiments should be considered as examples and not restrictive in all respects, and the technical scope of the present invention is not limited to the above-described embodiments, and the claims should be understood to include all changes falling within the meaning and range of equivalence to the range of.

The power conversion device disclosed in this specification can be used in a very wide range of fields such as in-vehicle equipment as well as consumer equipment and industrial equipment.

1 power conversion device 10 switch circuit (bidirectional inverter)
11 to 13 switch circuit unit 20 drive circuit 21 to 24 gate driver 100 digital control device 101 reference signal setting unit 102 coefficient setting unit 103 multiplication unit 104 addition unit 105 pulse width modulation unit 106 zero cross detection unit 107 signal switching unit 108 phase shift unit 109, 110 Signal switching unit AC1, AC2, AC11 to AC13 node (AC node)
C1, C2, C11, C12, C21 to C23 Capacitors DC1, DC2, DC11, DC12 Node (DC node)
E1, E2 Power supply L1 Inductor L11 Reactor Lm1, Lm2 Exciting inductance Ls1, Ls2 Leakage inductance M1 to M4, M11, M12, M21, M22 Transistor SW1, SW2 Bidirectional switch TCR11 Transformer coupling reactor TR11 Transformer Z1, Z2 Load

Claims (12)

A control device used as a control subject of a power conversion device including a switch circuit including a transistor,
A multiplied signal obtained by multiplying a current to be controlled of the switch circuit by a predetermined coefficient is subtracted from a reference signal set according to an operation method of the power converter, and a control signal for the transistor is generated based on the result of the calculation. A control device characterized by: The switch circuit is a PFC [power factor correction] circuit,
the reference signal is a constant value;
wherein the controlled current is an input current;
The control device according to claim 1, characterized in that: wherein the switch circuit is an inverter;
wherein the reference signal is a sinusoidal signal;
wherein the controlled current is an output current;
The control device according to claim 1, characterized in that: the switch circuit is a bidirectional inverter,
changing the waveform of the reference signal when switching the energization direction so that the reference signal has a constant value during AC/DC conversion and the reference signal becomes a sinusoidal signal during DC/AC conversion;
As the current to be controlled, the current flowing through the same node of the switch circuit is monitored during both AC/DC conversion and DC/AC conversion. At the time of conversion, the controlled current becomes an output current,
The control device according to claim 1, characterized in that: 5. The control device according to claim 2, wherein a modulation signal for giving distortion to said input current is superimposed on said reference signal. 5. The control device according to claim 3, wherein a modulation signal for canceling distortion of load current flowing from a commercial power source to another load is superimposed on said reference signal. a frequency of the reference signal is set to a frequency required for the output current;
the coefficient is set such that the multiplied signal is negligibly small compared to the reference signal;
5. The control device according to claim 3 or 4, characterized in that: a capacitor bridge including at least two capacitors connected in series between the first DC node and the second DC node;
a first transistor bridge and a second transistor bridge each comprising two transistors connected in series between said first DC node and said second DC node;
a first bidirectional switch and a second bidirectional switch respectively connected between a midpoint node of the capacitor bridge and an output node of each of the first transistor bridge and the second transistor bridge;
a transformer connected to each output node of the first transistor bridge and the second transistor bridge;
a reactor connected between the transformer and an AC node;
a capacitor connected between the AC node and a midpoint node of the capacitor bridge;
with
A bi-directional inverter, wherein the first transistor bridge and the second transistor bridge are operated with a phase difference of 180 degrees from each other. The first transistor bridge and the second transistor bridge, the first bidirectional switch and the second bidirectional switch, the transformer, the reactor, and the capacitor have a three-phase configuration, A bidirectional inverter according to claim 8 . 10. The bidirectional inverter according to claim 8, wherein said transformer and said reactor are formed as a transformer coupled reactor having both functions. Both sides according to any one of claims 8 to 10, characterized in that the transistor, the first bidirectional switch and the second bidirectional switch are each made of a wide bandgap semiconductor. direction inverter. A bidirectional inverter according to any one of claims 8 to 11;
The control device according to any one of claims 1 to 7, which is a control body of the bidirectional inverter;
A power converter, characterized by comprising:

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