JP7121924B2 - High frequency transformer and power supply circuit using the same - Google Patents

Description

The present invention relates to a high frequency transformer and a power supply circuit using the same.

In recent years, there has been a strong demand for smaller, lighter, and more efficient power supplies, and efforts are being made to raise the switching frequency of power supply circuits such as converters to 100 kHz or higher. Although a transformer used in a power supply circuit can be miniaturized by increasing the frequency, the copper loss caused by the windings of the transformer and the iron loss caused by the magnetic core increase. A low-loss magnetic core material can be selected to reduce iron loss, while a reduction in copper loss requires a reduction in DC resistance and AC resistance.

The main causes of transformer copper loss at high frequencies are the skin effect caused by uneven current density in the wire cross section of the coil and the AC resistance caused by the proximity effect. When a high-frequency current is passed through the coil, the current distribution is biased toward the surface of the wire due to the temporal change in the magnetic field caused by the current flowing through the winding. For example, in a wire with a circular cross section, if the skin thickness defined as the region in which the current distribution exists is less than the radius, the current density is greatly different between the outer periphery and the center of the wire, increasing the AC resistance.

Also, the LLC resonance drive system is known as a power supply circuit capable of soft switching and reducing switching loss. The eddy currents generated by interlinking the windings tend to increase the AC resistance. Furthermore, when the other winding is present in the magnetic field created by one winding, the magnetic flux interlinking with the other winding causes an eddy current and increases the AC resistance.

In other words, in order to reduce the copper loss, it is necessary to reduce the DC resistance of the winding, the skin effect in the conductor, the proximity effect between the conductors, and the AC resistance based on the eddy current due to the leakage magnetic flux.

In order to reduce the AC resistance, a twisted wire (also called litz wire) is generally used for the winding of a transformer. Patent Document 1 describes that the influence of the skin effect is reduced by using a twisted wire. By reducing the cross-sectional area (wire diameter) of the strands and increasing the number of strands, it is possible to reduce losses due to the skin effect, proximity effect, and eddy currents, thereby suppressing the AC resistance of the windings.

Further, in Patent Document 2, the contact area between the primary winding and the secondary winding of the transformer is reduced to suppress the temperature rise of the transformer due to the proximity effect between the primary winding and the secondary winding. An example of the structure of the transformer is shown in FIG. 24 as a cross-sectional view. A transformer 500 is formed by arranging a primary winding 506 and a secondary winding 505 formed by winding a copper wire in an umbrella shape or a truncated conical shape, and placing them in a magnetic core 520 so that their smaller diameter sides face each other.

JP-A-62-219605 JP 2012-169331 A

It is possible to reduce the AC resistance by using twisted wires for the windings of transformers, or by making the windings conical, etc. There is a problem of suppressing a decrease in efficiency of the power supply circuit. SUMMARY OF THE INVENTION Accordingly, an object of the present invention is to provide a high-frequency transformer capable of reducing AC resistance and a power supply circuit using the same.

A first invention comprises a closed magnetic circuit magnetic core, and a primary winding and a secondary winding wound around the closed magnetic circuit magnetic core, and the secondary winding is arranged along the winding axis direction of the primary winding. and the wire space factor of the primary winding and the wire space factor of the secondary winding satisfy the following condition (1) or (2).
(1) A single primary winding and a single secondary winding are arranged side by side with a gap, and are composed of a twisted wire with a wire diameter of φ0.15 mm or more and φ0.35 mm or less, and the primary winding The wire space factor and the wire space factor of the secondary winding are 4.0% or more and 13.5% or less. The primary winding is composed of a stranded wire with a wire diameter of φ0.05 mm or more and φ0.35 mm or less, and the wire space factor of the primary winding and the wire space of the secondary winding rate is 10.0% or more and 50.0% or less

Preferably, said single primary winding and said single secondary winding are configured in a conical or pyramidal spiral shape.

In addition, the two-part primary winding and the single secondary winding are configured in a conical or pyramidal shape, and the single secondary winding has a large diameter intermediate portion and both ends of the secondary winding. It is preferred that the portion has a small diameter.

Also, the closed magnetic circuit magnetic core includes a first magnetic core and a second magnetic core, and is preferably configured so as to have a magnetic gap between the first magnetic core and the second magnetic core.

Further, the high-frequency transformer of the present invention is preferably for LLC resonant power supplies with a switching frequency of 100 kHz to 500 kHz.

The present invention also provides a power supply circuit using the high-frequency transformer described above.

According to the present invention, it is possible to provide a high-frequency transformer capable of reducing AC resistance and a highly efficient power supply circuit using the same.

1 is a plan view of a high frequency transformer according to one embodiment of the present invention; FIG. FIG. 4 is a plan view of a high frequency transformer according to another embodiment of the present invention; FIG. 4 is an enlarged view of a cross section that appears by radially cutting the stranded wire from the set. FIG. 10 is an enlarged view of a cross section that appears by radially cutting a strand of composite strands in which the set strands are further bundled; FIG. 4 is a partial cross-sectional view of a simulation model showing the winding structure of the high-frequency transformer analyzed. FIG. 11 is a partial cross-sectional view of a simulation model showing another winding structure of the high-frequency transformer analyzed. FIG. 11 is a partial cross-sectional view of a simulation model showing another winding structure of the high-frequency transformer analyzed. FIG. 11 is a partial cross-sectional view of a simulation model showing another winding structure of the high-frequency transformer analyzed. FIG. 11 is a partial cross-sectional view of a simulation model showing another winding structure of the high-frequency transformer analyzed. FIG. 11 is a partial cross-sectional view of a simulation model showing another winding structure of the high-frequency transformer analyzed. (a) It is a front view which shows the model structure of the magnetic core used for simulation, (b) It is the side view. (a) It is a front view which shows the model structure of the bobbin used for simulation, (b) It is the flat side view. (a) is a front view showing another model structure of the bobbin used in the simulation, and (b) is a side view thereof. FIG. 4 is a diagram showing the relationship between the wire lamination factor of a primary winding having a wire diameter of φ0.2 mm and the AC resistance Rac1; FIG. 4 is a diagram showing the relationship between the wire lamination factor of a primary winding having a wire diameter of φ0.3 mm and the AC resistance Rac1; FIG. 4 is a diagram showing the relationship between the wire lamination factor of a primary winding having a wire diameter of φ0.1 mm and the AC resistance Rac1; FIG. 4 is a diagram showing the relationship between the wire lamination factor of a primary winding having a wire diameter of φ0.2 mm and the AC resistance Rac1; FIG. 5 is a diagram showing the relationship between the wire lamination factor of a secondary winding having a wire diameter of φ0.1 mm and AC resistance Rac2. FIG. 5 is a diagram showing the relationship between the wire lamination factor of a secondary winding having a wire diameter of φ0.2 mm and AC resistance Rac2; (a) It is a front view of the magnetic core used for the high frequency transformer of this invention, (b) It is the side view. 1 is a perspective view of a high frequency transformer of the present invention; FIG. FIG. 3 is a block diagram of an evaluation circuit for conversion efficiency of a high frequency transformer; It is a figure which shows the relationship between the frequency of the high frequency transformer of an Example and a comparative example, and conversion efficiency. 1 is a plan view of a conventional high frequency transformer; FIG.

A high-frequency transformer according to an embodiment of the present invention and a power supply circuit using the same will be specifically described below. However, the present invention is not limited to this. In addition, in some or all of the drawings, parts unnecessary for explanation are omitted, and some parts are enlarged or reduced in order to facilitate explanation. In addition, unless otherwise specified, the dimensions, shapes, relative positional relationships, and the like of constituent members shown in the description are not limited to these. Furthermore, in the description, the same names and symbols indicate the same or homogeneous members, and detailed description may be omitted even if they are illustrated.

FIG. 1 is a cross-sectional view showing a high frequency transformer according to one embodiment of the present invention. As shown in FIG. 1, the high-frequency transformer 1 includes a pair of magnetic cores 10a and 10b, a pair of windings 20a and 20b, and a winding frame (bobbin) 15 for holding the windings 20a and 20b. A gap G is formed in the magnetic path. In the following description, the winding 20a may be called the primary winding and the winding 20b may be called the secondary winding, but the distinction between the input side and the output side is not limited unless otherwise specified.

FIG. 2 is a cross-sectional view showing another configuration of the high frequency transformer according to one embodiment of the present invention. The difference from the high frequency transformer shown in FIG. 1 is that the primary winding 20a is divided into two windings 20a1 and 20a2. One primary winding 20a1 divided into two, a single secondary winding 20b, and the other primary winding 20a2 divided into two are arranged in order, and the divided primary windings 20a1 and 20a2 are connected in series. ing. In the illustrated example, the single secondary winding 20b is sandwiched between the divided primary windings 20a1 and 20a2, but the secondary winding 20b may be divided and the primary winding 20a may be sandwiched between them. . Note that, in relation to the divided winding, the undivided winding may be distinguished as a "single winding".

Each of the primary winding 20a and the secondary winding 20b is composed of a stranded wire formed by twisting a plurality of fine copper wires (strands) or a composite stranded wire formed by further bundling a plurality of thin copper wires (strands). A stranded wire uses strands with a smaller diameter than a single wire rod, so as described in Cited Document 1, the influence of the skin effect can be significantly reduced and the loss can be reduced.

FIG. 3 is an enlarged cross-sectional view of a bundle of twisted copper wires (strands) cut radially. The cross-sectional shape of the stranded wire 100 is equivalently circular as indicated by the dotted line. The wires 110 are copper wires or aluminum wires, each of which is provided with an insulating coating 120 . The insulating coating 120 is, for example, vinyl resin, fluororesin, formal resin, or the like. FIG. 4 is an enlarged view of a cross-section that appears by radially cutting a strand of composite strands in which the set strands are further bundled. The cross-sectional shape of the composite twist strand 130 is also equivalently circular, as indicated by the dashed line.

In the example of FIG. 3, seven strands 110 of the same wire diameter are formed into a set twisted wire 100, and in the example of FIG. 130 is configured, but the number of strands, wire diameter, etc. may be appropriately changed within the scope of the present invention, and is not limited to the illustrated example.

In the present invention, the primary winding 20a and the secondary winding 20b each have an upper limit of wire diameter of φ0.35 mm. This is because a wire diameter of φ0.35 mm is nearly twice the skin thickness of copper at 100 kHz, and if the wire diameter exceeds φ0.35 mm, AC resistance increases. When the primary winding 20a is divided into windings as shown in FIG. 2, the wire diameter is φ0.05 mm or more, and as shown in FIG. In the case of one winding, the diameter should be φ0.15 mm or more. If the wire diameter is less than a predetermined value, it may be difficult to obtain the effect of reducing AC resistance.

In order to obtain the effect of reducing the AC resistance of the primary winding 20a and the secondary winding 20b, when the primary winding 20a is divided into windings, the wire space factor should be 10.0% or more and 50.0% or less. do. Further, the wire space factor is preferably 15.0% or more, and preferably 40.0% or less. When both the primary winding 20a and the secondary winding 20b are not divided windings, the wire space factor is set to 4.0% or more and 13.5% or less. Further, the wire space factor is preferably 4.5% or more, and preferably 12.0% or less.

A bobbin 15 used in the high-frequency transformer 1 partitions between the primary winding 20a, the secondary winding 20b and the magnetic cores 10a and 10b, and the partitioning portion 15a between the connecting portions 13a and 13b of the magnetic cores 10a and 10b and the magnetic core 10a. , 10b, and a partition portion 15c between the primary winding 20a and the secondary winding 20b. The partition portion 15b has a tubular shape surrounding the middle legs 11a and 11b of the magnetic cores 10a and 10b, and the partition portions 15a and 15c are erected from the periphery of the partition portion 15b. A primary winding 20a and a secondary winding 20b are arranged side by side in a region defined by the partition portion 15a and the partition portion 15c. The bobbin 15 is a former for winding the primary winding 20a and the secondary winding 20b, and ensures insulation between the magnetic cores 10a and 10b and the primary winding 20a and the secondary winding 20b. The pair of magnetic cores 10a and 10b are fixed with an adhesive by passing the middle legs 11a and 11b through the tubular portion formed by the partition 15b of the bobbin 15, and restrained and fixed with an insulating tape, a fastener, or a band. .

The bobbin 15 is made of a resin having insulating properties, heat resistance, and moldability. Specifically, polyphenylene sulfide, liquid crystal polymer, polyethylene terephthalate, polybutylene terephthalate, and the like are preferable. They can be molded by a method such as an injection molding method or a resin molding method. In the illustrated example, the bobbin 15 is shared between the primary winding 20a and the secondary winding 20b, but may be a bobbin corresponding to each of the primary winding 20a and the secondary winding 20b. Moreover, it is preferable to form the partition portion 15b in accordance with the shape of the primary winding 20a and the secondary winding 20b, which will be described later. The shape may be adjusted to the inclination of the side surface.

Each of the magnetic cores 10a, 10b is preferably E-shaped or U-shaped. In the case of the E shape, the magnetic core 10a consists of a middle leg portion 11a, two side leg portions 12a, 12a, and a connecting portion 13a connecting them. Similarly, the magnetic core 10b is composed of a middle leg portion 11b, two side leg portions 12b, 12b, and a connecting portion 13b connecting them. The connecting portions 13a and 13b are formed in a substantially rectangular plate shape, the side legs 12a and 12b stand parallel to each other from the ends of the connecting portions 13a and 13b, and the middle legs 11a and 11b are divided into two It is located between the side legs 12a and 12b and rises from the central portions of the connecting portions 13a and 13b. The length of the middle legs 11a, 11b is shorter than that of the side legs 12a, 12b. The high-frequency transformer 1 is constructed by facing the middle legs 11a and 11b of the magnetic cores 10a and 10b and the two side legs 12a and 12b with the ends facing each other. A closed magnetic circuit is formed in the high-frequency transformer 1 through the middle legs 11a and 11b, the side legs 12a and 12b, and the connecting portions 13a and 13b, and a gap G is formed between the facing middle legs 11a and 11b. .

Each of the magnetic cores 10a and 10b is made of Ni-based, Mn-based soft ferrite, pure iron, Fe--Si-based magnetic alloy, Fe--Si--Al-based magnetic alloy, Fe--Si--Cr-based magnetic alloy, or Fe-based ferrite. , Co-based amorphous alloys, nanocrystalline alloys (typically Finemet (registered trademark)), and other metallic magnetic materials can be used. For example, when the load current at which magnetic saturation occurs is relatively small, it is preferable to use a low-loss soft ferrite with a small saturation magnetic flux density. It is preferable to use a metallic magnetic material with a high saturation magnetic flux density. When the metallic magnetic material is provided in the form of a ribbon, the ribbon may be punched into a predetermined shape to form a laminated magnetic core, or when provided in the form of powder, it is mixed with a resin or glass that will serve as an insulating coating, molded, and heat-treated. It is good also as a powder magnetic core comprised.

With respect to such a high-frequency transformer, the effect of suppressing AC resistance by simulation using finite element method (FEM) analysis will be described. The main analysis conditions are shown below.
・Software used for simulation JMAG-Designer 17.0.02v x64 edition was used.
Analysis method A two-dimensional frequency response magnetic field analysis was performed using a TS (transformer analysis) module and an FQ (frequency response magnetic field analysis) module.
The number of divisions of the winding varies depending on the winding conditions, and the conditions are such that the AC resistance can be calculated with high accuracy. In the analysis described later, for example, the conditions of a wire diameter of φ0.1 at 450 kHz (primary side 5-turn wire number 59 × 4 layers dense winding, secondary side 2-turn wire number 233 × 2 layers dispersion Vol.) has 44097 elements. There are 215,700 elements under the φ0.1 dense winding condition at 200 kHz (225 12-turn strands on the primary side and 225 18-turn strands on the secondary side).
・Material parameters The wire was made of copper, and the resistivity ρ was set to 1.673×10 ⁻⁸ Ωm.
Two types of Mn-based ferrite ML91S material and ML29D material manufactured by Hitachi Metals, Ltd. were used as magnetic core materials.
・Magnetic core shape PQ shape

5 to 10 are partial cross-sectional views of a simulation model showing the winding structure of the high-frequency transformer analyzed. 5 to 7 have the same configuration as the high frequency transformer shown in FIG. 1, and FIGS. 8 to 10 have the same configuration as the high frequency transformer shown in FIG. is different.

FIG. 5 is an example of a simulation model of a high frequency transformer. The primary winding 20a is stacked in four layers in the extending direction of the connecting portion 13a, and the twisted wire 100 is wound five times in each layer in the extending direction of the middle leg portion 11a. The secondary winding 20b has two layers, and the twisted wire 100 is wound twice on each layer. The primary winding 20a has 5 turns by connecting 4 layers in parallel, and the secondary winding 20b has 2 turns by connecting 2 layers in parallel. A structure in which twisted wires such as the primary winding 20a are aligned in multiple layers and wound densely will be referred to as dense winding for ease of explanation. Further, the secondary winding 20b has a wider wire interval than the windings of the densely wound structure, and is formed farther apart. The inner and outer peripheral surfaces of the primary winding 20a and the secondary winding 20b have a shape following the outer peripheral surface of the partition portion 15b of the bobbin 15 between the cores 10a and 10b and the middle legs 11a and 11b. Become. In the illustrated example, both the inner and outer peripheral surfaces of the primary winding 20a and the secondary winding 20b are circular. , the inner peripheral surface and the outer peripheral surface of the secondary winding 20b are substantially rectangular.

FIG. 6 is a simulation model of a high frequency transformer according to another embodiment of the present invention. The primary winding 20a and the secondary winding 20b have different winding diameters at both ends in the winding axis direction. In the illustrated example, the primary winding 20a provided on the magnetic core 10a side The winding diameter gradually increases from one side to the other. The winding diameter of the secondary winding 20b provided on the magnetic core 10b side is gradually reduced from one side to the other side toward the connecting portion 13b side. The primary winding 20a has a single layer, and the stranded wire 100 is wound five times to form a five-turn winding. The secondary winding 20b is formed by winding two turns of the two-layered stranded wire 100, forming a two-turn winding in which the two-layered stranded wire is connected in parallel. The peripheral surfaces of both the primary winding 20a and the secondary winding 20b are inclined with respect to the direction of the winding axis, and in the illustrated example, the shape of the peripheral surface is conical. Note that the shape of the peripheral surface may be a pyramid. Such a stranded wire winding structure is called a conical winding. The secondary winding 20b is a distributed winding at two locations, and the circumferential surface is an imaginary surface connecting the windings at the two locations.

FIG. 7 is a simulation model of a high frequency transformer according to another embodiment of the present invention. Each of the primary winding 20a and the secondary winding 20b is a distributed winding. In the illustrated example, the primary winding 20a is formed by winding the stranded wire 100 three times on the side of the connecting portion 13a of the magnetic core 10a, and then winding it two times at a position spaced apart in the direction of the winding axis, for a total of five turns. there is The secondary winding 20b has a similar configuration, and both have 5 turns.

FIG. 11 shows the model structure of the magnetic core used for the simulation. The magnetic core 10 has a PQ shape, and two of them are used in combination. It consists of a middle leg 11, two side legs 12, and a connecting portion 13 connecting them. The back surface that does not appear in the figure is a flat surface. Table 1 shows the dimensions of each part of the magnetic core 10 . ML91S material of Mn-based ferrite manufactured by Hitachi Metals, Ltd. was used as the magnetic core material, and the initial magnetization curve data of the DC BH curve was used for the simulation.

FIG. 12 shows the model structure of the bobbin used in the simulation. The bobbin 15 has partitions 15a, 15b, 15c and 15d between the primary winding 20a, the secondary winding 20b and the magnetic cores 10a and 10b. Table 2 shows the dimensions of each part of the bobbin 15 used in the simulation.

Table 3 shows the conditions under which the simulation was performed by changing the twisted wire and winding structure of the primary winding 20a and the secondary winding 20b, and Table 4 shows the results. The winding specifications are such that the self-inductance on the primary winding side (secondary winding side open) is 15.0 μH to 16.0 μH, and the leakage inductance (secondary winding side short circuit) is 3.0 μH to 3.3 μH. was set to be The distance LL between the primary winding and the secondary winding was set to 10 mm, and the gap length was set to 0.3 mm. The drive conditions in the simulation are such that the primary winding side is excited by a sine wave with a frequency of 450 kHz, and the secondary winding side output is a voltage of 48 V and a current of 20 A.

The wire diameter ds1 in Table 3 is the diameter of the copper wire (wire) 110 shown in FIG. The strand total cross-sectional area Sl of the stranded wire is the total cross-sectional area of each strand and was calculated by the following equation.
Sl = N x pi x 1/4 x ds1 ²
N Number of strands ds1 Nominal value of outside diameter of strands (not including insulation coating)

The wire lamination factor is a rectangle defined by the maximum width a1 in the x direction and the maximum height b1 in the y direction of the primary winding 20a in the cross section in the winding axis direction x and the direction y orthogonal thereto. Percentage of the ratio Sl/S of the total cross-sectional area Sl of the wire to each of the area S of the area and the area S of the rectangular area defined by the maximum width c1 in the x direction and the maximum height d1 in the y direction of the secondary winding 20b calculated as

FIG. 14 shows the relationship between the wire lamination factor of the primary winding and the AC resistance when the wire diameter is 0.2 mm (Nos. 4 to 15). FIG. 15 shows the relationship between the wire lamination factor of the primary winding and the AC resistance when the wire diameter is 0.3 mm (Nos. 17 to 30). The dashed line in the figure indicates No. 1, which has a large space factor. 3, No. 16 shows the level of AC resistance at 16. In the winding structure of distributed winding or conical winding, there is a range of wire space factor in which the AC resistance Rac1 can be reduced more than in the case of the winding structure with the same wire diameter and a relatively large wire space factor. I understand. The larger the wire diameter, the greater the reduction ratio of the AC resistance Rac1, and the conical winding reduced the AC resistance Rac1 more slightly than the distributed winding. Distributed winding reduced AC resistance by up to about 21%, and conical winding reduced by up to about 26%. By arranging the conically wound primary winding 20a and secondary winding 20b so that the large diameter sides of the windings face each other, the effect of leakage magnetic flux in the gap G is reduced and the proximity effect is suppressed. It seems to be the effect of

On the other hand, from Tables 3 and 4, the AC resistance Rac2 of the secondary winding 20b is also reduced. The larger the wire diameter, the greater the reduction ratio of the AC resistance Rac2, and the AC resistance was reduced by about 41% at maximum.

Furthermore, a simulation was performed by changing the structure of the windings to the divided windings of the high-frequency transformer shown in FIG. FIG. 8 is a simulation model of a high frequency transformer according to another embodiment of the present invention. This model has a split winding configuration in which the primary winding 20a is divided into two. The windings 20a1 and 20a2 obtained by dividing the primary winding 20a and the secondary winding 20b are both conical windings. The winding diameter of the secondary winding 20b gradually changes from one end to the other, and is configured such that the winding diameter is small at both ends and large at the center.

FIG. 9 is a simulation model of a high frequency transformer according to one embodiment of the present invention. This model is also a high-frequency transformer having the structure shown in FIG. It differs in that it is arranged so as to face the secondary winding 20b.

FIG. 10 is a simulation model of a high frequency transformer according to one embodiment of the present invention. This model is also a high-frequency transformer having the structure shown in FIG. It is different in that it is wound twice in the direction of the winding axis to form a conical winding.

Although not shown, the AC resistance of a high-frequency transformer according to another embodiment of the present invention, in which the primary winding 20a and the secondary winding 20b are densely wound or dispersedly wound, was analyzed by simulation.

The structure of the magnetic core in the simulation model is the same as in FIG. 11, and Table 5 shows the dimensions of each part of the magnetic core 10. A Mn-based ferrite ML24D material manufactured by Hitachi Metals, Ltd. was used as the magnetic core material, and the initial magnetization curve data of the DC BH curve was used for the simulation.

FIG. 13 shows the model structure of the bobbin used for the simulation. The bobbin 15 has partitions 15a, 15b, 15c and 15d between the primary winding 20a, the secondary winding 20b and the magnetic cores 10a and 10b. There are two places where the primary winding 20a is provided so as to sandwich the secondary winding 20b. Table 6 shows the dimensions of each part of the bobbin 15 used in the simulation.

Tables 7 and 8 show the results of simulations performed by changing the twisted wire conditions of the primary winding 20a and the secondary winding 20b and the winding structure. In the table, the winding mode in FIG. 8 is called conical winding A, the winding mode in FIG. 9 is called conical winding B, and the winding mode in FIG. 10 is called conical winding C. The primary winding 20a is divided into two parts, a primary winding 20a1 and a primary winding 20a2. Table 7 shows the twisted wire conditions of the primary winding 20a1. Description of the conditions of the primary winding 20a2 configured under the same conditions as the primary winding 20a1 is omitted in the table. The winding specifications were set so that the self-inductance of the primary winding (secondary winding side open) was 167 μH to 177 μH, and the leakage inductance (secondary winding side short) was 19 μH to 21 μH. The distances LL1 and LL2 between the primary and secondary windings were both set to 5.2 mm, and the gap length was set to 1.0 mm. Driving conditions in the simulation are such that the primary winding side is excited with a sine wave of frequency 200 kHz, and the secondary winding side output is a voltage of 360 V and a current of 10 A.

FIG. 16 shows the relationship between the wire lamination factor of the primary winding and the AC resistance when the wire diameter is 0.1 mm (Nos. 32 to 39). FIG. 17 shows the relationship between the wire lamination factor of the primary winding and the AC resistance when the wire diameter is 0.2 mm (Nos. 41 to 48). The dashed line in the figure indicates No. 1, which has a large space factor. 31, No. 40 shows the level of AC resistance. It can be seen that even when the primary winding 20a is distributed winding, there is a range of wire lamination factor in which the AC resistance Rac1 can be reduced. Further, the AC resistance Rac1 can be reduced in a range wider than that of the high frequency transformer shown in FIG. 1 in which the primary winding 20a is not divided windings, and includes a wire space factor of 10.0% or more and 50.0% or less.

FIG. 18 shows the relationship between the wire lamination factor of the secondary winding and the AC resistance when the wire diameter is 0.1 mm (Nos. 32 to 39). FIG. 19 shows the relationship between the wire lamination factor of the secondary winding and the AC resistance when the wire diameter is 0.2 mm (Nos. 41 to 48). The dashed line in the figure indicates No. The level of AC resistance at 31, 40 is shown. Also in this case, the AC resistance Rac2 can be reduced within a range including the wire space factor of 10.0% or more and 50.0% or less.

Using the magnetic core shown in FIG. 20, a high frequency transformer shown in FIG. 21 was manufactured. This high-frequency transformer has a structure similar to that shown in FIG. The primary winding 20a and the secondary winding 20b are juxtaposed with a predetermined spacing via a spacer 16 made of insulating resin. As an example, the magnetic core material is ML24D material manufactured by Hitachi Metals, the magnetic core shape is EER53, the gap is a spacer gap of 0.6 mm (total gap length is 1.2 mm) provided in the middle leg and the side leg, and the primary winding A wire 20a and a secondary winding 20b consisted of 40 twisted wires with a wire diameter of φ0.23 mm, each of which had five turns, and a high-frequency transformer composed of conical windings was obtained. As a comparative example, with the same core and gap as the example, 150 twisted wires with a wire diameter of φ0.23 mm are used for the primary winding 20a and the secondary winding 20b, and the number of turns is 5, and 3 layers of windings are arranged in parallel. A connected close-wound high-frequency transformer was also constructed. Table 9 shows the dimensions of each part of the magnetic core. The bobbin has a structure similar to that shown in FIG. 12, and Table 10 shows the dimensions of each part. In addition, Table 11 summarizes the wire lamination factor and winding interval LL of each wire. Insulating tape is wound around the partition portion 15b of the bobbin to form an inclination so that the primary winding 20a and the secondary winding 20b are conically wound.

Using the obtained high-frequency transformer, the power supply circuit of FIG. 22 was used to evaluate the conversion efficiency. The conversion efficiency of the transformer was calculated from the power values on the primary winding side and the secondary winding side obtained by a power analyzer WT3000 manufactured by Yokogawa Electric Corporation. The results are shown in FIG. Compared to the dense winding of the comparative example, the conical winding of the present invention provided a high efficiency of 98% or more in a wide frequency range, especially on the high frequency side.

It has been found that the high-frequency transformer of the present invention can reduce the influence of the proximity effect and reduce the AC resistance. Moreover, the conversion efficiency of the high-frequency transformer is high in a wide range and is suitable for power supply circuits.

1 high-frequency transformers 10a, 10b magnetic core 15 bobbin 20a primary winding 20b secondary winding 100 twisted wire 110 element wire (copper wire)
120 film

Claims (4)

a closed magnetic circuit magnetic core, and a primary winding and a secondary winding wound around the closed magnetic circuit magnetic core, wherein the secondary winding is arranged along the winding axis direction of the primary winding,
A high-frequency transformer in which the primary winding and the secondary winding satisfy the following condition (2) .
(2) A primary winding that is divided into two and arranged at a position sandwiching a single secondary winding, and is composed of a twisted wire with a wire diameter of φ0.05 mm or more and φ0.35 mm or less, and the primary winding The wire space factor of the winding and the wire space factor of the secondary winding are 10.0% or more and 50.0% or less, and the primary winding divided into two and the single secondary winding The windings are arranged in a conical or pyramidal spiral shape, and the single secondary winding has a large diameter in the middle part and a small diameter at both ends. A high frequency transformer according to claim 1 ,
A high-frequency transformer in which the closed magnetic circuit magnetic core includes a first magnetic core and a second magnetic core, which are combined so as to have a magnetic gap between the first magnetic core and the second magnetic core. The high frequency transformer according to claim 1 or 2 ,
A high-frequency transformer for LLC resonant power supplies with a switching frequency of 100 kHz to 500 kHz. A power supply circuit using the high-frequency transformer according to any one of claims 1 to 3 .

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