JP7090826B1 - Radar device - Google Patents

Abstract

The radar device (100) according to the present disclosure technology continuously generates a plurality of transmission pulse signals at a timing based on a pulse repetition cycle and a PRI control unit (150) that sets a predetermined pulse repetition cycle for each hit. A signal generation circuit (110), a transmission / reception unit (120) that sends a transmission pulse signal to an external space and receives a plurality of reflected wave signals corresponding to each of the transmission pulse signals from the external space, and a transmission / reception unit (120). A receiving circuit (140) that generates a plurality of received signals corresponding to each of the transmitted pulse signals by sampling each of the reflected wave signals received in the above, and a correlation processing unit (161) that pulse-compresses the received signals by a correlation calculation. ), A compensation processing unit (162) that performs motion compensation by a preset speed candidate, and a target candidate detection unit (163) that detects a target candidate based on the processed signal of the compensation processing unit (162). A speed measurement that estimates the unambiguous speed and distance of the target from the results from the target candidate detection unit processed using multiple motion compensation candidates at intervals wider than the unambiguous measurable speed interval. It is provided with a distance measuring unit (164).

Description

The disclosed technique relates to a radar device and a signal processing method for the radar device.

There are various types of radar devices. For example, a radar device includes a pulse Doppler system that transmits a plurality of pulse waves toward a target based on a pulse repetition period (PRI) and detects the distance and velocity of the target.

The technique relating to the pulse Doppler type radar device can be seen in, for example, Patent Document 1. Patent Document 1 discloses a technique relating to a radar device that performs high-precision Doppler correction even when ambiguity occurs to improve super-resolution ranging accuracy.

Japanese Unexamined Patent Publication No. 2010-281605

In the speed measurement by the radar device, the speed interval that can be solved without ambiguity is narrow when the PRI becomes long assuming a target moving at a long distance and at a high speed. Therefore, under such a situation, it is necessary to search with a wide speed interval of the assumed target. Further, in order to obtain the maximum strength of the target signal with high accuracy, it is necessary to make the step of the speed compensation candidate finer, which increases the calculation load. In the method of solving the velocity ambiguity by the search described in Patent Document 1, there is a trade-off relationship between the accuracy of solving without ambiguity and the amount of calculation required for the search.

The disclosed technique solves the above-mentioned problems, and an object thereof is to obtain a speed measuring means capable of accurately and reducing the amount of calculation even if the speed interval is wide.

The radar device according to the present disclosure technology includes a PRI control unit that sets a predetermined pulse repetition cycle for each hit, a signal generation circuit that continuously generates a plurality of transmission pulse signals at a timing based on the pulse repetition cycle, and a signal generation circuit. By transmitting the transmission pulse signal to the external space and sampling each of the transmission / reception unit that receives a plurality of reflected wave signals corresponding to each of the transmission pulse signals from the external space and the reflected wave signal received by the transmission / reception unit. A receiving circuit that generates a plurality of received signals corresponding to each transmitted pulse signal, a correlation processing unit that pulse-compresses the received signal by a correlation calculation, a compensation processing unit that performs motion compensation by a preset speed candidate, and compensation. Using a target candidate detection unit that detects target candidates based on the processed signal of the processing unit, and multiple motion compensation candidates at intervals that are two or more integral multiples of the speed interval (Δν _amb ) without ambiguity. It is provided with a speed measuring / measuring unit that estimates the speed and distance of the target from the results from the target candidate detection unit processed in the above process.

Since the radar device according to the present disclosed technique has the above configuration, the target speed is estimated based on a plurality of speed candidates in a wide range in which speed ambiguity appears, while accurately and reducing the calculation load. be able to.

FIG. 1 is a block diagram showing a functional configuration of a radar device according to the present disclosure technique. FIG. 2 is a block diagram showing a functional configuration of the radar signal processing circuit 160 of the radar device according to the first embodiment. FIG. 3 is a block diagram showing a functional configuration of the signal generation circuit 110 of the radar device according to the first embodiment. FIG. 4 is a flowchart showing a processing step of the radar signal processing circuit 160 of the radar device according to the first embodiment. FIG. 5 is a graph showing an image of the velocity interval without ambiguity and the detected intensity distribution in the velocity compensation candidate. FIG. 6 is a graph showing an image of an assumed speed interval of a target, a speed interval without ambiguity, and a candidate of a target. FIG. 7 is a graph showing an image in which the speed and distance of the peak intensity of each speed compensation candidate are superimposed. FIG. 8 is a graph showing an image of calculating an estimated value of a target speed from the peak intensity of each speed compensation candidate. FIG. 9 is a graph showing an image of calculating the estimated distance from the estimated speed of the target. FIG. 10 is an image diagram showing the relationship between the ratio of the sum and difference signals with respect to the signal strength and the speed. FIG. 11 is a hardware configuration diagram of the signal processing circuit 170 of the radar device according to the present disclosure technique. FIG. 12 is a block diagram showing a functional configuration of the radar signal processing circuit 160 (160B) of the radar device according to the second embodiment. FIG. 13 is a flowchart showing a processing step of the radar signal processing circuit 160 (160B) of the radar device according to the second embodiment. FIG. 14 is a flowchart showing the details of the CZT pulse Doppler processing step (ST165) in the flowchart shown in FIG. FIG. 15 is a block diagram showing a functional configuration of the radar signal processing circuit 160 (160C) of the radar device according to the third embodiment. FIG. 16 is a flowchart showing a processing step of the radar signal processing circuit 160 (160C) of the radar device according to the third embodiment. FIG. 17 is an image diagram shown in FIG. 8 applied to the third embodiment. FIG. 18 is an image diagram shown in FIG. 9 applied to the third embodiment. FIG. 19 is a block diagram showing a functional configuration of the radar signal processing circuit 160 (160D) of the radar device according to the fourth embodiment. FIG. 20 is a flowchart showing a processing step of the radar signal processing circuit 160 (160D) of the radar device according to the fourth embodiment. FIG. 21 is a reference diagram showing an image of integration performed by the inter-pulse stagger integration unit 167 according to the fourth embodiment.

The radar device 100 according to the present disclosed technique is a pulse Doppler system that transmits a plurality of pulse waves toward a target based on a pulse repetition period (Pulse Repetition Interval, PRI) and detects a target distance and speed. ..
The radar device 100 according to the present disclosed technique will be clarified by the description along the drawings for each embodiment. In addition, in the mathematical formula included in the explanation, a proviso (provided) is shown after for. In almost all proviso, variables are integer serial numbers and their range is indicated. Variables are shown as m, h, q, k, etc., but their usage and range are one and the same for one variable, unless otherwise specified. In addition, depending on the formula, the proviso relating to the explanation of variables may be omitted.

Embodiment 1.
FIG. 1 is a block diagram showing a functional configuration of the radar device 100 according to the disclosed technique. As shown in FIG. 1, the radar device 100 includes a signal generation circuit 110, a transmission / reception unit 120, an antenna 130, a reception circuit 140, a PRI control unit 150, and a radar signal processing circuit 160. Further, the radar device 100 may be connected to the display 200.

FIG. 2 is a block diagram showing a functional configuration of the radar signal processing circuit 160 of the radar device 100 according to the first embodiment. As shown in FIG. 2, the radar signal processing circuit 160 includes a correlation processing unit 161, a compensation processing unit 162, a target candidate detection unit 163, and a speed measurement / distance measurement unit 164.

The radar device 100 may have a millimeter wave band or a microwave band as a working frequency band.

The signal generation circuit 110 of the radar device 100 generates a plurality of transmission pulse signals. The mathematical model of the transmitted pulse signal is represented by T _x (h, t). Here, the argument h represents a pulse hit number (hereinafter, simply referred to as a “hit number”). h is an integer from 0 to H-1. H represents the number of pulse hits. The hit number h is derived from the acronym "hit" in English. That is, the radar device 100 transmits H transmission pulse signals from 0 to H-1. The argument t represents time.

The transmission pulse signal is generated at the timing of each pulse repetition cycle (Pulse Repetition Intervals, hereinafter referred to as "PRI"). The design value of PRI is expressed in T _pri (h) and has a unit of time. Thus, the PRI may have different time lengths in pulses with different hit numbers. The PRI is set by the PRI control unit 150. The transmission pulse signal generated by the signal generation circuit 110 is output to the transmission / reception unit 120.

The transmission pulse signal output to the transmission / reception unit 120 is further output to the antenna 130. The antenna 130 may have a linear shape called an antenna. The transmitted pulse signal radiated to the external space is reflected by the target and becomes a reflected wave signal. The mathematical model of the reflected wave signal is represented by R _x (h, t).

The reflected wave signal is received by the antenna 130 and output to the transmission / reception unit 120. The reflected wave signal is output to the receiving circuit 140 by the transmission / reception unit 120. That is, the transmission / reception unit 120 outputs the transmission pulse signal from the signal generation circuit 110 to the antenna 130, and outputs the reflected wave signal from the antenna 130 to the reception circuit 140. The transmission / reception unit 120 may be realized by a circulator having three ports.

The receiving circuit 140 performs analog signal processing on the plurality of input reflected wave signals to generate a plurality of received analog signals. The mathematical model of the received analog signal is represented by W ₀ (h, t). The receiving circuit 140 further converts the generated plurality of received analog signals into received digital signals. The mathematical model of the received digital signal is represented by V ₀ (h, t). Specifically, V ₀ (h, t) will be clarified by the explanation described later.

FIG. 3 is a block diagram showing a functional configuration of the signal generation circuit 110 of the radar device 100 according to the first embodiment. As shown in FIG. 3, the signal generation circuit 110 includes a local oscillator 111, a pulse generator 112, an in-pulse modulator 113, and an output unit 114.

The local oscillator 111 of the signal generation circuit 110 generates a local oscillation signal in the frequency band used. The generated local oscillation signal is output to the pulse generator 112 and the receiving circuit 140.

As the pulse generator 112, the in-pulse modulator 113, and the output unit 114, those of the prior art may be used.

FIG. 4 is a flowchart showing a processing step of the radar signal processing circuit 160 of the radar device 100 according to the first embodiment. As shown in FIG. 4, the radar signal processing circuit 160 includes a step of performing correlation processing (ST161), a compensation processing step of speed compensation candidates (ST162), a target candidate detection step (ST163), and speed measurement / distance measurement. Includes a step (ST164) of performing the process.

The correlation processing unit 161 of the radar signal processing circuit 160 modulates the reference signal and performs correlation processing for correlating the reference signal with the received digital signal (step shown in ST161 in FIG. 3). The correlation processing unit 161 performs correlation processing on a plurality of received digital signals and generates a plurality of corresponding pulse-compressed signals. In other words, the correlation processing unit 161 performs pulse compression. The details of the correlation processing performed by the correlation processing unit 161 will be clarified by the explanation based on the mathematical model described later.

ここで、ｍはＰＲＩ内におけるサンプリングに付される通し番号であり、０からＭ－１までの整数である。Δｔは、ＰＲＩ内のサンプリング周期を表す。すなわちｍΔｔはサンプリング開始からの時間を表すが、離散的な表現を用いて単に「時刻ｍ」と称されることもある。ＭはＰＲＩ内のサンプリングの数を表す。ｈは、前述したヒット番号である。
ｆ_０は、送信周波数を表す。Ｒ（ｈ、ｍΔｔ）は、目標までのレンジ、すなわち相対距離を表す。ｃは、光速を表す。Ｂ_０は、変調帯域幅を表す。Ｔ_０は、送信パルス長を表す。Ａ_０は、振幅を表す定数である。The mathematical model of the received digital signal is represented by _V0 . Specifically, the mathematical model of the received digital signal is expressed by the following equation (1).

Here, m is a serial number assigned to sampling in the PRI, and is an integer from 0 to M-1. Δt represents the sampling period in the PRI. That is, mΔt represents the time from the start of sampling, but it may be simply referred to as “time m” using a discrete expression. M represents the number of samplings in the PRI. h is the hit number described above.
f ₀ represents the transmission frequency. R (h, mΔt) represents the range to the target, that is, the relative distance. c represents the speed of light. B ₀ represents the modulation bandwidth. T ₀ represents the transmission pulse length. A ₀ is a constant representing the amplitude.

ここでν_ｃａｎ（ｑ）は、速度補償の候補値（以降、単に「速度補償候補」と称する）を表す。下添え字のｃａｎは、候補を意味するｃａｎｄｉｄａｔｅの最初の３文字である。式（２）の右辺におけるＡ_ｒｅｆは、参照信号の振幅である。The mathematical model of the reference signal is represented by V _ref (m). The subscript ref is the first three letters of reference, which means reference. Specifically, the mathematical model of the reference signal is expressed by the following equation (2).

Here, ν _can (q) represents a candidate value for speed compensation (hereinafter, simply referred to as “speed compensation candidate”). The subscript can is the first three letters of the candidate, which means a candidate. Aref on the right side of equation (2) is the amplitude of the _reference signal.

The radar device 100 according to the first embodiment includes a Doppler frequency term in the reference signal used as shown in the equation (2). Including the Doppler frequency term in the reference signal has the effect of suppressing Doppler coupling in the correlation processing. That is, including the Doppler frequency term in the reference signal has the effect of reducing the integration loss between the received signal and the reference signal.

ここで、ν_{ｃａｎ＿ｃｒｉ}は、速度補償候補の基準値（以降、単に「速度補償候補基準」と称する）を表す。下添え字のｃｒｉは、基準を表すｃｒｉｔｅｒｉｏｎの最初の３文字である。ｑは速度候補に付された通し番号（以降、「速度候補番号」と称する）であり、０からＱ－１までの整数である。Ｑは速度候補の総数（以降、単に「速度候補数」と称する）を表す。Δν_ｃａｎは、速度候補を変化させる刻み幅である。速度補償候補は、想定される目標（ターゲット）の速度の範囲内で設定されるとよい。本開示技術に係るレーダ装置１００は、速度候補数であるＱと、速度候補を変化させる刻み幅であるΔν_ｃａｎと、それぞれを設定可能である。
用語の「速度候補」と「速度補償候補」とは、結果的に同じ変数を指してもよいが、厳密には意味が異なり、区別して使用されるべきものである。「速度候補」の用語は、目標について検出すべき速度の候補値を意味するときに使用される。一方、「速度補償候補」の用語は、後述する速度補償に用いられる値を意味するときに使用される。速度補償候補は、速度候補から選定される。特に、速度候補に付される速度候補番号のｑと、後述する速度補償候補に付される速度補償候補番号のｑ_ｋとは、区別して用いられる。
用語の「目標」は、一般用語として辞書的な意味に用いられる場合と、レーダの測定対象を指す場合と、明確に区別される必要がある。そこで本明細書中では、前者の辞書的な意味では「目標」の用語は使わず、別の表現、例えば「目的、めあて」等を使って区別する。Specifically, the speed compensation candidate is represented by the following equation (3).

Here, ν _{can_cri} represents a reference value of a speed compensation candidate (hereinafter, simply referred to as “speed compensation candidate reference”). The subscript cri is the first three letters of the criterion representing the reference. q is a serial number assigned to the speed candidate (hereinafter referred to as “speed candidate number”), and is an integer from 0 to Q-1. Q represents the total number of speed candidates (hereinafter, simply referred to as "number of speed candidates"). Δν _can is a step size that changes the velocity candidate. The speed compensation candidate may be set within the range of the assumed target speed. In the radar device 100 according to the present disclosure technique, Q, which is the number of speed candidates, and Δν _can , which is the step size for changing the speed candidates, can be set respectively.
The terms "velocity candidate" and "velocity compensation candidate" may eventually refer to the same variable, but they have different meanings and should be used separately. The term "velocity candidate" is used to mean a candidate velocity value to be detected for a target. On the other hand, the term "speed compensation candidate" is used to mean a value used for speed compensation described later. Speed compensation candidates are selected from speed candidates. In particular, the speed candidate number q assigned to the speed candidate and the speed compensation candidate number q _k attached to the speed compensation candidate described later are used separately.
The term "goal" needs to be clearly distinguished between when it is used in a dictionary sense as a general term and when it refers to a radar measurement target. Therefore, in the present specification, the term "goal" is not used in the former dictionary sense, but another expression such as "purpose, target" is used for distinction.

ここでΔν_ａｍｂは、Ｔ_ｐｒｉ（ｈ）に基づくアンビギュイティが生じることなく測定可能な速度の間隔（以降、単に「アンビギュイティがない速度間隔」と称する）、を表す。下添え字のａｍｂは、アンビギュイティの英語表現であるａｍｂｉｇｕｉｔｙの最初の３文字である。アンビギュイティは、あいまいさを意味する。またＮ_{ν，ａｍｂ}は、正の整数である。具体的にΔν_ａｍｂは、以下の式（５）で与えられる。

このようにΔν_ｃａｎを定義することで、スロータイム方向のフーリエ変換後の信号どうしを同一のサンプリング番号で表すことができ、ペアリング及び検出の効率が良くなる。As the Δν _can used in the equation (3), specifically, the one given by the following equation (4) may be adopted.

Here, Δν _amb represents a velocity interval that can be measured without occurrence of ambiguity based on T _pri (h) (hereinafter, simply referred to as “velocity interval without ambiguity”). The subscript amb is the first three letters of ambiguity, the English expression for ambiguity. Ambiguity means ambiguity. Further, N _{ν and amb} are positive integers. Specifically, Δν _amb is given by the following equation (5).

By defining Δν _can in this way, the signals after the Fourier transform in the slow time direction can be represented by the same sampling number, and the efficiency of pairing and detection is improved.

FIG. 5 is a graph showing an image of the velocity interval without ambiguity and the detected intensity distribution in the velocity compensation candidate. In the graph shown in FIG. 5, the vertical axis represents intensity and the horizontal axis represents velocity. Further, the plot of the star mark in the graph shown in FIG. 5 shows the true value for the target, and the plot of the cross mark shows the candidate speed of the target including the speed ambiguity.

The graph shown in FIG. 5 shows an example in which the intensity is maximized at the true value of the target velocity. This phenomenon indicates that the range walk component after motion compensation remains as the velocity deviates from the true value, and the intensity of the integrated result decreases.

The Δν _can shown in the equation (4) appears as N _{ν, amb} times the plot interval of the adjacent star or cross in the graph shown in FIG. FIG. 5 shows that the Δν _can adopted by the present disclosure technique is wider than the Δν _amb . In FIG. 5, an example in which N ν and _amb are set to 2 is illustrated, but N _{ν and amb} may be integers larger than 2.

In the graph shown in FIG. 5, qk represents the _kth speed compensation candidate number. In the example shown in FIG. 5, since N _{ν and amb} are set to 2, the speed compensation candidate numbers are q ₀ = 0, q ₁ = 2, q ₂ = 4, .... That is, unlike the speed candidate number q, the speed compensation candidate number q _k can be a non-continuous and discrete number. Since the speed compensation candidate number q _k is a number assigned in order from 0 for each speed interval without ambiguity, it is a multiple of N _{ν and amb} .
Further, P (q _k ) represents the intensity at q _k . The graph shown in FIG. 5 shows an example in which the k + 1th velocity compensation candidate number, q _{k + 1} , matches the true value of the target velocity.

ここでΔＰ_ｄｅｓは、設計した強度差分（以降、「強度差分設計値」と称す）である。下添え字のｄｅｓは、設計を表すｄｅｓｉｇｎの最初の３文字である。ΔＰ_ｄｅｓは、例えば３［ｄｂ］等の、経験的に得られた値であってよい。このように強度差分設計値は、Ｎ_{ν，ａｍｂ}をどの値にするか、すなわち速度補償候補の間隔をどの程度とするか、ということの判断の材料にしてよい。The modulation band, pulse width, and velocity compensation candidate step may be determined so that the difference between P (q _k ) and P (q _{k + 1} ) is larger than the pre-designed design value.

Here, ΔP _des is the designed strength difference (hereinafter referred to as “strength difference design value”). The subscript des is the first three letters of the design representing the design. ΔP _des may be an empirically obtained value such as 3 [db]. In this way, the strength difference design value may be used as a material for determining which value N _{ν and amb} should be, that is, how much the interval between the speed compensation candidates should be.

ここで、式（７）右辺の総和区間に登場するＭ_Ｐは、パルス内のサンプリング点の数を表す。また、式（７）の右辺に登場する上線は、複素共役の操作を表す。すなわちＶ_ｒｅｆ（ｐ）に上線を付したものは、Ｖ_ｒｅｆ（ｐ）の複素共役である。なお、式（７）は代表的な相関演算に用いられる離散時間の畳み込み演算であるが、相関演算にはいろいろな流儀のものが存在する。本開示技術に係るレーダ信号処理回路１６０は、式（７）に代えて、公知の別の流儀の相関演算が用いられてもよい。パルス圧縮処理された信号（以降、「パルス圧縮信号」と称する）は、式（７）の左辺のＦ_{ｖ０ｖｒｅｆ}（ｈ，ｍ）として表される。Specifically, the correlation processing performed by the correlation processing unit 161 may be performed by the following mathematical expression processing.

Here, MP appearing in the _summation section on the right side of equation (7) represents the number of sampling points in the pulse. The overline appearing on the right side of the equation (7) represents the operation of the complex conjugate. That is, what is overlined in V _ref (p) is the complex conjugate of V _ref (p). The equation (7) is a discrete-time convolution operation used for a typical correlation operation, but there are various styles of the correlation operation. In the radar signal processing circuit 160 according to the present disclosure technique, a correlation calculation of another known style may be used instead of the equation (7). The pulse-compressed signal (hereinafter referred to as “pulse-compressed signal”) is represented as F _v0vref (h, m) on the left side of the equation (7).

The compensation processing unit 162 of the radar signal processing circuit 160 performs motion compensation processing on the pulse compressed signal. As shown in FIG. 1, the radar device 100 according to the present disclosure technique is assumed to be a monostatic radar using an antenna 130 for both transmission and reception. In the case of monostatic radar, since the receiving gate cannot be opened during pulse transmission, a so-called blind area is created due to the lack of data in the received signal. The motion compensation processing performed by the compensation processing unit 162 interpolates the information in this blind region from the information of the received signals of adjacent pulses on the time axis. The pulse compression signal subjected to the motion compensation process is referred to as a motion compensation candidate in the present specification.

ここで、式（８）の左辺のスクリプト書体のＫは、その大かっこの中身に対して速度補償処理を行う操作を表している。また、式（８）の左辺のスクリプト書体のＦは、その大かっこの中身に対してフーリエ変換を行う操作を表している。速度補償処理とフーリエ変換とを合わせた処理が、運動補償処理である。
式（８）の右辺に現れた関数の引数のｋは、離散フーリエ変換を行うことにより得られるスペクトルに付される通し番号である。式（８）の詳細は、後述の説明により明らかとなる。The compensation processing unit 162 of the radar signal processing circuit 160 performs a Fourier transform on the pulse compression signal subjected to the speed compensation processing.

Here, K in the script typeface on the left side of the equation (8) represents an operation of performing speed compensation processing on the contents of the brackets. Further, F of the script typeface on the left side of the equation (8) represents an operation of performing a Fourier transform on the contents of the parentheses. The process that combines the velocity compensation process and the Fourier transform is the motion compensation process.
The argument k of the function appearing on the right side of the equation (8) is a serial number assigned to the spectrum obtained by performing the discrete Fourier transform. The details of the formula (8) will be clarified by the explanation described later.

In the compensation processing performed by the compensation processing unit 162, it is advisable to determine in advance which speed candidate is used for compensation. For example, it may be decided in advance that the compensation process is compensated by the speed compensation candidate ν _can (q) whose speed candidate number is q.

FIG. 6 is a graph showing an image of an assumed speed interval of a target, a speed interval without ambiguity, and a candidate of a target.
Generally, when the PRF (Pulse Repetition Frequency), which is the reciprocal of PRI, is lowered for the purpose of observing a long distance, the so-called virtual image due to ambiguity does not appear at the speed interval (“speed interval without ambiguity”). Same as) becomes narrower.
Also, in general, when the target is moving at high speed, it is necessary to take a wide speed interval. The radar according to the prior art must take a large number of speed candidates and perform a speed search in order to obtain a target speed with high accuracy.
FIG. 6 shows a situation in which the assumed speed interval of the target is wide and the PRI is small. Further, FIG. 6 shows an example in which the intervals of speed compensation candidates are aligned with the intervals of velocities that can be solved without ambiguity (also the same as “speed intervals without ambiguity”). FIG. 6 also shows that the most probable value for the target must be selected by velocity search from the large number of candidates generated by velocity ambiguity.

ここで式（９）の左辺のＲ_ｃａｎ（ｈ）は、補償距離を表す。用いられているアルファベットのＲは、距離を意味するレンジの英語表記であるＲａｎｇｅの頭文字である。引数のｈは、前述のヒット番号である。式（９）で示される数理モデルは、単に速度と、時間と、距離と、の関係を表したものである。なお式（９）の右辺にマイナスの符号が用いられているのは、速度の軸と、距離の軸と、それぞれの向きが反対になるように定義したに過ぎない。

ここで式（１０）の左辺のＶ_ＲＣＡＮ（ｈ）は、補償距離から作成した補償信号を表す。なお、式（１０）の右辺には振幅を示していないが、補償信号についての数理モデルは、式（１０）の右辺に振幅を表す定数が乗じられていてもよい。Using ν _can , which is a speed compensation candidate, the compensation distance and compensation signal can be calculated based on the following mathematical model.

Here, R _can (h) on the left side of the equation (9) represents the compensation distance. The letter R used is an acronym for Range, which is an English notation for the range meaning distance. The argument h is the hit number described above. The mathematical model represented by Eq. (9) simply expresses the relationship between velocity, time, and distance. It should be noted that the negative sign used on the right side of the equation (9) is merely defined so that the axes of velocity and the axis of distance are opposite to each other.

Here, V _RCAN (h) on the left side of the equation (10) represents a compensation signal created from the compensation distance. Although the amplitude is not shown on the right side of the equation (10), in the mathematical model for the compensation signal, a constant representing the amplitude may be multiplied on the right side of the equation (10).

式（１１）は、式（８）の速度補償処理の部分についての詳細である。
ここでスクリプト書体のＦに下添え字のｒａｎｇｅは、その大かっこの中身に対して距離方向のフーリエ変換を行う操作を表している。距離方向のフーリエ変換は、ファストタイム領域からファストタイム周波数領域への変換を意味する。距離方向の逆フーリエ変換は、その逆の変換、すなわちファストタイム周波数領域からファストタイム領域への変換を意味する。By using the results of the equations (7) and (10), the signal after the speed compensation (hereinafter, simply referred to as "the signal after the speed compensation") is derived.

Equation (11) is a detail about the speed compensation processing portion of equation (8).
Here, the subscript range in F of the script typeface represents an operation of performing a Fourier transform in the distance direction with respect to the contents of the parentheses. The Fourier transform in the distance direction means the transformation from the fast time domain to the fast time frequency domain. The inverse Fourier transform in the distance direction means the inverse transform, that is, the transformation from the fast-time frequency domain to the fast-time domain.

式（１２）は、式（８）のフーリエ変換の部分についての詳細である。式（１２）は、一般的な離散フーリエ変換を表すものと相違ない。式（１２）の右辺が示すように、変数のｋは、離散フーリエ変換を行うときのサンプリング番号である。式（１２）の例は、あるサンプリングから次のサンプリングまでに、すなわちｈが１つ増えるごとに、位相が２πｋをＫで割った値の分だけ進む基本波による離散フーリエ変換であることを示している。基本波に相当する式（１２）右辺のｅｘｐの項は、複素平面上の単位円を等分割した点を表すため、回転子と呼ばれることもある。By further Fourier transforming the left side of the equation (11) in the slow time direction, the result obtained by the equation (11), that is, the signal after the motion compensation processing (hereinafter, simply referred to as "the signal after the motion compensation") is obtained. Be guided.

Equation (12) is a detail about the Fourier transform part of equation (8). Equation (12) is no different from that representing a general discrete Fourier transform. As shown by the right side of equation (12), the variable k is the sampling number when performing the discrete Fourier transform. The example of Eq. (12) shows that it is a discrete Fourier transform with a fundamental wave whose phase advances by the value of 2πk divided by K from one sampling to the next, that is, for each increment of h. ing. The exp term on the right-hand side of equation (12), which corresponds to the fundamental wave, is sometimes called a rotor because it represents the point where the unit circle on the complex plane is equally divided.

The above is a general pulse Doppler technique with a speed compensation process added, but the disclosed technique is not limited to this. The radar device 100 according to the present disclosure technique may be subjected to another process to the same effect.

The target candidate detection unit 163 of the radar signal processing circuit 160 calculates the relative distance and the relative speed with respect to the target candidate (step shown in ST163 in FIG. 4).
It is conceivable that the target candidate detection unit 163 detects the target candidate by using an algorithm such as CA-CFAR (Cell Average-Constant False Allarm Rate).
CA-CFAR can maximize the detection probability so that the false alarm probability becomes a constant value. That is, the CA-CFAR has a feature that false detection can be controlled and a target candidate can be detected based on the signal strength without detecting noise as much as possible.

As shown in FIG. 4, the radar signal processing circuit 160 repeats the processing steps of ST161, ST162, and ST163 for the number of speed candidates.

式（１３）を用いて各ピーク強度の距離をシフトすれば、複数の速度候補のピーク強度をインコヒーレント積分が可能である。ここで、距離方向シフト後の運動補償後信号は、以下の式で表される。

ただし、ｒ＿ｓｈｉｆｔは、その大かっこの中身に対して、運動補償後信号（Ｆ_ＦＦＴ）を速度候補番号と距離間隔（Δｒ）の積で表される量で距離シフトする処理を表す。By the above processing, the detection results of the target candidates for the number of speed candidates can be obtained, and the detection performance can be further improved by incoherently integrating each of them. FIG. 7 is a graph in which the peak intensities of each velocity candidate are arranged. The horizontal axis represents speed and the vertical axis represents distance. If the speed candidates are set as an integral multiple of the speed that can be solved without ambiguity, the peak intensities are lined up at distance intervals according to the speed candidates in the same speed direction and in the distance direction because they are integrated in the same speed bin. Letting the speed interval of the speed candidate be Δν, Δr of the distance interval corresponding to the speed candidate is expressed by the following equation.

By shifting the distance of each peak intensity using the equation (13), it is possible to incoherently integrate the peak intensities of a plurality of velocity candidates. Here, the motion-compensated signal after the shift in the distance direction is expressed by the following equation.

However, r_shift represents a process of distance-shifting the motion-compensated signal ( _FFFT ) with respect to the contents of the large parenthesis by an amount represented by the product of the velocity candidate number and the distance interval (Δr).

ここで、変数に付された下添え字のＩＣは、インコヒーレントの英語表記に由来する。なお、式（１５）は積分記号ではなく総和記号が用いられているが、これは単に離散的に積分を行ったことを表しているに過ぎない。The incoherent integral of the peak intensities of multiple velocity candidates is expressed by the following equation.

Here, the subscript IC attached to the variable is derived from the incoherent English notation. In addition, although the sum symbol is used in the equation (15) instead of the integral symbol, this merely indicates that the integral was performed discretely.

The radar device 100 according to the present disclosure technique improves the detection performance by using the F _{FFT_IC} (k, m) obtained by incoherent integration instead of using each of the _F'FFT (k, m). It has the effect of being able to make it.

The speed measuring / distance measuring unit 164 of the radar signal processing circuit 160 calculates the peak intensities of each of the speed candidates, and calculates the peak intensities corresponding to each of the speed compensation candidates. Further, the speed measuring / distance measuring unit 164 calculates an estimated value of a plausible target speed from the information of the peak having the maximum intensity (step shown in ST164 in FIG. 4).

FIG. 8 is a graph showing an image of calculating an estimated value of a target speed from the peak intensity of each speed compensation candidate. In FIG. 8, the horizontal axis of the graph represents the value of the speed compensation candidate, and the vertical axis of the graph represents the intensity. FIG. 8 shows that the intensity is maximized when the value of the speed compensation candidate is closest to the true value of the target speed. Being able to estimate something close to the true value of the target velocity means that the velocity ambiguity is solved and that the integration efficiency of the target velocity within the PRF in the range of velocity candidates does not deteriorate, that is, it improves. ..
FIG. 8 also shows that the target velocity can be estimated from the plot tendency even with discrete velocity compensation candidates wider than the velocity ambiguity interval. In FIG. 8, the plot shows a linear tendency, that is, the linear and intensity decreases as the velocity estimate deviates from the true value. However, it is possible to estimate plausible values even if the plot cannot always be approximated by two straight lines.

For p (0), ..., p (q _k ), p (q _{k + 1} ), ... Shown in FIG. 8, the speed compensation candidate numbers are q _k (q ₀ = 0, q ₁ = 2, q ₂ = 4). , ...,). In the example shown in FIG. 8, the graph increases monotonically on the left side and decreases monotonically on the right side. More specifically, the graph increases monotonically up to p (q _k ) and decreases monotonically after p (q _k ). Therefore, in the example of FIG. 8, it can be inferred that the intensity is maximized at the velocity between q _k and q _{k + 1} .

In this way, when the intensity is plotted for each speed compensation candidate, the plot changes from an increasing tendency to a decreasing tendency at a certain speed compensation candidate number. The present disclosure technique divides the first half group of plots that are increasing and the second half group of plots that is decreasing, and mathematically models each of them. The mathematical model shown in FIG. 8 uses, but is not limited to, a linear approximation based on the least squares method. The mathematical model may be a higher-dimensional curve approximation or may be based on another interpolation method. Then, the target velocity can be obtained as an intersection of a curve or the like which is a mathematical model of the first half group and a curve or the like which is a mathematical model of the second half group.

ここで式（１６）の左辺のν_ｒｅｓは、目標の速度の推定結果である。下添え字のｒｅｓは、結果を意味するｒｅｓｕｌｔの最初の３文字である。式（１６）の右辺第１項のνにダッシュが付されたものは、速度補償候補の中で最も強度が高いものである。図９におけるνにダッシュが付されたプロットは、「最も近い速度プロット」と称する。式（１６）の右辺第２項のｄν_ｒｅｓは、２つの数理モデルの交点の位置から求めた速度差分値を表す。
すなわち本開示技術に係るレーダ装置１００は、速度候補のそれぞれの検出強度に基づいて、速度候補を、増加傾向にある前半グループと、減少傾向にある後半グループとに分け、前半グループと後半グループのそれぞれで最小二乗法に基づいた近似直線又は近似曲線を求め、近似直線又は近似曲線のそれぞれが交わる交点として前記目標の速度を求める。The estimated value of the target velocity obtained as the intersection of the two mathematical models can be expressed as follows.

Here, ν _res on the left side of the equation (16) is the estimation result of the target velocity. The subscript res is the first three letters of the result, which means the result. The ν in the first term on the right side of the equation (16) with a dash is the one with the highest strength among the speed compensation candidates. The plot in which ν is dashed in FIG. 9 is referred to as the “closest velocity plot”. The second term dν _res on the right side of equation (16) represents the velocity difference value obtained from the position of the intersection of the two mathematical models.
That is, the radar device 100 according to the present disclosure technology divides the speed candidates into an increasing first half group and a decreasing second half group based on the respective detection intensities of the speed candidates, and divides the speed candidates into the first half group and the second half group. An approximate straight line or an approximate curve based on the least squares method is obtained for each, and the target speed is obtained as an intersection where each of the approximate straight lines or the approximate curves intersects.

ここで式（１７）の左辺のΔｒ_ｃａｎは、速度候補の刻み幅であるΔν_ｃａｎに対応する距離候補の刻み幅である。
式（１６）及び式（１７）から、目標の距離の推定値は、以下のように求められる。

ここで式（１８）の左辺のｒ_ｒｅｓは、目標の距離の推定結果である。式（１８）の右辺第１項のｒにダッシュが付されたものは、式（１６）の右辺第１項のνにダッシュが付されたもの、すなわち最も近い速度プロット、に対応する距離である。FIG. 9 is a graph showing an image of calculating the estimated distance from the estimated speed of the target. As shown in the lower graph of FIG. 9, the estimated distance can be obtained from the relational expression between the distance and the speed.

Here, Δr _can on the left side of the equation (17) is a step width of the distance candidate corresponding to the step width of the velocity candidate Δν _can .
From the equations (16) and (17), the estimated value of the target distance is obtained as follows.

Here, r _res on the left side of the equation (18) is the estimation result of the target distance. The r in the first term on the right side of the equation (18) with a dash is the distance corresponding to the ν in the first term on the right side of the equation (16) with a dash, that is, the closest velocity plot. be.

例えば図８に例示したようにｑ_ｋ番目の速度補償候補とｑ_ｋ＋１番目の速度法相候補との間に強度の極大値が存在するとわかった場合、ｑ_ｋ番目とｑ_ｋ＋１番目との間における速度補償候補をより細かい刻みで変化させ、式（１９）に基づいて、強度が最大となる速度を求めることもできる。
すなわち実施の形態１に係るレーダ装置１００は、速度補償候補の検出結果から最大強度と2番目に最大となる強度を選択し、前記目標の速度の推定を行う。In general, there is a method of obtaining a maximum value and a minimum value from a derivative. That is, the slope at the extreme value is 0, and this property can be applied to obtain the velocity candidate having the highest intensity in the present disclosed technique. The velocity candidate with the highest intensity can be obtained by obtaining the difference in intensity between adjacent velocity candidates as follows.

For example, if it is found that there is a maximum intensity value between the qkth velocity compensation candidate and the _{qk + 1st} velocity method candidate as illustrated in FIG. 8, the velocity between the qkth and _{qk + 1st} velocities. It is also possible to change the compensation candidates in finer steps and obtain the speed at which the intensity is maximized based on the equation (19).
That is, the radar device 100 according to the first embodiment selects the maximum intensity and the second maximum intensity from the detection results of the speed compensation candidates, and estimates the target speed.

ここで、ｐ_Σは信号強度の和信号であり、ｐ_Δは信号強度の差信号である。
図１０に示されたディスクリカーブの横軸は、速度である。また、図１０に示されたディスクリカーブの縦軸は、以下のように算出される、信号強度に関する和信号対差信号の比である。

The target velocity may be estimated using a discrecurve. FIG. 10 is an image diagram showing the relationship between the ratio of the sum and difference signals with respect to the signal strength and the speed. The curve in FIG. 10 is a discrete curve prepared in advance. Here, the sum signal and the difference signal regarding the signal strength are obtained as follows.

Here, p _Σ is a sum signal of signal strength, and p _Δ is a difference signal of signal strength.
The horizontal axis of the discrete curve shown in FIG. 10 is velocity. Further, the vertical axis of the discrete curve shown in FIG. 10 is the ratio of the sum signal pairing signal regarding the signal strength calculated as follows.

In FIG. 10, the first term on the right side of Eq. (14) and the second term on the right side of Eq. (14) are obtained using the discrete curve, and ν _res is the estimation result of the target velocity according to Eq. (14). It shows that can be obtained.

FIG. 11 is a diagram showing an example of the hardware configuration of the signal processing circuit 170 of the radar device 100 according to the present disclosure technique. Here, the signal processing circuit 170 is hardware that realizes the PRI control unit 150 and the radar signal processing circuit 160, which are the functional configurations of the radar device 100. The signal processing circuit 170 includes a CPU (Central Processing Unit, a central processing unit, a processing unit, an arithmetic unit, a microprocessor, a microcomputer, a processor, and a DSP) that executes a program stored in a memory even if it is dedicated hardware. It may be). Note that FIG. 11 illustrates a case where the signal processing circuit 170 is realized by a CPU that executes a program stored in the memory. The signal processing circuit 170 includes a processor 171, a memory 172, a storage device 173, an input / output interface 174, and a signal path 175. Information from the receiving circuit 140 is input to the signal processing circuit 170, and an external display 200 is connected to the signal processing circuit 170. Here, the term "processor" is used as a general noun, and the processor 171 of the component constituting the signal processing circuit 170 is used as a proper noun, and the two are distinguished from each other. Further, the term "memory" is used as a general noun, and the memory 172 of the component constituting the signal processing circuit 170 is used as a proper noun, and both are distinguished.

When the signal processing circuit 170 is dedicated hardware, the signal processing circuit 170 corresponds to, for example, a single circuit, a decoding circuit, a programmed processor, a parallel programmed processor, an ASIC, an FPGA, or a combination thereof. do. The functions of the PRI control unit 150 and the radar signal processing circuit 160 may be realized by different processing circuits, or may be realized by one processing circuit collectively.

When the signal processing circuit 170 is configured by a CPU, the functions of the PRI control unit 150 and the radar signal processing circuit 160 are realized by software, firmware, or a combination of software and firmware. Both software and firmware are written as programs and stored in memory. The signal processing circuit 170 realizes the functions of each part by reading and executing the program stored in the memory. That is, when the function is executed by the signal processing circuit 170, the radar device 100 has a process of setting PRI, a step of performing correlation processing (ST161), a compensation processing step of speed compensation candidates (ST162), and a target. A memory for storing a program in which the candidate detection step (ST163) and the step of performing speed measurement / distance measurement processing (ST164) are to be executed as a result is provided. Further, it can be said that these programs cause a computer to execute the procedures and methods of the PRI control unit 150 and the radar signal processing circuit 160.

The processor 171 that executes the program may be configured by, for example, an LSI.

Here, the memory may be, for example, a non-volatile or volatile semiconductor memory such as RAM, ROM, flash memory, EPROM, EEPROM or the like. Further, the memory may be a magnetic disk, a flexible disk, a compact disk, a mini disk, a DVD, or the like. The memory may be in the form of HDD, SSD, or the like.
FIG. 11 includes a memory 172 and a storage device 173 to show that the memory can take various forms. The memory 172 and the storage device 173 are used for digital signal processing, a program memory for storing various program codes to be executed by the processor 171 and a work memory used when the processor 171 executes digital signal processing. Includes a temporary storage memory in which the data to be stored is stored.

The functions of the PRI control unit 150 and the radar signal processing circuit 160 may be partially realized by dedicated hardware and partially realized by software or firmware. For example, the PRI control unit 150 realizes its function by a processing circuit as dedicated hardware, and the radar signal processing circuit 160 realizes its function by reading and executing a program stored in the memory. It is also possible.

As described above, the signal processing circuit 170 can realize the functions of the PRI control unit 150 and the radar signal processing circuit 160 by hardware, software, firmware, or a combination thereof.

Since the radar device 100 according to the first embodiment has the above configuration, it is possible to estimate the target speed based on a plurality of speed candidates in a wide range in which the speed ambiguity appears. Due to this effect, the radar device 100 according to the first embodiment can estimate the target distance and speed with a smaller amount of calculation than the conventional one without being affected by the speed ambiguity.

Embodiment 2.
The radar device 100 according to the second embodiment includes a radar signal processing circuit 160 (160B) having a different configuration from the radar signal processing circuit 160 according to the first embodiment. Unless otherwise specified, the second embodiment uses the same reference numerals as those of the first embodiment, and duplicate description is omitted as appropriate.

FIG. 12 is a block diagram showing a functional configuration of the radar signal processing circuit 160 (160B) of the radar device 100 according to the second embodiment. As shown in FIG. 12, the radar signal processing circuit 160 (160B) includes a CZT pulse Doppler processing unit 165 instead of the correlation processing unit 161 and the compensation processing unit 162. As shown in FIG. 12, the CZT pulse Doppler processing unit 165 is provided in the front stage of the radar signal processing circuit 160 (160B), the target candidate detection unit 163 is provided in the subsequent stage, and the speed measurement / distance measurement unit 164 is further in the subsequent stage. It is provided. CZT is an acronym for Chirp Z-Transform and means chirp z-transform.

The chirp z-transform is the FFT algorithm proposed by Bluestain in 1968, and is also called the FFT algorithm of Bluestain. The chirp z-transform has a feature that discrete data of arbitrary length can be fast Fourier transformed.

FIG. 13 is a flowchart showing a processing step of the radar signal processing circuit 160 (160B) of the radar device 100 according to the second embodiment. As shown in FIG. 13, the processing steps of the radar signal processing circuit 160 (160B) are the CZT pulse Doppler processing step (ST165) performed by the CZT pulse Doppler processing unit 165 and the target candidate detection step (ST163) performed by the target candidate detection unit 163. ) And the speed measuring / measuring step (ST164) performed by the speed measuring / measuring unit 164.

The radar signal processing circuit 160 (160B) according to the second embodiment performs correlation processing and speed compensation using the chirp z-transform. Specifically, the CZT pulse Doppler processing unit 165 of the radar signal processing circuit 160 (160B) performs CZT processing in the hit direction in consideration of speed compensation on the received digital signal represented by the equation (1), and further performs correlation processing. (Step shown in ST165 in FIG. 13).

FIG. 14 is a flowchart showing the details of the CZT pulse Doppler processing step (ST165) in the flowchart shown in FIG. As shown in FIG. 14, the CZT pulse Doppler processing step (ST165) includes a processing step of FFT in the distance direction (ST165A), a processing step of CZT in the hit direction (ST165B), and a processing step of multiplying by a reference function (ST165C). And the processing step of the IFF in the distance direction (ST165D).

ここで、ｍ_ｐは、１つのＰＲＩにおけるサンプリング番号である。Ｍ_ｐは、１つのＰＲＩにおけるサンプリングの総数である。Ｍ_ｆｆｔは、フーリエ変換点の総数である。ｋ_ｒは、距離方向のＦＦＴの結果に付される通し番号である。別の表現をすればｋ_ｒは、ファストタイム周波数スペクトルに付されるビン番号である。ビン番号の詳細は、後述の説明により明らかとなる。The CZT pulse Doppler processing unit 165 executes FFT processing in the distance direction exemplified by the following formula for the received digital signal (step shown by ST165A in FIG. 14).

Here, _mp is a sampling number in one PRI. M _p is the total number of samplings in one PRI. M _fft is the total number of Fourier transform points. _kr is a serial number assigned to the result of the FFT in the distance direction. In other words, _kr is a bin number assigned to the fast-time frequency spectrum. The details of the bin number will be clarified by the explanation described later.

ここでスクリプト書体のＦに下添え字のＣＺＴは、その大かっこの中身に対してヒット方向のチャープｚ変換を行う操作を表している。ヒット方向とは、要するにヒット番号が増減する方向とイメージすればよく、距離方向又はレンジ方向と同義である。このことは、式（２３）が、式（２２）の距離方向のＦＦＴの結果に対して行う処理である、ということからも理解できる。言い換えれば実施の形態２に係るレーダ装置１００は、受信信号に対して、ファストタイム方向を周波数領域に変換し、チャープｚ変換を用いている。
ｈ_ｃｚｔは、ＣＺＴ処理後の信号のドップラ速度のビン番号を表す。ビン番号は、ＦＦＴビンに付された通し番号である。ＦＦＴビン又はビンは、英語のｂｉｎに由来し、周波数スペクトルが短冊のように並んでいることから、その短冊状のものを指す用語として使われている。Ｈ_ｃｚｔは、ＣＺＴ処理後に得られるビンの総数である。ｈ_ｃｚｔがドップラ速度のビン番号であることと対比して、ｋ_ｒをレンジのビン番号と称してもよい。The CZT pulse Doppler processing unit 165 executes the processing of the CZT in the hit direction exemplified by the following mathematical formula (step shown by ST165B in FIG. 14).

Here, CZT, which is a subscript to F in the script typeface, represents an operation of performing a chirp z-transform in the hit direction for the contents of the parentheses. The hit direction can be imagined as a direction in which the hit number increases or decreases, and is synonymous with a distance direction or a range direction. This can be understood from the fact that the equation (23) is a process performed on the result of the FFT in the distance direction of the equation (22). In other words, the radar device 100 according to the second embodiment converts the fast time direction into the frequency domain with respect to the received signal, and uses the chirp z-transform.
h _czt represents the bin number of the Doppler velocity of the signal after CZT processing. The bin number is a serial number attached to the FFT bin. The FFT bin or bin is derived from the English bin, and since the frequency spectra are arranged like a strip, it is used as a term to refer to the strip-shaped one. H _czt is the total number of bins obtained after CZT treatment. In contrast to h _czt being the bin number of the Doppler velocity, kr may be referred to as the bin number of the range _.

ここで式（２４）の左辺のＡ_ｋｒは、ｋ_ｒに対応する変換の開始位相に係る回転子である。ｆ_ｋｒは、ｋ_ｒに位置するスペクトルのファストタイム周波数である。ν_ｓｔは、変換開始の速度である。ν_ｓｔの下添え字のｓｔは、開始を意味するｓｔａｒｔの最初の２文字である。
式（２５）の左辺のＷ_ｋｒ（ｈ_ｃｚｔ）は、ｋ_ｒに対応する変換の位相変化幅に係る回転子である。Δν_ｃｚｔは、速度の変化幅である。
言い換えれば実施の形態２に係るレーダ装置１００は、補償処理部１６２において、チャープｚ変換を用いて位相補償を実施し、予め設定された速度による前記運動補償候補で前記運動補償を行う。The rotor term on the right side of equation (21) is specifically defined as follows.

Here, A _kr on the left side of the equation (24) is a rotor related to the start phase of the conversion corresponding to _kr . f _kr is the fast-time frequency of the spectrum located at _kr . ν _st is the speed at which conversion starts. The subscript st of ν _st is the first two letters of start, which means start.
W _kr (h _czt ) on the left side of the equation (25) is a rotor related to the phase change width of the conversion corresponding to kr _. Δν _czt is the range of change in velocity.
In other words, the radar device 100 according to the second embodiment performs phase compensation using the chirp z-transform in the compensation processing unit 162, and performs the motion compensation with the motion compensation candidate at a preset speed.

There is no difference between the process represented by the equation (22) and the process represented by the equation (23) being an FFT. The FFT represented by the formula (22) is the first FFT and is sometimes referred to as a Range-FFT. The FFT represented by the formula (23) is a second FFT for the result of the first FFT, and is sometimes referred to as a Doppler-FFT. Generally, by performing two FFTs of Range-FFT and Doppler-FFT, it is possible to measure the velocity of each of a plurality of objects at the same distance.

式（２６）の左辺のν_ｃｚｔ（ｈ_ｃｚｔ）は、ドップラ速度のビン番号がｈ_ｃｚｔであるときの相対速度を表す。The relative velocities to be obtained by CZT processing appear in equations (24) and (25). Specifically, the relative velocity is a value according to the following formula.

The ν _czt (h _czt ) on the left side of the equation (26) represents the relative speed when the bin number of the Doppler speed is h _czt .

ここでν_ｅｎは、変換終了の速度である。ν_ｅｎの下添え字のｅｎは、終了を意味するｅｎｄの最初の２文字である。式（２７）は、単に、速度の全体幅をサンプリングの点の数で割ることで刻み幅が求められることを表している。The Δν _czt used in the formula (25) has the relationship shown below with H _czt .

Here, ν _en is the speed at which the conversion ends. The subscript en of ν _en is the first two letters of end, which means the end. Equation (27) simply expresses that the step size can be obtained by dividing the total width of the velocity by the number of sampling points.

ここでｋ_νは、或る定数である。ｄ_νは、速度分解能、すなわち速度の刻み幅について最も小さくできるもの、を表す。ｄ_νは、数式で表せば、以下のように示せる。

The ν _st and ν _en used in the formula (27) have the relationship shown below with the ν _can .

Here, k _ν is a constant. d _ν represents the velocity resolution, that is, the one that can be made the smallest with respect to the step size of the velocity. d _ν can be expressed as follows when expressed by a mathematical formula.

このように式（３０）に示される参照信号は、式（２６）に示された相対速度が使われている。Similar to the above processing for the received digital signal, the Fourier transform is also performed on the reference signal. The reference signal used in the second embodiment has the same structure as the reference signal of the first embodiment shown in the equation (2), but the details are shown below.

As described above, the reference signal shown in the equation (30) uses the relative velocity shown in the equation (26).

なお、式（３１）に示されたフーリエ変換は、式（２２）で示されたものと同じである。The Fourier transform performed on the reference signal is, for example, as follows.

The Fourier transform shown in Eq. (31) is the same as that shown in Eq. (22).

The CZT pulse Doppler processing unit 165 performs a process of multiplying the received digital signal obtained by Fourier transform and the reference function, respectively (step shown by ST165C in FIG. 14).

式（３３）により得られたＲ_ｐｃ（ｈ_ｃｚｔ、ｌ）は、相関処理及び補償処理が施された信号である。Ｒ_ｐｃ（ｈ_ｃｚｔ、ｌ）は、実施の形態１における式（１２）で示されたＦ_ＦＦＴ（ｋ，ｍ）と等価なものである。よって、以降の処理は、実施の形態１に示されたものを行えばよい。The CZT pulse Doppler processing unit 165 performs an inverse Fourier transform in the distance direction on the result of multiplying the two signals shown in the equation (32) (step shown by ST165D in FIG. 14).

The R _pc (h _czt , l) obtained by the equation (33) is a signal subjected to the correlation processing and the compensation processing. R _pc (h _czt , l) is equivalent to _FFFT (k, m) represented by the formula (12) in the first embodiment. Therefore, the subsequent processing may be performed as shown in the first embodiment.

As described above, since the radar device 100 according to the second embodiment has the above configuration, in addition to the effect shown in the first embodiment, the integral loss due to the Doppler coupling is reduced, and the distance shift due to the Doppler coupling is performed. It has the effect of being able to be made smaller.

The hardware of the radar device 100 according to the second embodiment may be realized with the same configuration as that of the first embodiment.

Embodiment 3.
The radar device 100 according to the third embodiment includes a radar signal processing circuit 160 (160C) based on the configuration of the radar device 100 according to the second embodiment and having a certain function added. The radar device 100 according to the third embodiment has a configuration particularly effective when the radar device 100 adopts a staggered PRI method in which PRIs are made unequal intervals for each pulse. The term "staga" is derived from the English word Staged. Staged originally means "staggering" or "staggered", but in this art it means unequal intervals.
Unless otherwise specified, the third embodiment uses the same reference numerals as those of the above-described embodiments, and duplicate explanations are appropriately omitted.

FIG. 15 is a block diagram showing a functional configuration of the radar signal processing circuit 160 (160C) of the radar device 100 according to the third embodiment. As shown in FIG. 15, the radar signal processing circuit 160 (160C) includes a CZT pulse Doppler processing unit 165, a target candidate detection unit 163, a speed measurement / distance measurement unit 164, and an averaging unit 166.

The staga PRI method (also simply referred to as "staga method") is an effective means for solving the ambiguity of Doppler and range. In the stagger method, the pulse repetition cycle changes for each hit number. Therefore, in the third embodiment, the notation of T _PRI (0), T _PRI (1), ..., T _PRI (h), ..., T _PRI (H-1) is used for the pulse repetition period.

The PRI control unit 150 according to the third embodiment has a pulse width of T ₀ and a pulse repetition period of T _PRI (0), T _PRI (1), ..., T _PRI (h), ..., T _PRI ( A timing signal for generating a series of signals according to H-1) is generated. The generated timing signal is sent to the signal generation circuit 110.
The series of signals adopted by the radar device 100 according to the third embodiment may be created based on the idea of a reference period. That is, the series of signals may be, for example, one in which the period is linearly increased or decreased starting from the reference period.

FIG. 16 is a flowchart showing a processing step of the radar signal processing circuit 160 (160C) of the radar device 100 according to the third embodiment. As shown in FIG. 16, the processing flow of the radar signal processing circuit 160 (160C) includes a double processing loop including an inner processing loop related to the speed candidate and an outer processing loop related to the pulse repetition cycle. The inner processing loop related to the speed candidate is the same as the processing loop related to the speed candidate in the flowchart shown in FIG.

As described above, in the radar signal processing circuit 160 (160C) of the radar device 100 according to the third embodiment, the pulse repetition period is T _PRI (0), T _PRI (1), ..., T _PRI (h), ..., T. In each of the _PRI (H-1), the estimated distance and velocity of the target are calculated. The purpose of the radar signal processing circuit 160 (160C) to include the averaging unit 166 is to calculate a more probable estimated value by averaging the estimated values calculated in each of the pulse repetition cycles. The averaging unit 166 calculates the average value of the estimated values calculated in each of the pulse repetition cycles (step shown by ST166 in FIG. 16).

FIG. 17 is an image diagram shown in FIG. 8 applied to the third embodiment. Further, FIG. 18 is an image diagram shown in FIG. 9 applied to the third embodiment.

As described above, since the radar device 100 according to the third embodiment has the above configuration, the staga PRI method can be adopted in addition to the effects shown in the second embodiment, and the estimated values calculated in each of the pulse repetition cycles can be adopted. By taking the average of, it is possible to calculate a more probable estimated value.

The hardware of the radar device 100 according to the third embodiment may be realized with the same configuration as that of the first embodiment.

Embodiment 4.
The radar device 100 according to the fourth embodiment is a radar signal processing circuit 160 based on the configuration of the radar device 100 according to the first embodiment, in which the correlation processing unit 161 and the compensation processing unit 162 are replaced with the inter-pulse stagger integration unit 167. (160D) is provided. The radar device 100 according to the fourth embodiment has a configuration that is particularly effective when the radar device 100 adopts the staggered PRI method. Unless otherwise specified, the fourth embodiment uses the same reference numerals as those of the above-described embodiments, and duplicate explanations are appropriately omitted.

The staggered method assumed in the fourth embodiment is a special one in which several PRI pulses at the same interval are continuously formed to form a group. This is a staggered method in which there are multiple groups and different groups have different PRIs. For example, it is assumed that PRIs at the same interval form a continuous group for 4 pulses. In this case, T _PRI (0) to T _PRI (3) form the first group and have the same value. T _PRI (4) to T _PRI (7) constitute the second group, which are different from the first group and have different common values. Hereinafter, the groups are similarly divided, and in the case of this example, a group of the number obtained by dividing H by 4 is assumed in one series. After the irradiation of the signal of the first series is performed, the hit number returns to 0 and the signal of the second series is irradiated again. Also in the second series, T _PRI (0) to T _PRI (3) form the first group, and T _PRI (4) to T _PRI (7) form the second group. The radar device 100 according to the fourth embodiment repeats the irradiation of the pulse in this way.

The PRI control unit 150 according to the fourth embodiment generates a timing signal for generating a staggered signal forming the above group. The generated timing signal is sent to the signal generation circuit 110.

FIG. 20 is a flowchart showing a processing step of the radar signal processing circuit 160 (160D) of the radar device 100 according to the fourth embodiment. As shown in FIG. 20, the processing steps of the radar signal processing circuit 160 (160D) are the inter-pulse stagger integration step (ST167) performed by the inter-pulse stagger integration unit 167 and the target candidate detection step (ST167) performed by the target candidate detection unit 163. ST163) and the speed measurement / distance measurement step (ST164) performed by the speed measurement / distance measurement unit 164 are included.

It is conceivable to improve the signal-to-noise ratio by adding and integrating the reflected signals of PRIs having the same interval one after another. The purpose of the radar signal processing circuit 160 (160D) according to the fourth embodiment to include the inter-pulse stagger integrating unit 167 is to exert this effect.

FIG. 21 is a reference diagram showing an image of integration performed by the inter-pulse stagger integration unit 167 according to the fourth embodiment. More specifically, FIG. 21 shows the PRI for each hit by describing the number of the group to which the PRI belongs. The right direction in FIG. 21 is the direction in which the hit number increases. FIG. 21 illustrates a case where PRIs at the same interval form a continuous group for 4 pulses. Further, FIG. 21 illustrates a case where H is 16 and the number of groups is 4. The downward direction in FIG. 21 indicates the direction in which the number of series increases.

The inter-pulse stagger integration unit 167 according to the fourth embodiment performs coherent integration with respect to a plurality of series of reflected signals shown in the reference diagram of FIG. 21, for example. The inter-pulse stagger integration unit 167 first performs coherent integration on reflected signals of PRIs having the same interval, and then performs coherent integration on reflected signals of PRIs having different intervals (step shown in ST167 in FIG. 19). Here, the coherent integral performed by the inter-pulse stagger integral unit 167 may be the CZT described above, or may be a discrete Fourier transform of another style. The coherent integral may also utilize a projection matrix or apply a filter to enhance its effect.

Coherent integration for reflection signals of PRIs with the same spacing may be performed for reflection signals belonging to the same group in the same series (see “Integration Direction 1” in FIG. 21), or reflections of the same group number in different series. This may be done for the signal (see “Integral Direction 2” in FIG. 21).

The radar device 100 according to the fourth embodiment performs coherent integration on signals for each pulse included in the pulse repetition cycle of the same type, and then performs coherent integration in different types of pulse repetition cycles. To detect the target.

The radar device 100 according to the fourth embodiment has a configuration including an inter-pulse stagger integration unit 167, and improves the SN ratio by integration after detection.
It is considered that the improvement of the signal-to-noise ratio can also be achieved by the pre-detection integration. In the radar device 100 according to the present disclosure technique, a filter such as a Matched Filter may be used in order to improve the SN ratio.

As described above, since the radar device 100 according to the fourth embodiment has the above configuration, in addition to the effect shown in the second embodiment, the reflected signals of the PRI having the same interval are integrated after detection, so that the SN ratio is used. It has the effect of improving. The post-detection integral is expected to have the same effect as the averaging shown in the third embodiment. Therefore, it can be said that the radar device 100 according to the fourth embodiment has the effect of being able to calculate a more probable estimated value.

The radar device 100 according to the present disclosure technique can be used as a measuring device for measuring a target distance and speed, and has industrial applicability.

100 radar device, 110 signal generation circuit, 111 local oscillator, 112 pulse generator, 113 in-pulse modulator, 114 output unit, 120 transmission / reception unit, 130 antenna, 140 reception circuit, 150 PRI control unit, 160, (160B), (160C), (160D) Radar signal processing circuit, 161 correlation processing unit, 162 compensation processing unit, 163 target candidate detection unit, 164 speed measurement / distance measurement unit, 165 CZT pulse Doppler processing unit, 166 averaging unit, 167 pulse Inter-stagger integrator, 170 signal processing circuit, 171 processor, 172 memory, 173 storage device, 174 input / output interface, 175 signal path, 200 display.

Claims (13)

A PRI control unit that sets a predetermined pulse repetition cycle for each hit,
A signal generation circuit that continuously generates a plurality of transmission pulse signals at a timing based on the pulse repetition cycle, and a signal generation circuit.
A transmission / reception unit that sends the transmission pulse signal to the external space and receives a plurality of reflected wave signals corresponding to each of the transmission pulse signals from the external space.
A receiving circuit that generates a plurality of received signals corresponding to the transmitted pulse signals by sampling each of the reflected wave signals received by the transmitting / receiving unit.
A correlation processing unit that pulse-compresses the received signal by correlation calculation,
A compensation processing unit that performs motion compensation with motion compensation candidates at a preset speed,
A target candidate detection unit that detects a target candidate based on the processed signal of the compensation processing unit, and a target candidate detection unit.
Velocity measurement that estimates the target speed and distance from the results from the target candidate detection unit processed using the plurality of motion compensation candidates at intervals that are two or more integral multiples of the speed interval without ambiguity. A radar device equipped with a distance measuring unit. The radar device according to claim 1, wherein the interval between the motion compensation candidates is calculated based on the integration loss of the target in the target candidate detection unit. The radar device according to claim 1, wherein the signal generated by the motion compensation candidate is incoherently integrated. The target candidate detection unit is
The radar device according to claim 1, wherein the relative distance and the relative speed with respect to the target candidate are calculated. The radar device according to claim 1, wherein the number of speed candidates required to perform the motion compensation and the step size of the speed candidates can be set. Based on the respective detection intensities of the speed candidates, the speed candidates are divided into an increasing first half group and a decreasing second half group.
Obtain an approximate straight line or an approximate curve based on the least squares method in each of the first half group and the second half group.
The radar device according to claim 1, wherein the target speed is obtained as an intersection where each of the approximate straight line or the approximate curve intersects. The radar device according to claim 1, wherein the maximum intensity and the second maximum intensity are selected from the detection results of the speed compensation candidates, and the target speed is estimated. The radar device according to claim 6 , wherein the speed compensation candidate number when the difference value before and after is the smallest is selected from the detection result of the speed compensation candidate. The radar device according to claim 6 , wherein the target speed is estimated based on the sum and difference of two signal intensities selected from the result of processing the speed compensation candidate. The radar device according to claim 1, further comprising an averaging unit for calculating an average value of estimated values calculated for each type of pulse repetition cycle. Claim 1 for detecting the target by performing coherent integration on signals for each pulse included in the pulse repetition cycle of the same type, and then performing coherent integration in different types of pulse repetition cycles. The radar device described. The radar device according to claim 1, wherein the compensation processing unit performs phase compensation using the chirp z-transform, and performs the motion compensation with the motion compensation candidate at a preset speed. The radar device according to claim 12 , wherein the fast time direction of the received signal is converted into a frequency domain, and the chirp z-transform is used.

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