JP7089377B2 - Power converter - Google Patents

Description

The present invention relates to a power conversion device, and more particularly to control of a switching circuit included in the power conversion device.

Electric vehicles such as hybrid vehicles and electric vehicles are widely used. The electric vehicle is equipped with a battery for supplying electric power to the drive motor. In a hybrid vehicle, the battery is charged by the driving force of the engine and the electric power generated by regenerative braking. Further, in an electric vehicle having a plug-in function, the battery is charged by the electric power supplied from the commercial power source. To charge the battery, the electric vehicle is equipped with a power converter. The power conversion device converts the voltage input for charging the battery into an appropriate voltage and applies it to the battery.

The following Patent Documents 1 and 2 describe a power conversion device related to the present invention. Patent Document 1 discloses a power conversion device in which two switching circuits are magnetically coupled by windings connected to each circuit, and electric power is transmitted between the two switching circuits. By magnetically coupling the two switching circuits, the insulation between the two switching circuits is enhanced. Patent Document 2 describes a power conversion device that controls an output voltage by adjusting the frequencies of the first and second boost converters while improving the power factor by pulse width modulation of the first and second boost converters. It is shown. For an electric vehicle having a plug-in function, a technique for adjusting the power factor of input power by using a boost converter (switching circuit) as described in Patent Document 2 has been proposed.

Japanese Unexamined Patent Publication No. 2011-193713 Japanese Unexamined Patent Publication No. 2010-183726

In recent years, a technology called V2H (Vehicle to Home) that supplies AC power from a battery mounted on an electric vehicle to electric equipment in a general household, and a battery mounted on an electric vehicle are used to AC power to a power supply system such as a commercial power supply system. Research is being conducted on a technology called V2G (Vehicle to Grid) that supplies power. In V2H and V2G, it is necessary to have sufficient insulation between the place operated by the user and the battery. Therefore, a power conversion device has been proposed in which two switching circuits are magnetically coupled by a transformer (a winding connected to each switching circuit) to transmit electric power between the two switching circuits. However, in such a power conversion device, it cannot be said that a technique for improving the power transmission characteristics such as the power factor of the input power and the transmission loss has been sufficiently studied.

An object of the present invention is to insulate a DC power supply from an electric device to which an AC power is supplied and to improve the power transmission characteristics of a power conversion device that converts DC power into AC power.

The present invention is a power conversion device , wherein a first switching circuit and a second switching circuit, a primary winding having both ends connected to the first switching circuit, and both ends connected to the second switching circuit are connected. The first switching circuit comprises a secondary winding that is magnetically coupled to the primary winding, and the first switching circuit is a full bridge in which two half bridges are connected in parallel, and each half bridge has its own. The primary winding comprises a full bridge comprising two switching elements, one ends of which are commonly connected, and a first and second rectifying switching elements of which one end is commonly connected. It is connected between the connection point of the two switching elements in one of the two half bridges and the connection point of the two switching elements in the other of the two half bridges, and is the first rectification switching. The other end of the element is connected to one of the two parallel connection points in the full bridge, and the other end of the second rectifying switching element is connected to the other of the two parallel connection points. Based on the DC power input to the second switching circuit from between the path drawn from the middle point of the primary winding and the connection point of the first rectifying switching element and the second rectifying switching element. The output AC voltage is output, and the power converter further switches the first switching circuit according to the difference between the direct current flowing into the second switching circuit and the target value with respect to the direct current. A control unit that controls the difference between the phase and the phase for switching the second switching circuit, and a capacitor connected to two parallel connection points in the full bridge, the control unit is located between the terminals of the capacitor. The target value for the voltage, the target value for the direct current flowing into the second switching circuit, the duty ratio when switching the first switching circuit and the second switching circuit, and the leakage inductance in the primary winding. It is characterized in that the difference between the phase for switching the first switching circuit and the phase for switching the second switching circuit is controlled based on the switching frequency .

Further, the present invention is a power conversion device, in which a first switching circuit and a second switching circuit, a primary winding having both ends connected to the first switching circuit, and both ends connected to the second switching circuit. The first switching circuit is a full bridge in which two half bridges are connected in parallel, and each half bridge is provided with a secondary winding that is magnetically coupled to the primary winding. The primary winding comprises a full bridge comprising two switching elements, one end of which is commonly connected, and a first rectifying switching element and a second rectifying switching element, one end of which is commonly connected. Is connected between the connection point of the two switching elements in one of the two half bridges and the connection point of the two switching elements in the other of the two half bridges. The other end of the rectifying switching element is connected to one of the two parallel connection points in the full bridge, and the other end of the second rectifying switching element is connected to the other of the two parallel connection points. The DC power input to the second switching circuit from between the path drawn from the middle point of the primary winding and the connection point of the first rectifying switching element and the second rectifying switching element. The output AC voltage is output based on the above, and the power conversion device further obtains a target value for a capacitor connected to two parallel connection points in the full bridge, a voltage between terminals of the capacitor, and a voltage between terminals of the capacitor. The first switching circuit and the control unit for switching the second switching circuit are provided at a duty ratio corresponding to the difference between the control unit and the output AC voltage and the intermediate point of the primary winding. It is characterized in that the first switching circuit and the second switching circuit are switched at a duty ratio corresponding to a midpoint current flowing in a path drawn from the circuit.

Desirably, the control unit switches the first switching circuit and the second switching circuit with a duty ratio corresponding to the difference between the target value for the output AC voltage and the voltage between the terminals of the capacitor.

Further, the present invention is a power conversion device, in which a first switching circuit and a second switching circuit, a primary winding having both ends connected to the first switching circuit, and both ends connected to the second switching circuit. The first switching circuit is a full bridge in which two half bridges are connected in parallel, and each half bridge is provided with a secondary winding that is magnetically coupled to the primary winding. The primary winding comprises a full bridge comprising two switching elements, one end of which is commonly connected, and a first rectifying switching element and a second rectifying switching element, one end of which is commonly connected. Is connected between the connection point of the two switching elements in one of the two half bridges and the connection point of the two switching elements in the other of the two half bridges. The other end of the rectifying switching element is connected to one of the two parallel connection points in the full bridge, and the other end of the second rectifying switching element is connected to the other of the two parallel connection points. The DC power input to the second switching circuit from between the path drawn from the middle point of the primary winding and the connection point of the first rectifying switching element and the second rectifying switching element. The output AC voltage is output based on the above, and the power conversion device further performs the first switching circuit and the second switching at a duty ratio according to the difference between the output AC voltage and the target value with respect to the output AC voltage. It is characterized by including a control unit for switching a circuit.

Desirably, a capacitor connected to two parallel connection points in the full bridge is provided, and the control unit has a duty ratio according to a difference between a target value for the output AC voltage and a voltage between terminals of the capacitor. The first switching circuit and the second switching circuit are switched.

Desirably , the first switching circuit is switched according to the difference between the capacitor connected to the two parallel connection points in the full bridge, the voltage between the terminals of the capacitor, and the target value with respect to the voltage between the terminals of the capacitor. A control unit for controlling the difference between the phase to be switched and the phase for switching the second switching circuit is provided.

Desirably, the second switching circuit is a secondary full bridge in which two secondary half bridges are connected in parallel, and each of the secondary half bridges has one end connected in common. It comprises a secondary full bridge with one switching element, the secondary winding being the connection point of the two switching elements in one of the two secondary half bridges and the two secondary halves. It is connected between the connection points of the two switching elements on the other side of the bridge, and DC power is input from the two parallel connection points on the secondary side full bridge.

Desirably, each of the first rectifying switching element and the second rectifying switching element is turned on and off according to the polarity of the output AC voltage.

According to the present invention, it is possible to insulate the electric device to which the AC power is supplied from the DC power supply and improve the power transmission characteristics.

It is a figure which shows the structure of the power conversion device for vehicle mounting. It is a figure which shows the operation timing of the power factor improvement circuit in the half cycle when the input AC voltage becomes a positive value. It is a figure which shows the operation timing of the power factor improvement circuit in a half cycle in which an input AC voltage becomes a negative value. It is a figure which shows the time waveform of the primary winding voltage, the secondary winding voltage, and the secondary winding current. It is a figure which shows the connection configuration example in a power supply mode. It is a figure which shows the time waveform of the primary winding voltage, the secondary winding voltage, and the primary winding current. It is a figure which shows the equivalent circuit of a power supply system. It is a figure which shows the structural example of the control part in the input current control mode. It is a figure which shows the simulation result about the operation of the input current control mode. It is a figure which shows the structural example of the control part in the output voltage control mode. It is a figure which shows the simulation result about the operation of the output voltage control mode.

(1) Outline of Configuration of Power Conversion Device FIG. 1 shows a configuration of a power conversion device for mounting on a vehicle according to the basic technique of the present invention. The power conversion device includes a power factor improving circuit 10, a voltage converter circuit 14, and a control unit 22. An AC voltage source 18 is connected to the power factor improving circuit 10. The AC voltage source 18 is, for example, a commercial power source, and when the vehicle on which the power conversion device is mounted has a plug-in function, the power supply connector becomes the AC voltage source 18. A DC power supply circuit 20 is connected to the voltage converter circuit 14. The DC power supply circuit 20 is, for example, a battery or a charging circuit for charging the battery. The control unit 22 controls on / off of each switching element included in the power factor improving circuit 10 and the voltage converter circuit 14.

The power factor improving circuit 10 adjusts the time waveform of the current flowing in from the AC voltage source 18 by switching, and improves the power factor when the power conversion device side is viewed from the AC voltage source 18. The power factor improving circuit 10 and the voltage converter circuit 14 are coupled by a transformer T, and the electric power output from the AC voltage source 18 is transmitted from the power factor improving circuit 10 to the voltage converter circuit 14. The voltage converter circuit 14 converts the AC voltage obtained from the secondary winding T2 of the transformer T into a DC voltage, and outputs a DC voltage of an appropriate magnitude to the DC power supply circuit 20. According to the power factor improving circuit 10 and the voltage converter circuit 14, power is efficiently supplied from the AC voltage source 18 to the DC power supply circuit 20. When the DC power supply circuit 20 is a battery, the battery as the DC power supply circuit 20 is charged, and when the DC power supply circuit 20 is a charging circuit for charging the battery, the battery is charged by the DC power supply circuit 20. To.

(2) Configuration of Power Factor Improvement Circuit and Operation The configuration of the power factor improvement circuit 10 will be described. The power factor improving circuit 10 includes a filter capacitor Cf, a primary winding T1, and a first switching circuit 12.

The first switching circuit 12 includes a half bridge U composed of switching elements S1 and S2, a half bridge V composed of switching elements S3 and S4, a rectifying switching element S9, a rectifying switching element S10, and an intermediate capacitor Cbuf. There is. The half bridge U connects one end of the switching element S1 and one end of the switching element S2. Diodes are connected to both ends of the switching element S1 with the side of the connection point with the switching element S2 as the anode. Diodes are connected to both ends of the switching element S2 with the side of the connection point with the switching element S1 as the cathode. As the switching elements S1 and S2, for example, an IGBT (Insulated Gate Bipolar Transistor) is used. In this case, the emitter of the IGBT as the switching element S1 and the collector of the IGBT as the switching element S2 are connected.

Similarly, the half bridge V connects one end of the switching element S3 and one end of the switching element S4. Diodes are connected to both ends of the switching element S3 with the side of the connection point with the switching element S4 as an anode. Diodes are connected to both ends of the switching element S4 with the side of the connection point with the switching element S3 as the cathode. As the switching elements S3 and S4, for example, an IGBT is used. In this case, the emitter of the IGBT as the switching element S3 and the collector of the IGBT as the switching element S4 are connected.

A primary winding T1 is connected between the connection points of the switching elements S1 and S2 and the connection points of the switching elements S3 and S4. The tap m (midway point) of the primary winding T1 is connected to the AC terminal 24-2.

The half bridges U and V are connected in parallel to form a full bridge. That is, the terminal on the side opposite to the switching element S2 side of the switching element S1 (the terminal on the upper side in the figure) and the terminal on the side opposite to the switching element S4 side in the switching element S3 (the terminal on the upper side in the figure) are connected. ing. Further, the terminal on the side opposite to the switching element S1 side of the switching element S2 (the terminal on the lower side in the figure) and the terminal on the side opposite to the switching element S3 side of the switching element S4 (the terminal on the lower side in the figure) are It is connected.

One end of the rectifying switching element S9 is connected to one end of the rectifying switching element S10. The other end of the rectifying switching element S9 is connected to the upper terminals of the half bridges U and V, and the other end of the rectifying switching element S10 is connected to the lower terminals of the half bridges U and V. The connection points of the rectifying switching elements S9 and S10 are connected to the AC terminal 24-1. Diodes are connected to both ends of the rectifying switching element S9 with the side of the connection point with the rectifying switching element S10 as an anode. Diodes are connected to both ends of the rectifying switching element S10 with the side of the connection point with the rectifying switching element S9 as the cathode. As the rectifying switching elements S9 and S10, for example, an IGBT is used. In this case, the emitter of the IGBT as the rectifying switching element S9 and the collector of the IGBT as the rectifying switching element S10 are connected.

An intermediate capacitor Cbuf is connected between the connection point of the switching element S1, the switching element S3, and the rectifying switching element S9 and the connection point of the switching element S2, the switching element S4, and the rectifying switching element S10.

The primary winding T1 is magnetically coupled to the secondary winding T2 included in the voltage converter circuit 14, and the primary winding T1 and the secondary winding T2 constitute a transformer T. The primary winding T1 is represented by connecting a first inductor L1 as a self-inductor, a second inductor L2 as a self-inductor, and a mutual inductor M in series. The first inductor L1 is connected between one end of the mutual inductor M and the connection points of the switching elements S1 and S2, and the second inductor is connected between the other end of the mutual inductor M and the connection points of the switching elements S3 and S4. L2 is connected. The coupling coefficient k of the transformer T has a relationship of k = M / (L1 + L2). The first inductor L1 and the second inductor L2 may be magnetically coupled. Further, an inductor (reactor) may be connected between the AC terminal 24-2 and the tap m.

A filter capacitor Cf is connected between the AC terminal 24-1 and the AC terminal 24-2. Further, an AC voltage source 18 is connected between the AC terminal 24-1 and the AC terminal 24-2. When the AC voltage source 18 is a commercial power source, a power plug is connected to the AC terminals 24-1 and 24-2 via a cable, and the power plug is inserted into the power connector.

The operation of the power factor improving circuit 10 will be described. The AC voltage source 18 outputs an input AC voltage Vac, which is a sinusoidal voltage, to the AC terminals 24-1 and 24-2. The filter capacitor Cf is generated in the power factor improving circuit 10 and suppresses the high frequency current flowing out to the AC voltage source 18 side.

The control unit 22 outputs the control signals Cn1 to Cn4 to the switching elements S1 to S4, respectively, and controls the switching elements S1 to S4 on and off. When the control signal Cni is high, the switching element Si is turned on, and when the control signal Cni is low, the switching element Si is turned off. However, i is an integer of 1 to 4. The control signal Cn2 is the inverted high and low with respect to the control signal Cn1, and the control signal Cn4 is the inverted high and low with respect to the control signal Cn3. Further, the phases of the control signals Cn3 and Cn4 are 180 ° behind the control signals Cn1 and Cn2, respectively.

As a result, the switching element S1 and the switching element S2 are alternately turned on and off. That is, when the switching element S1 is turned from off to on, the switching element S2 is changed from on to off, and when the switching element S1 is turned from on to off, the switching element S2 is changed from off to on. Similarly, the switching element S3 and the switching element S4 are alternately turned on and off. The on / off phase of the switching elements S3 and S4 is delayed by 180 ° with respect to the on / off phase of the switching elements S1 and S2.

The control unit 22 has, for example, an intermediate voltage (intermediate voltage Vb) between terminals of the intermediate capacitor Cbuf, an intermediate point current iL flowing through a path between the AC voltage source 18 and the tap m of the primary winding T1, and an AC voltage source 18. The duty ratio (time ratio) of the control signals Cn1 to Cn4 is changed according to the input AC voltage Vac or the like to be output. Further, when the polarity of the input AC voltage Vac is positive on the side of the AC terminal 24-1 (during a half cycle in which the input AC voltage Vac becomes a positive value), the control unit 22 sets the rectifying switching element S9. Turn off and turn on the rectifying switching element S10. Then, when the polarity of the input AC voltage Vac is negative on the side of the AC terminal 24-1 (during a half cycle in which the input AC voltage Vac becomes a negative value), the control unit 22 sets the rectifying switching element S9. Turn on and turn off the rectifying switching element S10. As a result, the time waveform of the current flowing through the AC terminals 24-1 and 24-2 is approximated or matched with the time waveform of the input AC voltage Vac, and the phase of the current flowing through the AC terminals 24-1 and 24-2 is adjusted. Approximately or match the phase of the input AC voltage Vac.

FIGS. 2A to 2E show the operation timing of the power factor improving circuit 10 in a half cycle in which the input AC voltage Vac becomes a positive value. FIG. 2A shows the inverted value of the control signal Cn1 and the time waveform of the control signal Cn2, and FIG. 2B shows the inverted value of the control signal Cn3 and the time waveform of the control signal Cn4. There is. The "-" symbol attached above the code means the inverted value of the control signal represented by the code. FIG. 2C shows a time waveform of the potential Vu (U-phase potential Vu) at the connection point of the switching elements S1 and S2, and FIG. 2D shows the connection with the switching elements S3 and S4. The time waveform of the point potential Vv (V-phase potential Vv) is shown. Further, FIG. 2E shows a time waveform of the voltage Vuv (primary winding voltage Vuv) applied to the primary winding T1. The inverted value of the control signal Cn1 and the inverted value of the control signal Cn3 are the same as those of the control signals Cn2 and Cn4, respectively. Further, the reference of the U-phase potential Vu and the V-phase potential Vv is the potential of the ground conductor G1.

As shown in FIG. 2A, the inversion value of the control signal Cn1 and the period of the control signal Cn2 are P, and the inversion value of the control signal Cn1 and the control signal Cn2 are in high time during one period P. Only δ becomes high. As shown in FIG. 2B, the inverted value of the control signal Cn3 and the control signal Cn4 are delayed by half a cycle, that is, 180 ° with respect to the inverted value of the control signal Cn1 and the control signal Cn2. The duty ratio α is α = δ / P, the high time δ changes according to the instantaneous value of the input AC voltage Vac, and the duty ratio α changes with the change of the high time δ.

Here, the operation of the power factor improving circuit 10 will be described assuming that the intermediate capacitor Cbuf is charged to a constant voltage Vb.

While the control signal Cn1 is high and the control signal Cn2 is low, the switching element S1 is turned on and the switching element S2 is turned off. As a result, the U-phase potential Vu becomes the charging voltage (intermediate voltage Vb) of the intermediate capacitor Cbuf. On the other hand, while the control signal Cn1 is low and the control signal Cn2 is high, the switching element S1 is turned off and the switching element S2 is turned on. As a result, the U-phase potential becomes 0. Therefore, as shown in FIG. 2 (c), the U-phase potential Vu becomes Vb during the time (P-δ) in the period P and becomes 0 in the other time zones.

While the control signal Cn3 is high and the control signal Cn4 is low, the switching element S3 is turned on and the switching element S4 is turned off. As a result, the V-phase potential Vv becomes an intermediate voltage Vb. On the other hand, while the control signal Cn3 is low and the control signal Cn4 is high, the switching element S3 is turned off and the switching element S4 is turned on. As a result, the V-phase potential becomes 0. Therefore, as shown in FIG. 2D, the V-phase potential Vv has the same time waveform as the U-phase potential Vu, and is 180 ° out of phase with the U-phase potential Vu.

The primary winding voltage Vuv is a voltage obtained by subtracting the V-phase potential Vv from the U-phase potential Vu. As a result, as shown in FIG. 2 (e), the primary winding voltage Vuv becomes a positive-negatively symmetric time waveform at the peak value Vb.

FIGS. 3A to 3E show the operation timing of the power factor improving circuit 10 in a half cycle in which the input AC voltage Vac becomes a negative value. FIG. 3A shows the inverted value of the control signal Cn1 and the time waveform of the control signal Cn2, and FIG. 3B shows the inverted value of the control signal Cn3 and the time waveform of the control signal Cn4. There is. FIG. 3C shows a time waveform of the U-phase potential Vu, and FIG. 3D shows a time waveform of the V-phase potential Vv. Further, FIG. 3 (e) shows a time waveform of the primary winding voltage Vuv. The same matters as those shown in FIG. 2 are designated by the same reference numerals and the description thereof will be omitted.

As shown in FIG. 3A, the inversion value of the control signal Cn1 and the cycle of the control signal Cn2 are P, and the inversion value of the control signal Cn1 and the control signal Cn2 are in high time during one cycle P. Only γ becomes high. As shown in FIG. 3B, the inverted value of the control signal Cn3 and the control signal Cn4 are delayed by half a cycle, that is, 180 ° with respect to the inverted value of the control signal Cn1 and the control signal Cn2. The duty ratio α is α = γ / P, the high time γ changes according to the instantaneous value of the input AC voltage Vac, and the duty ratio α changes with the change of the high time γ.

As shown in FIG. 3C, in the half cycle in which the input AC voltage Vac becomes a negative value by the same operation as the half cycle in which the input AC voltage Vac becomes a positive value, the U phase potential Vu Is Vb during the time (P-γ) in the period P, and becomes 0 in the other time zones. Further, as shown in FIG. 3D, the V-phase potential Vv has the same time waveform as the U-phase potential Vu, and is a potential whose phase is delayed by 180 ° from the U-phase potential Vu. The primary winding voltage Vuv is a voltage obtained by subtracting the V-phase potential Vv from the U-phase potential Vu. As a result, as shown in FIG. 3 (e), the primary winding voltage Vuv becomes a positive-negatively symmetric time waveform at the peak value Vb. As a result, the secondary winding voltage Vwx based on the primary winding voltage Vuv is generated in the secondary winding T2.

Further, by controlling the switching S1 to S4 on and off according to the control signals Cn1 to Cn4, the switching elements S1 to S4, the rectifying switching elements S9 and S10 operate as a rectifier circuit, and the voltage Vuv between the terminals of the primary winding T1 is increased. It is rectified and applied to the intermediate capacitor Cbuf. As a result, the intermediate capacitor Cbuf is charged based on the input AC voltage Vac.

The period of the control signals Cn1 to Cn4 is sufficiently shorter than the period of the input AC voltage Vac. The time waveform of the current flowing through the primary winding T1 is shaped by the switching of the switching elements S1 to S4, and the power factor improving operation is executed.

(3) Configuration and Operation of Voltage Converter Next, the configuration of the voltage converter circuit 14 will be described with reference to FIG. The voltage converter circuit 14 includes a secondary winding T2 and a second switching circuit 16.

The second switching circuit 16 includes a secondary side half bridge W composed of switching elements S5 and S6, a secondary side half bridge X composed of switching elements S7 and S8, and a secondary side capacitor Cout. The secondary side half bridge W connects one end of the switching element S5 and one end of the switching element S6. Diodes are connected to both ends of the switching element S5 with the side of the connection point with the switching element S6 as an anode. Diodes are connected to both ends of the switching element S6 with the side of the connection point with the switching element S5 as the cathode. As the switching elements S5 and S6, for example, an IGBT is used. In this case, the emitter of the IGBT as the switching element S5 and the collector of the IGBT as the switching element S6 are connected.

Similarly, the half bridge X connects one end of the switching element S7 and one end of the switching element S8. Diodes are connected to both ends of the switching element S7 with the side of the connection point with the switching element S8 as an anode. Diodes are connected to both ends of the switching element S8 with the side of the connection point with the switching element S7 as the cathode. As the switching elements S7 and S8, for example, an IGBT is used. In this case, the emitter of the IGBT as the switching element S7 and the collector of the IGBT as the switching element S8 are connected.

A secondary winding T2 is connected between the connection points of the switching elements S5 and S6 and the connection points of the switching elements S7 and S8.

The secondary side half bridges W and X are connected in parallel to form a secondary side full bridge. That is, the upper terminal of the switching element S5 and the upper terminal of the switching element S6 are connected, and the lower terminal of the switching element S7 and the lower terminal of the switching element S8 are connected. A secondary capacitor Cout is connected between the upper terminals of the secondary half bridges W and X and the lower terminals of the secondary half bridges W and X. Further, the positive electrode terminal 26P is connected to the upper terminal of the secondary side half bridge W and X, and the negative electrode terminal 26N is connected to the lower terminal of the secondary side half bridge W and X. Further, a DC power supply circuit 20 is connected between the positive electrode terminal 26P and the negative electrode terminal 26N.

The operation of the voltage converter circuit 14 will be described. A voltage is generated in the secondary winding T2 according to the voltage applied to the primary winding T1 of the power factor improving circuit 10, and the voltage generated in the secondary winding T2 is connected to the connection points of the switching elements S5 and S6. It is applied between the connection points of the switching elements S7 and S8.

The control unit 22 outputs the control signals Cn5 to Cn8 to the switching elements S5 to S8, respectively, and controls the switching elements S5 to S8 on and off. When the control signal Cni is high, the switching element Si is turned on, and when the control signal Cni is low, the switching element Si is turned off. However, i is an integer of 5 to 8. The control signal Cn6 has high and low inverted with respect to the control signal Cn5, and the control signal Cn8 has high and low inverted with respect to the control signal Cn7. Further, the phases of the control signals Cn7 and Cn8 are 180 ° behind the control signals Cn5 and Cn6, respectively.

As a result, the switching elements S5 and S6 are alternately turned on and off. That is, when the switching element S5 goes from off to on, the switching element S6 goes from on to off, and when the switching element S5 goes from on to off, the switching element S6 goes from off to on. Similarly, the switching elements S7 and S8 are alternately turned on and off. The on / off phase of the switching elements S7 and S8 is delayed by 180 ° with respect to the on / off phase of the switching elements S5 and S6. The control unit 22 matches the duty ratio in the voltage converter circuit 14 with the duty ratio in the power factor improving circuit 10.

The control unit 22 delays the phase of switching the second switching circuit 16 with respect to the first switching circuit 12 according to the difference between the voltage between the terminals of the secondary capacitor Cout and the target value thereof.

Here, the operation of the voltage converter circuit 14 will be described assuming that the secondary side capacitor Cout is charged to a constant voltage Vd.

FIG. 4A shows the time waveforms of the primary winding voltage Vuv and the secondary winding voltage Vwx. Here, the secondary winding voltage Vwx is the voltage at the connection point of the switching elements S5 and S6 with reference to the potential of the connection point of the switching elements S7 and S8. The primary winding voltage Vuv is a rectangular wave having a peak value of Vb, and the secondary winding voltage Vwx is a rectangular wave having a peak value of Vd. The phase of the secondary winding voltage Vwx is delayed by φ with respect to the primary winding voltage Vuv. FIG. 4B shows a time waveform of the current id flowing through the secondary winding T2. The secondary winding current id is positive in the direction from the secondary side half bridge X to the secondary side half bridge W.

During the period τ1 when the primary winding voltage Vuv rises from 0 to Vb and the secondary winding voltage Vwx is 0, the secondary winding current id rapidly increases from 0 in the positive direction. After that, the change of the secondary winding current id becomes gradual during the period τ2 in which the secondary winding voltage Vwx rises to Vd and the primary winding voltage Vv is Vb and the secondary winding voltage Vwx is Vd. .. Further, the primary winding voltage Vuv drops from Vb to 0, and the secondary winding current id decreases sharply toward 0 during the period τ3 when the secondary winding voltage Vwx is Vd.

During the period τ4 when the primary winding voltage Vuv and the secondary winding voltage Vwx are 0, the secondary winding current id is 0.

During the period τ5 when the primary winding voltage Vuv drops from 0 to −Vb and the secondary winding voltage Vwx is 0, the secondary winding current id rapidly increases from 0 in the negative direction. After that, the secondary winding voltage Vwx drops to -Vd, and the change in the secondary winding current id during the period τ6 when the primary winding voltage Vuv is -Vb and the secondary winding voltage Vwx is -Vd. Becomes gradual. Further, the primary winding voltage Vuv rises from −Vb to 0, and the secondary winding current id decreases sharply toward 0 during the period τ7 when the secondary winding voltage Vwx is −Vd.

During the period τ1 and τ5 from the rise or fall of the primary winding voltage Vuv to the rise or fall of the secondary winding voltage Vwx, energy is supplied from the primary winding T1 to the secondary winding T2. At the same time, the secondary winding T2 stores energy. Then, during the periods τ2, τ3, τ6, and τ7, the voltage converter circuit 14 outputs the power determined by the product of the secondary winding voltage Vwx and the secondary winding current id to the DC power supply circuit 20.

The larger the phase difference φ between the primary winding voltage Vuv and the secondary winding voltage Vwx, the longer the period τ1 and τ5 in which energy is stored in the secondary winding T2, and in the periods τ2, τ3, τ6 and τ7. The absolute value of the secondary winding current id becomes large. However, the phase difference φ is a value less than 180 °. Therefore, the larger the phase difference φ between the primary winding voltage Vuv and the secondary winding voltage Vwx, the more the power factor improvement circuit 10 transmits to the voltage converter circuit 14, and the voltage converter circuit 14 outputs to the DC power supply circuit 20. Power increases.

The control unit 22 switches and controls the switching elements S5 to S8 as described above, so that the secondary winding current id flows from the upper end to the lower end of the secondary side capacitor Cout, and the secondary side capacitor Cout is predetermined. It is charged with the voltage Vd of.

The voltage converter circuit 14 is magnetically coupled to the power factor improving circuit 10 by a transformer T composed of a primary winding T1 and a secondary winding T2. Therefore, the power factor improving circuit 10 is electrically isolated from the voltage converter circuit 14, and it is avoided that the current due to the high voltage generated in the voltage converter circuit 14 flows to the power factor improving circuit 10. Further, as described above, since the primary winding voltage Vuv applied to the primary winding T1 has a time waveform of positive and negative symmetry, electric power is transmitted from the power factor improving circuit 10 to the voltage converter circuit 14. The loss that occurs in the transformer T when it is done is reduced.

(4) Power supply mode (input current control mode and output voltage control mode)
In the above, the operation in which the AC voltage source 18 is connected to the AC terminals 24-1 and 24-2 and the power is supplied from the AC voltage source 18 to the DC power supply circuit 20 has been described. In this operation, when the DC power supply circuit 20 is a battery, the battery as the DC power supply circuit 20 is charged, and when the DC power supply circuit 20 is a charging circuit for charging the battery, the DC power supply circuit 20 is used. The battery is charged. In addition to such a charging mode, the power conversion device operates in a power supply mode in which a load circuit is connected to the AC terminals 24-1 and 24-2 and power is supplied from the DC power supply circuit 20 to the load circuit.

In FIG. 5, as an example of the connection configuration in the power supply mode, the battery 28 is connected to the positive electrode terminal 26P and the negative electrode terminal 26N as the DC power supply circuit 20, and the load circuit 30 is connected to the AC terminals 24-1 and 24-2. An example of the connection configuration is shown.

The power supply mode includes an input current control mode and an output voltage control mode. In the input current control mode, the current flowing from the battery 28 to the positive electrode terminal 26P and the current flowing out from the negative electrode terminal 26N to the battery 28 are brought close to the target value or adjusted to the target value.

The input current control mode may be an operation mode in V2G. In V2G, a power supply system is connected to AC terminals 24-1 and 24-2 as a load circuit 30. Since the transmission voltage is maintained constant in the power supply system, the output AC voltage appearing at the AC terminals 24-1 and 24-2 to which the power supply system is connected is maintained at a predetermined value. Further, the voltage output from the battery 28 to the positive electrode terminal 26P and the negative electrode terminal 26N is constant. Therefore, the current flowing from the battery 28 to the positive electrode terminal 26P and the current flowing out from the negative electrode terminal 26N to the battery 28 are brought close to or matched with the target values, so that the current flowing through the AC terminals 24-1 and 24-2 is predetermined. Approached or matched to the value. As a result, the electric power supplied from the electric power converter to the electric power supply system as the load circuit 30 is brought close to or adjusted to a predetermined value.

On the other hand, the output voltage control mode may be the operation mode in V2H. In V2H, general electric equipment is connected to AC terminals 24-1 and 24-2 as a load circuit. Examples of electrical equipment include home appliances and office equipment operated by a commercial AC power source. In the output voltage control mode, the output AC voltage appearing at the AC terminals 24-1 and 24-2 is brought close to or adjusted to a predetermined value. This stabilizes the power supply voltage applied to the electrical equipment connected to the AC terminals 24-1 and 24-2.

The switching timing of the switching elements S1 to S4 in the power supply mode is the same as the switching timing in the charging mode. The switching timings of the rectifying switching elements S9 and S10 in the power supply mode are as follows. When the output AC voltage between the AC terminals 24-1 and 24-2 is positive on the side of the AC terminal 24-1, the control unit 22 turns off the rectifying switching element S9 and turns on the rectifying switching element S10. To. When the output AC voltage is negative on the side of the AC terminal 24-1, the control unit 22 turns on the rectifying switching element S9 and turns off the rectifying switching element S10.

The switching timing of the switching elements S5 to S8 in the power supply mode is also the same as the switching timing in the charging mode. However, in the charging mode, the control unit 22 delays the phase of switching the switching elements S5 to S8 with respect to the switching elements S1 to S4, whereas in the power supply mode, the control unit 22 causes the switching elements S5 to S8. The switching phase is advanced with respect to the switching elements S1 to S4.

FIG. 6A shows the time waveforms of the primary winding voltage Vuv and the secondary winding voltage Vwx in the power supply mode. The phase of the secondary winding voltage Vwx is advanced by φ with respect to the primary winding voltage Vuv. FIG. 6B shows a time waveform of the current ie flowing through the primary winding T1. The primary winding current ie is positive in the direction from the half bridge V to the half bridge U.

The primary winding current ie rapidly increases from 0 in the positive direction during the period τ1 when the secondary winding voltage Vwx rises from 0 to Vd and the primary winding voltage Vuv is 0. After that, the primary winding voltage Vuv rises to Vb, and the change in the primary winding current ie becomes gradual during the period τ2 in which the secondary winding voltage Vwx is Vd and the primary winding voltage Vv is Vb. .. Further, the secondary winding voltage Vwx drops from Vd to 0, and the primary winding current ie decreases sharply toward 0 during the period τ3 when the primary winding voltage Vuv is Vb.

During the period τ4 where the secondary winding voltage Vwx and the primary winding voltage Vuv are 0, the primary winding current ie is 0.

During the period τ5 when the secondary winding voltage Vwx falls from 0 to −Vd and the primary winding voltage Vuv is 0, the primary winding current ie rapidly increases from 0 in the negative direction. After that, the primary winding voltage Vuv drops to -Vb, and the change in the primary winding current ie during the period τ6 where the secondary winding voltage Vwx is -Vd and the primary winding voltage Vv is -Vb. Becomes gradual. Further, the secondary winding voltage Vwx rises from −Vd to 0, and the primary winding current ie decreases sharply toward 0 during the period τ7 when the primary winding voltage Vuv is −Vb.

During the period τ1 and τ5 from the rise or fall of the secondary winding voltage Vwx to the rise or fall of the primary winding voltage Vuv, energy is supplied from the secondary winding T2 to the primary winding T1. At the same time, the primary winding T1 stores energy. Then, during the periods τ2, τ3, τ6 and τ7, the power factor improving circuit 10 loads the power determined by the product of the primary winding voltage Vuv and the primary winding current ie from the AC terminals 24-1 and 24-2. Output to circuit 30.

The larger the phase difference φ between the secondary winding voltage Vwx and the primary winding voltage Vuv, the longer the period τ1 and τ5 in which energy is stored in the primary winding T1, and in the periods τ2, τ3, τ6 and τ7. The absolute value of the primary winding current ie becomes large. However, the phase difference φ is a value less than 180 °. Therefore, the larger the phase difference φ between the secondary winding voltage Vwx and the primary winding voltage Vuv, the more it is transmitted from the voltage converter circuit 14 to the power factor improving circuit 10 and output from the power factor improving circuit 10 to the power supply system. Power increases.

The control unit 22 switches and controls the switching elements S1 to S4 as described above, so that the primary winding current ie flows from the upper end to the lower end of the intermediate capacitor Cbuf, and the intermediate capacitor Cbuf has a predetermined voltage Vb. It will be charged.

(5) Input Current Control Mode An input current control mode, which is one of the power supply modes, will be specifically described. In the input current control mode, for example, a power supply system is connected to the AC terminals 24-1 and 24-2 as a load circuit 30. FIG. 7 shows an equivalent circuit of the power supply system 32. The power supply system 32 includes two power transmission lines 25, a load circuit 36, and a power transmission / distribution facility 34. The power transmission / distribution equipment 34 supplies power to the load circuit 36 while maintaining a constant voltage between the two power transmission lines 25. The two power transmission lines 25 are connected to the AC terminals 24-1 and 24-2, and power is supplied from the power conversion device to the load circuit 36 via the two power transmission lines 25. As a result, power is supplied from the power conversion device to the load circuit 36 in addition to the power transmission / distribution equipment 34. The output AC voltage (system AC voltage Vs) between the AC terminals 24-1 and 24-2 is maintained constant by the power transmission / distribution equipment 34.

FIG. 8 shows a block diagram of the control unit 22A as a configuration example of the control unit 22 in the input current control mode. The control unit 22A may include a processor that realizes each component shown in FIG. 8 by executing a program. Further, each component may be individually configured by an electronic circuit as hardware.

When the control unit 22A generates each control signal, the measured value Ibtm of the battery current ibt flowing from the battery 28 to the positive electrode terminal 26P and flowing from the negative electrode terminal 26N to the battery 28, the measured value Vbm of the intermediate voltage Vb, and the system AC voltage. The measured value Vsm of Vs and the measured value IL of the midpoint current iL are used. The power conversion device is provided with sensors (not shown) for measuring these.

The control unit 22A has the difference between the intermediate voltage measured value Vbm, which is the measured value of the intermediate voltage Vb, and its target value Vb ^* , the system AC voltage measured value Vsm, the midpoint current measured value IL, the system AC voltage target value Vs ^* , and the system AC voltage target value Vs *. The duty ratio target value α is obtained based on the intermediate voltage measurement value Vbm. Further, the control unit 22A is a second switching circuit 16 based on the difference between the battery current measured value Ibtm and its target value Ibt ^* , the battery current target value Ibt ^* , the intermediate voltage target value Vb ^* , and the duty ratio target value α. Each control signal that changes the phase of switching with respect to the first switching circuit 12 is generated.

The process of obtaining the duty ratio target value α will be described. The subtractor 38 subtracts the intermediate voltage measurement value Vbm from the intermediate voltage target value Vb ^* to obtain the first error, and outputs the first error to the voltage PI control unit 40. The voltage PI control unit 40 obtains a first control value by proportional integration control and outputs it to the multiplier 42. The multiplier 42 subtracts the midpoint current target value iL ^* obtained by multiplying the system AC voltage measured value Vsm by the first control value and further multiplying by the inverse of the square of the time average value of the system AC voltage measured value Vsm. Output to 44. The subtractor 44 subtracts the midpoint current measurement value IL from the midpoint current target value iL ^* to obtain the second error, and outputs the second error to the current PI control unit 46. The current PI control unit 46 obtains a second control value by proportional integration control and outputs it to the adder 48. The adder 48 adds the half-cycle target value 1-Vs ^* / Vbm to the second control value to obtain the pre-adjustment target value α0. The half-cycle target value 1-Vs ^* / Vbm has significance as a duty ratio at which the power factor improving effect can be obtained in the power factor improving circuit 10 when the system AC voltage Vs is a positive half-cycle value.

When the system AC voltage measurement value Vsm is 0 or a positive value, the duty ratio determination unit 50 obtains the duty ratio target value α with α = α0. Further, when the system AC voltage measurement value Vsm is a negative value, the duty ratio determination unit 50 obtains the duty ratio target value α with α = 1-α0.

The process of changing the phase of switching the second switching circuit 16 with respect to the first switching circuit 12 will be described. The subtractor 52 obtains a third error obtained by subtracting the battery current measurement value ibtm from the battery current target value ibt ^* , and outputs the third error to the phase PI control unit 54. The phase PI control unit 54 obtains a third control value by proportional integration control and outputs it to the adder 55.

The FF value determining unit 56 obtains the feedforward value FF for the third control value based on the following (Equation 1), and outputs the feedforward value FF to the adder 55.

However, ωsw is a switching angular frequency when switching the first switching circuit 12 and the second switching circuit 16. Leq is the leakage inductance in the primary winding T1, where the inductances of the first inductor L1 and the second inductor L2 are L1 and L2, respectively, and the coupling coefficient between the primary winding T1 and the secondary winding T2 is k. Then, there is a relationship of Leq = (L1 + L2) · (1-k).

The adder 55 adds the feedforward value FF to the third control value to obtain the phase control value, and outputs the phase control value to the phase adjustment unit 60. The carrier generation unit 58 outputs a carrier signal for performing pulse width modulation to the phase adjustment unit 60. The carrier signal is, for example, a signal having a triangular wave as a time waveform.

The phase adjusting unit 60 further changes the phase of the carrier signal output from the carrier generation unit 58 based on the phase control value, and outputs the phase to the UV phase buffer amplifier 62. For example, the phase adjusting unit 60 delays the phase of the carrier signal as the phase control value becomes larger. The UV phase buffer amplifier 62 outputs the carrier signal output from the phase adjusting unit 60 to the UV phase control signal generation unit 64 as the U phase carrier signal CU. Further, the UV phase buffer amplifier 62 delays the carrier signal output from the phase adjusting unit 60 by 180 ° and outputs the carrier signal as a V phase carrier signal CV to the UV phase control signal generation unit 64.

The phase adjusting unit 60 outputs a carrier signal to the WX phase buffer amplifier 66. The WX phase buffer amplifier 66 outputs the carrier signal output from the phase adjusting unit 60 to the WX phase control signal generation unit 68 as the W phase carrier signal CW. Further, the WX phase buffer amplifier 66 delays the carrier signal output from the phase adjusting unit 60 by 180 ° and outputs the carrier signal as the X phase carrier signal CX to the WX phase control signal generation unit 68.

Here, the process of delaying the phase of the signal output to the UV phase buffer amplifier 62 by the phase adjusting unit 60 based on the phase control value has been described, but the phase adjusting unit 60 has described the process of delaying the phase of the signal output to the WX phase buffer amplifier 66. The process of advancing the phase based on the phase control value may be executed.

The UV phase control signal generation unit 64 generates control signals Cn1 to Cn4 as shown in FIGS. 2 and 3 based on the duty ratio target value α, the U phase carrier signal CU, and the V phase carrier signal CV. The WX phase control signal generation unit 68 generates control signals Cn5 to Cn8 whose phase is delayed by 180 ° with respect to the control signals Cn1 to Cn4 based on the duty ratio target value α, the W phase carrier signal CW and the X phase carrier signal CX. Generate.

As described above, according to the configuration shown in FIG. 8, the control unit 22A has the first control unit 22A according to the difference between the direct current ibt flowing into the second switching circuit 16 and the target value ibt ^* with respect to the direct current. The difference between the phase for switching the switching circuit 12 and the phase for switching the second switching circuit 16 is controlled.

Further, according to (Equation 1), the control unit 22A has a target value Vb ^* for the intermediate voltage Vb, a target value ibt ^* for the DC current flowing into the second switching circuit 16, and the first switching circuit 12 and the second switching circuit 16. The phase for switching the first switching circuit 12 and the phase for switching the second switching circuit 16 based on the duty ratio at the time of switching, the leakage inductance Leq in the primary winding T1, and the switching angle frequency ωsw. Control the difference with.

Further, the control unit 22A switches the first switching circuit 12 and the second switching circuit 16 at a duty ratio corresponding to the difference between the intermediate voltage Vb and the target value Vb ^* with respect to the intermediate voltage Vb.

Further, the control unit 22A has a duty ratio corresponding to the output AC voltage Vs (system AC voltage Vs) and the midpoint current iL flowing in the path drawn from the midpoint of the primary winding, in the first switching circuit. 12 and the second switching circuit 16 are switched.

Further, the control unit 22A switches the first switching circuit 12 and the second switching circuit 16 with a duty ratio corresponding to the difference between the target value Vs ^* with respect to the output AC voltage and the intermediate voltage Vb.

According to such control, the midpoint current target value iL ^* is obtained according to the difference between the intermediate voltage target value Vb ^* and the intermediate voltage measured value Vbm, and the midpoint current measured value IL is the midpoint current target value iL. A duty ratio target value α that approaches or matches ^* is obtained. Further, when the battery current measured value ibtm is less than the battery current target value ibt ^* , the phase for switching the second switching circuit 16 becomes larger than the phase for switching the first switching circuit 12. When the battery current measured value ibtm exceeds the battery current target value ibt ^* , the phase lead for switching the second switching circuit 16 becomes smaller than the phase for switching the first switching circuit 12. As a result, the battery current measurement value ibtm approaches or coincides with the battery current target value ibt ^* . Further, the feedforward value FF obtained by the FF value determining unit 56 is added to the third control value for controlling the switching phase difference | φ | of the first switching circuit 12 and the second switching circuit 16, and the phase is added. A control value is required. By such feed-forward control, even if the power consumption in the power supply system 32 changes, the power supplied from the power conversion device to the power supply system 32 quickly follows the change in the power consumption.

FIG. 9 shows the simulation results for the operation of the input current control mode. The horizontal axis in FIGS. 9 (a) to 9 (e) indicates time. FIG. 9A shows the time waveforms of the system AC voltage Vs and the midpoint current iL. However, the midpoint current iL is not obtained by simulation, but is schematically a time waveform represented by multiplying the scale of the vertical axis by 20. FIG. 9B shows the time waveforms of the intermediate voltage Vb and the battery current ibt. However, the scale of the vertical axis of the battery current ibt is set to 50 times. FIG. 9C shows the states of the switching elements S9 and S10. FIG. 9D shows a time waveform having a duty ratio α. FIG. 9E shows a phase delay φ for switching the second switching circuit 16 with respect to the phase for switching the first switching circuit 12.

(6) Output Voltage Control Mode An output voltage control mode, which is another one of the power supply modes, will be specifically described. In the output voltage control mode, for example, an electric device is connected to the AC terminals 24-1 and 24-2 as a load circuit 30. The power consumption of an electric device changes according to its operating state, and when the power supply voltage fluctuates due to the fluctuation of the power consumption, sufficient performance may not be exhibited. Therefore, in the output voltage control mode, the output AC voltage output from the AC terminals 24-1 and 24-2 is brought closer to the target value or adjusted to the target value.

FIG. 10 shows a block diagram of the control unit 22B as a configuration example of the control unit 22 in the output voltage control mode. The control unit 22B may include a processor that realizes each component shown in the figure by executing a program. Further, each component may be individually configured by an electronic circuit as hardware.

When the control unit 22B generates each control signal, the measured value Vbm of the intermediate voltage Vb and the measured value VLm of the load voltage VL (voltage between the AC terminals 24-1 and 24-2) are used. The power conversion device is provided with sensors (not shown) for measuring these.

The control unit 22B obtains the duty ratio target value α based on the difference between the absolute value of the load voltage measurement value VLm and the target value VL ^* , the load voltage target value VL ^* , and the intermediate voltage measurement value Vbm. Further, the control unit 22B changes the phase for switching the second switching circuit 16 with respect to the first switching circuit 12 based on the difference between the intermediate voltage measured value Vbm and the intermediate voltage target value Vb ^* . To generate.

The process of obtaining the duty ratio target value α will be described. The subtractor 38 subtracts the absolute value | VLm | of the load voltage measurement value VLm from the target value VL ^* with respect to the absolute value of the load voltage measurement value VLm to obtain the first voltage error, and outputs the first voltage error to the voltage PI control unit 40. The voltage PI control unit 40 obtains a duty ratio control value by proportional integration control and outputs it to the adder 48. The adder 48 adds the offset value VL ^* / Vbm to the duty ratio control value to obtain the pre-adjustment target value α0. The offset value VL ^* / Vbm has a significance as an error expressing the difference between the target value VL ^* of the absolute value of the load voltage measurement value and the intermediate voltage measurement value Vbm as a ratio.

When the load voltage measurement value VLm is 0 or a positive value, the duty ratio determination unit 50 obtains the duty ratio target value α with α = α0. Further, when the load voltage measurement value VLm is a negative value, the duty ratio determination unit 50 obtains the duty ratio target value α with α = 1-α0.

The process of changing the phase of switching the second switching circuit 16 with respect to the first switching circuit 12 will be described. The subtractor 52 obtains a second voltage error obtained by subtracting the intermediate voltage measurement value Vbm from the intermediate voltage target value Vb ^* , and outputs the second voltage error to the phase PI control unit 54. The phase PI control unit 54 obtains a phase control value by proportional integration control and outputs it to the phase adjustment unit 60.

The phase adjusting unit 60, the UV phase buffer amplifier 62, and the UV phase control signal generation unit 64 generate control signals Cn1 to Cn4 based on the carrier signal, the phase control value, and the duty ratio target value α. Further, the phase adjusting unit 60, the WX phase buffer amplifier 66, and the WX phase control signal generation unit 68 generate control signals Cn5 to Cn8 based on the carrier signal, the phase control value, and the duty ratio target value α.

According to the configuration shown in FIG. 10, the control unit 22B has a phase for switching the first switching circuit 12 and a second switching circuit 16 according to the difference between the intermediate voltage Vb and the target value Vb ^* with respect to the intermediate voltage. Controls the difference from the phase to be switched.

Further, the control unit 22B switches the first switching circuit 12 and the second switching circuit 16 with a duty ratio according to the difference between the absolute value of the output AC voltage VL and the target value VL ^* with respect to the absolute value of the output AC voltage. do.

Further, the control unit 22B switches the first switching circuit 12 and the second switching circuit 16 at a duty ratio according to the difference between the target value VL ^* with respect to the absolute value of the output AC voltage VL and the intermediate voltage Vb.

According to the control performed by the control unit 22A, the absolute value | VLm | of the load voltage measurement value VLm approaches or matches the target value, and further, the target value VL ^* of | VLm | and the intermediate voltage measurement value Vbm are A matching duty ratio target value α is obtained. Further, when the intermediate voltage measurement value Vbm is less than the intermediate voltage target value Vb ^* , the phase advance for switching the second switching circuit 16 becomes larger than the phase for switching the first switching circuit 12. When the intermediate voltage measurement value Vbm exceeds the intermediate voltage target value Vb ^* , the phase lead for switching the second switching circuit 16 becomes smaller than the phase for switching the first switching circuit 12. As a result, the intermediate voltage measurement value Vbm approaches or coincides with the intermediate voltage target value Vb ^* . Therefore, the absolute value of the load voltage VL is brought close to or adjusted to a predetermined value, and the load voltage VL is stabilized.

FIG. 11 shows the simulation results for the operation of the output voltage control mode. The horizontal axis in FIGS. 11 (a) to 11 (e) indicates time. FIG. 11A shows the time waveforms of the load voltage VL and the midpoint current iL. FIG. 11B shows a time waveform of the intermediate voltage Vb. FIG. 11C shows the states of the rectifying switching elements S9 and S10. FIG. 11D shows a time waveform having a duty ratio α. FIG. 11E shows a phase delay φ for switching the second switching circuit 16 with respect to the phase for switching the first switching circuit 12.

(7) Winding ratio of transformer T In the power conversion device, the power transmitted from the power factor improving circuit 10 to the voltage converter circuit 14 or the power transmitted from the voltage converter circuit 14 to the power factor improving circuit 10 is constant. If this is the case, the switching phase difference | φ | of the first switching circuit 12 and the second switching circuit 16 tends to be as follows. That is, the closer the duty ratio is to 1 or 0, the larger the switching phase difference, and the closer the duty ratio is to 0.5, the smaller the switching phase difference tends to be. This is because the closer the duty ratio is to 0.5, the longer the periods τ2 and τ6 in FIGS. 4 and 6, and the longer the time for power transmission. That is, under the condition that the periods τ2 and τ6 are long and the transmission power is constant, the periods τ1 and τ5 are lengthened (switching phase difference is increased) to increase the primary winding T1 or the secondary winding T2. This is because it is not necessary to increase the current flowing through the power transfer. Under the condition that the transmission power is constant, the closer the duty ratio is to 0.5, the smaller the switching phase difference becomes, and the smaller the current flowing through the primary winding T1 or the secondary winding T2. As a result, the loss due to Joule heat is reduced and the transmission efficiency is improved. In the power conversion circuit according to the present embodiment, by setting the winding ratio of the transformer T to an appropriate value, the duty ratio is brought close to 0.5 and the transmission efficiency is improved, as described below.

In the power factor improving circuit 10 described above, when the execution value of the output AC voltage V (= Vac, Vs or VL) between the AC terminal 24-1 and the AC terminal 24-2 is Vrms, the intermediate voltage target value Vb By setting ^* to 1.8 times Vrms, the duty ratio of the first switching circuit 12 and the second switching circuit 16 can be brought closer to 0.5. However, the time waveform of the output AC voltage V between the AC terminal 24-1 and the AC terminal 24-2 is a sine wave. The time average value of the time waveform obtained by full-wave rectifying a sine wave having an effective value of Vrms is 0.9 Vrms, and the duty ratio is 0 by setting the intermediate voltage Vb to twice this voltage. This is because it can be brought closer to 5. Therefore, the duty ratio approaches 0.5 by setting N = Vbat / (1.8 · Vrms) as the winding ratio N defined as the number of turns of the secondary winding T2 with respect to the number of turns of the primary winding T1. Be done. However, Vbat is the output voltage of the battery 28. Therefore, under the condition that the transmission power is constant, the duty ratio is approached to 0.5 by setting N = Vbat / (1.8 · Vrms), and the transmission efficiency is improved.

10 Power factor improvement circuit, 12 1st switching circuit, 14 Voltage converter circuit, 16 2nd switching circuit, 18 AC voltage source, 20 DC power supply circuit, 22 Control unit, 24-1, 24-2 AC terminal, 26P positive terminal , 26N Negative terminal, 28 battery, 30, 36 load circuit, 32 power supply system, 34 power transmission and distribution equipment, 38, 44, 48, 52 subtractor, 40 voltage PI control unit, 42 multiplier, 46 current PI control unit, 50 Duty ratio determination unit, 54 Phase PI control unit, 56 FF value determination unit, 58 carrier generation unit, 60 phase adjustment unit, 62 UV phase buffer amplifier, 64 UV phase control signal generation unit, 66 WX phase buffer amplifier, 68 WX Phase control signal generator.

Claims (8)

It ’s a power converter,
The first switching circuit and the second switching circuit,
A primary winding having both ends connected to the first switching circuit,
A secondary winding, both ends of which are connected to the second switching circuit and magnetically coupled to the primary winding, is provided.
The first switching circuit is
A full bridge in which two half bridges are connected in parallel, each said half bridge comprising two switching elements, one end of which is commonly connected.
A first rectifying switching element and a second rectifying switching element, one of which is connected in common, are provided.
The primary winding is
It is connected between the connection point of two switching elements in one of the two half bridges and the connection point of two switching elements in the other of the two half bridges.
The other end of the first rectifying switching element is connected to one of the two parallel connection points in the full bridge.
The other end of the second rectifying switching element is connected to the other of the two parallel connection points.
Output AC based on the DC power input to the second switching circuit from between the path drawn from the middle point of the primary winding and the connection point of the first rectifying switching element and the second rectifying switching element. The voltage is output ,
The power conversion device further
The difference between the phase for switching the first switching circuit and the phase for switching the second switching circuit is controlled according to the difference between the direct current flowing into the second switching circuit and the target value with respect to the direct current. Control unit and
A capacitor connected to two parallel connection points in the full bridge.
The control unit
The target value for the voltage between terminals of the capacitor, the target value for the DC current flowing into the second switching circuit, the duty ratio when switching between the first switching circuit and the second switching circuit, and the first winding. A power conversion device characterized in that the difference between the phase for switching the first switching circuit and the phase for switching the second switching circuit is controlled based on the leakage inductance in the line and the switching frequency. .. It ’s a power converter,
The first switching circuit and the second switching circuit,
A primary winding having both ends connected to the first switching circuit,
A secondary winding, both ends of which are connected to the second switching circuit and magnetically coupled to the primary winding, is provided.
The first switching circuit is
A full bridge in which two half bridges are connected in parallel, each said half bridge comprising two switching elements, one end of which is commonly connected.
A first rectifying switching element and a second rectifying switching element, one of which is connected in common, are provided.
The primary winding is
It is connected between the connection point of two switching elements in one of the two half bridges and the connection point of two switching elements in the other of the two half bridges.
The other end of the first rectifying switching element is connected to one of the two parallel connection points in the full bridge.
The other end of the second rectifying switching element is connected to the other of the two parallel connection points.
Output AC based on the DC power input to the second switching circuit from between the path drawn from the middle point of the primary winding and the connection point of the first rectifying switching element and the second rectifying switching element. The voltage is output,
The power conversion device further
Capacitors connected to the two parallel connection points in the full bridge,
A control unit that switches between the first switching circuit and the second switching circuit at a duty ratio corresponding to a difference between the voltage between the terminals of the capacitor and the target value with respect to the voltage between the terminals of the capacitor is provided .
The control unit
Switching the first switching circuit and the second switching circuit at a duty ratio corresponding to the output AC voltage and the midpoint current flowing in the path drawn from the midpoint of the primary winding. Characterized power conversion device. In the power conversion device according to claim 2 ,
The control unit
A power conversion device characterized in that the first switching circuit and the second switching circuit are switched at a duty ratio corresponding to a difference between a target value with respect to the output AC voltage and a voltage between terminals of the capacitor. It ’s a power converter,
The first switching circuit and the second switching circuit,
A primary winding having both ends connected to the first switching circuit,
A secondary winding, both ends of which are connected to the second switching circuit and magnetically coupled to the primary winding, is provided.
The first switching circuit is
A full bridge in which two half bridges are connected in parallel, each said half bridge comprising two switching elements, one end of which is commonly connected.
A first rectifying switching element and a second rectifying switching element, one of which is connected in common, are provided.
The primary winding is
It is connected between the connection point of two switching elements in one of the two half bridges and the connection point of two switching elements in the other of the two half bridges.
The other end of the first rectifying switching element is connected to one of the two parallel connection points in the full bridge.
The other end of the second rectifying switching element is connected to the other of the two parallel connection points.
Output AC based on the DC power input to the second switching circuit from between the path drawn from the middle point of the primary winding and the connection point of the first rectifying switching element and the second rectifying switching element. The voltage is output,
The power conversion device further
A power conversion device including a control unit that switches between the first switching circuit and the second switching circuit at a duty ratio corresponding to a difference between the output AC voltage and a target value with respect to the output AC voltage. .. In the power conversion device according to claim 4 ,
With capacitors connected to the two parallel connection points in the full bridge
The control unit
A power conversion device characterized in that the first switching circuit and the second switching circuit are switched at a duty ratio corresponding to a difference between a target value with respect to the output AC voltage and a voltage between terminals of the capacitor. In the power conversion device according to claim 4 or 5.
Capacitors connected to the two parallel connection points in the full bridge,
The difference between the phase for switching the first switching circuit and the phase for switching the second switching circuit is controlled according to the difference between the voltage between the terminals of the capacitor and the target value with respect to the voltage between the terminals of the capacitor. Control unit and
A power conversion device characterized by being equipped with. The power conversion device according to any one of claims 1 to 6.
The second switching circuit is
A secondary side full bridge in which two secondary side half bridges are connected in parallel, and each of the secondary side half bridges includes two switching elements in which one end thereof is commonly connected. Equipped with a bridge
The secondary winding is
It is connected between the connection point of the two switching elements in one of the two secondary side half bridges and the connection point of the two switching elements in the other of the two secondary side half bridges. ,
A power conversion device characterized in that DC power is input from two parallel connection points in the secondary side full bridge. The power conversion device according to any one of claims 1 to 7 .
A power conversion device characterized in that each of the first rectifying switching element and the second rectifying switching element is turned on and off according to the polarity of the output AC voltage.

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