JP6971957B2 - Power factor compensation power supply and LED lighting - Google Patents

Description

The present application relates to a power factor compensating power supply and an LED lighting device.

Conventionally, a power factor compensating power supply device having a power factor improvement (PFC: Power Factor Direction) function and converting AC power into DC power is equipped with a power factor improvement circuit having a full-wave rectifier circuit and a converter, and the output voltage of the converter. A switching element that calculates the reference on-time from the detected value, generates a correction signal based on the phase information of the input voltage after full-wave rectification, and uses the reference on-time corrected by this correction signal to form a converter. On / off control was performed (see, for example, Patent Document 1).

International Publication No. 2018-0879060

In the power factor improving function as described above, there is a period that contributes to the power factor improvement and a period that does not contribute to the power factor improvement depending on the phase of the input AC voltage. However, in the power factor compensating power supply device described in Patent Document 1, since the switching element is controlled on / off during the entire period, switching loss is generated even in a region that does not contribute to the improvement of the power factor, and the switching loss increases. There is a problem that efficiency is reduced.
The present application discloses a technique for solving the above-mentioned problems, and obtains a power factor compensating power supply device and an LED lighting device capable of achieving a high power factor while suppressing an increase in switching loss. With the goal.

The power factor compensating power supply device disclosed in the present application includes a power supply main circuit unit and a power supply control unit that controls the power supply main circuit unit, and the power supply main circuit unit performs full-wave rectification of the AC voltage supplied from the AC power supply. By switching the switching element with a converter that has a full-wave rectifying circuit and a switching element, and the AC voltage that is full-wave rectified by the full-wave rectifying circuit is input as an input voltage and converts the input voltage to the target output voltage. It has an input capacitor that suppresses the conduction of generated noise to an AC power supply, an input voltage detection unit that detects the input voltage, and an output voltage detection unit that detects the output voltage. The reference on time of the switching element is determined based on this, and the reference on time is corrected by the correction amount calculated based on the phase difference between the input current from the AC power supply and the input voltage and the phase of the input voltage to turn on the switching element. It is a power supply control unit that calculates the time and controls the switching element based on the on-time. The power supply control unit compares the predetermined upper limit threshold and lower limit threshold with the correction amount, and the correction amount is equal to or larger than the upper limit threshold. In the case of, and when the correction amount is equal to or less than the lower limit threshold value, the on-time is set to zero and the switching of the switching element is stopped.

Further, the LED lighting device disclosed in the present application includes a power supply main circuit unit to which an LED module is connected to the output side and a power supply control unit that controls the power supply main circuit unit, and the power supply main circuit unit is from an AC power source. A full-wave rectifier circuit that full-wave rectifies the supplied AC voltage, and a converter that has a switching element and converts the input current obtained by the full-wave rectifier circuit to full-wave rectify the AC voltage into the target output current. An input capacitor that suppresses the conduction of noise generated by switching of the switching element to the AC power supply, and an input voltage detector that detects the input voltage obtained by the full-wave rectifier circuit performing full-wave rectification of the AC voltage. It has an output current detector that detects the output current, and the power supply control unit determines the reference on-time of the switching element based on the output current, and the phase difference between the input current and the input voltage from the AC power supply and the input voltage. It is a power supply control unit that corrects the reference on-time by the correction amount calculated based on the phase of, calculates the on-time of the switching element, and controls the switching element based on the on-time. When the correction amount is equal to or greater than the upper limit threshold and the correction amount is equal to or less than the lower limit threshold, the on-time is set to zero and the switching of the switching element is stopped by comparing the determined upper limit threshold and the lower limit threshold with the correction amount.

According to the power factor compensating power supply device disclosed in the present application, it is possible to realize a high power factor while suppressing an increase in switching loss.

It is a block diagram which shows the structure of the power factor compensation power supply device in Embodiment 1. FIG. It is a figure which shows the relationship between the pulsating current voltage and the input voltage phase which concerns on Embodiments 1-5. It is a figure explaining the determination of the reference on time which concerns on Embodiment 1. FIG. It is a figure explaining the detection of the phase of a pulsating current voltage. It is a figure explaining the example of the on / off control of a switching element by the switch control unit which concerns on Embodiment 1. FIG. It is a figure explaining another example of on / off control of a switching element by a switch control part which concerns on Embodiment 1. FIG. It is a figure which shows the equivalent circuit of the input filter and a converter which concerns on Embodiment 1. FIG. It is a vector diagram of the AC input voltage and the AC input current when the phase difference between the AC input voltage and the AC input current is not adjusted. It is a vector diagram of the AC input voltage and the AC input current when adjusting the phase difference of the AC input voltage and the AC input current. It is a figure which shows the structural example of the input coil which concerns on Embodiment 1. FIG. It is a figure which shows the structural example of the input coil which concerns on Embodiment 1. FIG. It is a figure which shows the structural example of the input coil which concerns on Embodiment 1. FIG. It is a figure which shows the input voltage of the converter which concerns on Embodiment 1. FIG. It is a waveform diagram which shows the simulation result of a reference example. It is a waveform diagram which shows the simulation result of Embodiment 1. FIG. It is a figure which shows the example of the upper limit threshold value and the lower limit threshold value corresponding to each phase difference in Embodiment 2. It is a waveform diagram which shows the waveform of the AC input current when there is a phase lead. It is a waveform diagram which shows the waveform of the AC input current when there is a phase lag. It is a figure which shows the limiter correction amount when there is a phase lead in an AC input current. It is a figure which shows the limiter correction amount when there is a phase lag in an AC input current. It is a figure which shows the relationship between the phase of an AC input voltage, and the limiter correction amount when the phase difference of an AC input voltage and an AC input current is changed. It is a figure which shows the relationship between the output power, the phase difference of AC input voltage, and AC input current. It is a figure which shows the relationship between the AC input voltage, and the phase difference of the AC input voltage and the AC input current. It is a figure which shows the relationship between the AC input voltage frequency, and the phase difference of AC input voltage and AC input current. It is a waveform diagram which shows the simulation result of the reference example when the output power is 50W. It is a waveform diagram which shows the simulation result of Embodiment 3 when the output power is 50W. It is a waveform diagram which shows the simulation result of Embodiment 1 when the output power is 50W. It is a waveform diagram which shows the simulation result of the reference example when the output power is 5W. It is a waveform diagram which shows the simulation result of Embodiment 3 when the output power is 5W. It is a waveform diagram which shows the simulation result of Embodiment 1 when the output power is 5W. It is a block diagram which shows the structure of the LED lighting apparatus in Embodiment 4. It is a block diagram which shows the structure of the LED lighting apparatus in Embodiment 5.

Embodiment 1.
Hereinafter, the first embodiment will be described with reference to FIGS. 1 to 10. FIG. 1 is a block diagram showing a configuration of a power factor compensating power supply device according to the first embodiment. The power factor compensating power supply device 1 according to the first embodiment includes a power supply main circuit unit 2 and a power supply control unit 3 for direct current.
First, the power supply main circuit unit 2 will be described. The power supply main circuit unit 2 is mainly composed of a full-wave rectifier circuit 14, an input filter 21, a DC / DC converter (hereinafter, simply referred to as a converter) 4, and an output capacitor 15.
The full-wave rectifier circuit 14 is composed of a diode bridge (not shown), and is a full-wave rectified voltage | vac | (hereinafter referred to as pulsating voltage) by full-wave rectifying the AC input voltage vac supplied from the AC power supply 5. Is output. The input filter 21 is composed of an input coil 20, an input capacitor 6a, and an input capacitor 6 having a capacitance of Cin, and noise generated by switching to an AC power supply 5 by a switching element 7 described later is generated. Suppresses conduction to the AC power supply 5.

The converter 4 converts the pulsating current voltage | vac |, which has been fully wave rectified by the full wave rectifier circuit 14, into a direct current and adjusts the voltage to a target voltage for output. The output capacitor 15 having a capacitance of Co smoothes the pulsation of the voltage output by the converter 4 and outputs it as a DC output voltage vo to the output side of the power supply main circuit unit 2. A load 8 is connected to the output side of the DC power, and an output voltage vo is applied to the load 8.

In the first embodiment, the converter 4 is composed of a step-up chopper circuit. More specifically, in the converter 4, the connection point between one end of the full-wave rectifier circuit 14 and one end of the input capacitor 6 is connected to one end of the reactor 18 as an inductance element, and the other end of the reactor 18 is the anode of the diode 17. Is connected to. Further, the cathode terminal of the diode 17 is connected to the connection point between one end of the output capacitor 15 and one end of the load 8. Further, the connection point between the reactor 18 and the diode 17 is connected to one end of the switching element 7, and the other end of the switching element 7 is connected to the connection point between the other end of the output capacitor 15 and the other end of the load 8. The inductance of the reactor 18 is L.

A FET (Field Effect Transistor) element driven by an element drive signal Vg generated by the power supply control unit 3, an IGBT (Insulated Gate Bipolar Transistor) element, or the like is applied to the switching element 7. When the switching element 7 is an FET, one end and the other end of the switching element 7 described above are a drain terminal and a source terminal in the case of the FET element.

It is also possible to change the diode 17 to a switching element 171 (not shown) such as an FET element or an IGBT element, and use a synchronous rectification method in which the switching elements 7 and 171 are turned on and off by reverse logic.

In addition to the configuration described above, the power supply main circuit unit 2 is provided with a zero current detection unit 16, an input voltage detection unit 10, and an output voltage detection unit 9. The input voltage detection unit 10 is composed of a plurality of voltage dividing resistors connected in series (not shown), detects a voltage value of pulsating voltage | vac |, and outputs it as an input voltage detection value Vinsen which is an electric signal. do. The output voltage detection unit 9 is composed of a plurality of voltage dividing resistors (not shown) connected in series, detects the voltage value of the DC output voltage vo, and outputs it as the output voltage detection value vosen.

The zero current detection unit 16 has, for example, a current detection resistor, and detects the current iL flowing through the reactor 18 based on the potential difference generated between both ends of the current detection resistor. Further, the zero current detection unit 16 outputs the potential difference between both ends of the above-mentioned current detection resistor as a voltage conversion value iLsen. The voltage conversion value iLsen is a voltage value corresponding to the reactor current iL. The zero current detection unit 16 is not limited to the method of detecting on the low potential side as shown in FIG. 1, and the current detection resistor is connected in series with the reactor 18 to perform voltage conversion corresponding to the reactor current iL on the high potential side. A method of detecting the value iLsen can also be adopted. Further, the reactor 18 may be provided with an auxiliary winding having the opposite polarity without using a current detection resistor, and the voltage obtained from the auxiliary winding may be detected as the voltage conversion value iLsen. The voltage conversion value iLsen is used to determine the turn-on timing of the switching element 7 as described later.

Next, the power supply control unit 3 will be described. The power supply control unit 3 includes an input voltage phase detection unit 12, an output voltage control unit 11, and a switch control unit 13. The input voltage phase detection unit 12 detects the phase vinphase of the pulsating voltage | vac | (hereinafter, this may be referred to as θ) from the input voltage detection value vinsen. Hereinafter, the phase θ (= vinphase) will be referred to as an input voltage phase θ. In the first embodiment and the second to fifth embodiments described later, the input voltage phase θ is expressed with reference to the bottom of the pulsating voltage | vac | as shown in FIG. Since the pulsating voltage | vac | has a frequency twice that of the AC input voltage vac, one cycle of the pulsating voltage | vac is expressed from 0 degrees to 180 degrees (0 to π).

The output voltage control unit 11 acquires the output voltage detection value vosen and the preset target voltage value voref, obtains the difference between the two, and performs a control calculation to perform a control calculation, so that the time becomes a reference for the on-time of the switching element 7 ( Hereinafter, ton1 (referred to as reference on time) is determined. The output voltage control unit 11 determines the reference on-time ton1 at the timing when the input voltage phase detection unit 12 detects the bottom of the pulsating voltage | vac |.

The switch control unit 13 calculates the on-time ton2 of the element drive signal Vg for driving the switching element 7. Specifically, first, the switch control unit 13 acquires the reference on-time ton1 from the output voltage control unit 11 and the input voltage phase θ (= vinphase) from the input voltage phase detection unit 12. Next, the switch control unit 13 determines the correction amount Kphase according to the input voltage phase θ, and further determines the limiter correction amount Kphase * by performing limiter processing on the correction amount Kphase. After determining the limiter correction amount Kphase *, the switch control unit 13 determines the on-time ton2 of the element drive signal Vg by multiplying the reference on-time ton1 by the limiter correction amount Kphase *. The correction amount Kphase and the limiter correction amount Kphase * will be described later.

Further, the switch control unit 13 determines the on-timing of the element drive signal Vg according to the voltage conversion value iLsen corresponding to the reactor current iL detected by the zero current detection unit 16. By determining the on-time ton2 and the on-timing of the element drive signal Vg in this way, the switch control unit 13 controls the switching element 7.

For the realization of each functional unit of the power supply control unit 3, a digital circuit including an IC may be used, or a digital control circuit not including an IC (including a circuit by software having the same function) may be used. Further, a part or all of the components may be analog control circuits. In the first embodiment, a case where the configuration is made by a digital control circuit using a microcomputer is described as an example.

Next, the processing operations of the output voltage control unit 11, the input voltage phase detection unit 12, and the switch control unit 13 constituting the power supply control unit 3 will be described more specifically.
First, a method of determining the reference on-time ton1 by the output voltage control unit 11 will be described. The output voltage control unit 11 is the timing of the input voltage phase θ (= vinphase) of 0 and π, that is, the frequency of the pulsating voltage | vac | after full-wave rectification when the commercial frequency is 50 Hz (period 20 ms). The reference on-time ton1 is determined at intervals of 100 Hz (period 10 ms). As a result, the average current iLmean per unit time of the reactor current iL can be controlled in the same phase as the pulsating current voltage | vac |, so that the power factor improvement control can be performed. For example, when the converter 4 is a step-up chopper circuit, the pulsating current voltage | The relationship is proportional to vac |. Utilizing this fact, during one cycle of the pulsating current voltage | vac |, as shown in FIG. 3, the reference on-time ton1 of the switching element 7 is kept constant to perform the current critical operation.

ΔiL = ton / L × | vac | (1)

For example, in FIG. 3, the reference on time is set to ton1a in one cycle Ta of the pulsating current voltage | vac |, and the reference on time is set to ton1b in the next one cycle Tb. Therefore, the average current iLmean is controlled in phase with the pulsating current voltage | vac | in each cycle. The bottoms 0 and π of the input voltage phase θ of the pulsating current voltage | vac | are detected by the input voltage phase detection unit 12, and the method of the detection timing will be described later.

The output voltage control unit 11 calculates the difference between the output voltage detection value vosen and the target voltage value voref, and determines the reference on-time ton1 for the switching element 7 so that the difference becomes zero. Specifically, the reference on-time ton1 is determined by classical control such as proportional integral (PI) control or proportional integral differential (PID) control, or modern control such as H∞ (H-infinity) control. Here, in order to perform power factor improvement control, it is described that the output voltage control unit 11 performs a control calculation of the reference on-time ton1 at the timing when the input voltage phase θ (= vinphase) is 0 or π. However, the present invention is not limited to this method, and for example, a control calculation may be performed at each turn-on timing of the switching element 7. In this case, the high frequency component of the output voltage detection value vosen can be sufficiently attenuated by adjusting an RC filter or the like (not shown) provided in the output voltage detection unit 9, or the control coefficient (gain) used for the control calculation is sufficiently sufficient. By making it smaller, the same effect as the above-mentioned method can be obtained.

The input voltage phase detection unit 12 detects the phase (0 to π) of the pulsating voltage | vac | from the input voltage detection value vinsen, for example, as follows. That is, the input voltage detection value vinsen is sampled at regular intervals by an AD converter or the like (not shown), and the latest sampling value is x (n) and the value before one sampling is x as shown by the ● mark in FIG. (N-1) When the value before 2 samplings is x (n-2) ..., The timing is such that x (n-2)> x (n-1) <x (n). The point x (n) is the timing at which the bottom of the pulsating voltage | vac | is detected. This bottom detection timing is set to "zero", and while performing count-up processing for each sampling of the AD converter or the like, the sampling frequency fsamp, the commercial frequency fcom of the AC power supply 5, and the count-up number n are used to perform the following (2). ), The input voltage phase θ (= frequency) is calculated.

θ = vinphase = 2 × fcom × n × π / fsamp (2)

In the above-mentioned input voltage phase θ (= vinphase) detection method, a pulsating voltage bottom detection delay of up to two sampling cycles occurs. In order to reduce the influence of this pulsating voltage bottom detection delay, for example, the input voltage phase θ may be determined using the following formula (3) assuming a one-cycle delay, which is the average delay value. good.

θ = vinphase = 2 × fcom × (n + 1) × π / fsamp (3)

The switch control unit 13 * determines the limiter correction amount Kphase by the correction amount adjustment formula shown in the following equation (8) according to the input voltage phase θ calculated by the above method. Further, the limiter correction amount Kphase * is determined from the correction amount Kphase by the limiter process described later. The limiter correction amount Kphase * is a correction signal for the reference on-time ton1, and the switch control unit 13 multiplies the reference on-time ton1 by the limiter correction amount Kphase * as shown in the following equation (4) to obtain an element drive signal. The on-time ton2 of Vg is determined.

ton2 = ton1 x Kphase * (4)

It is preferable that the correction operation shown in the equation (4) is performed every time the output cycle of the element drive signal Vg, that is, the element drive signal Vg is output.

Next, on / off control of the switching element 7 by the switch control unit 13 will be described. FIG. 5A is a diagram showing the relationship between the voltage conversion value iLsen obtained by the zero current detection unit 16 and the element drive signal Vg, and is a diagram illustrating an example of on / off control of the switching element 7 by the switch control unit 13. .. The switch control unit 13 turns off the switching element 7 by setting the element drive signal Vg to zero after keeping the switching element 7 on-controlled by the element drive signal Vg in the on state for the on-time ton2. .. When the switching element 7 is turned off, the current iL flowing through the reactor 18 gradually decreases, and the voltage conversion value iLsen obtained by the zero current detection unit 16 also decreases as shown in FIG. 5A. The switch control unit 13 switches the element drive signal Vg to the on state again at the timing when the threshold voltage iLth preset in the comparator (not shown) falls below the voltage conversion value iLsen, and maintains the on state for the on time ton2. do.
The comparison of the voltage conversion value iLsen with the threshold voltage iLth uses a comparator here, but is not limited to this, and for example, the voltage conversion value iLsen is used as a drive signal input terminal (gate terminal in the case of FET) of the switching element 7. ), And the on-threshold voltage of the switching element 7 is configured as a substitute for iLth, so that the timing at which the voltage conversion value iLsen falls below iLth can be detected. In this case, it is preferable to adjust the threshold voltage iLth so as to perform critical operation in consideration of the delay of the element such as the comparator used for comparing the capacity of the capacitor included in the zero current detection unit 16 and the voltage level in advance.

As described above, the on / off control of the switching element 7 is performed by determining the on control timing of the switching element 7 based on the voltage conversion value iLsen and determining the on time ton2 of the switching element 7 by the equation (4). By performing the above, it is possible to achieve not only the efficiency improvement by the critical operation but also the power factor improvement by the correction calculation while controlling the output voltage vo to the target value.
In the example shown in FIG. 5A, the zero current detection unit 16 is configured by the current detection resistor to obtain the voltage conversion value iLsen as shown in FIG. 1. However, as described above, the reactor 18 has an auxiliary winding having the opposite polarity. The waveform shown in FIG. 5B can be obtained when the voltage obtained from the auxiliary winding is obtained as the voltage conversion value iLsen. Even in this case, similarly to the example shown in FIG. 5A, the voltage conversion value iLsen and the threshold voltage iLth can be compared by the comparator to determine the timing of the on-control of the switching element 7.

Next, the correction amount Kphase and the limiter correction amount Kphase * will be described. FIG. 6A is an equivalent circuit of the input filter 21 and the converter 4. Here, the current flowing through the converter 4 is represented by a current source I * (hereinafter, a vector is indicated by marking the current or voltage with *), and the input coil 20 constituting the input filter 21 is used. In the reactor L, the input capacitors 6 and 6a are represented as the combined capacitance C. The full-wave rectifier circuit 14 is omitted for simplification. Further, as the input coil 20 shown in FIG. 6A, a hybrid choke coil 20A showing an equivalent circuit in FIG. 7A can be used. In this case, the input coil 20 simultaneously reduces common mode noise and normal mode noise. The input coil 20 is not limited to the hybrid choke coil 20A described above, and the normal mode choke coil 20B shown in FIG. 7B may be used, or the common mode choke coil 20C shown in FIG. 7C may be used. You may use it.

First, due to the influence of the combined capacitance C of the capacitors constituting the input filter 21, a phase difference Φ which is a leading phase with respect to the AC input voltage vac * is generated in the AC input current iac *. When this phase difference is not adjusted, the vector diagram of the AC input current iac * and the AC input voltage vac * is as shown in FIG. 6B. Here, assuming that the current I * flowing through the converter 4 is zero (I * = 0), the current Ic * flowing through the combined capacitance C of the input filter 21 can be obtained by the following equation (5).

入力フィルタ２１のＬＣ共振周波数は、交流電源５の周波数よりも十分に高いので（５）式の分母は「正」となり、合成容量Ｃに流れる電流Ｉｃ＊は進み電流となるため、交流入力電流ｉａｃの位相が交流入力電圧ｖａｃの位相に対して所定の位相差Φ分だけ進んでいることがわかる。

Since the LC resonance frequency of the input filter 21 is sufficiently higher than the frequency of the AC power supply 5, the denominator of Eq. (5) is "positive", and the current Ic * flowing through the combined capacitance C becomes the lead current, so that the AC input current It can be seen that the phase of iac advances by a predetermined phase difference Φ with respect to the phase of the AC input voltage vac.

Therefore, as shown in FIG. 6C, the current I * = Ip * -Ic * (however, Ip *) for the current I * flowing through the converter 4 so that the AC input current iac has the same phase as the AC input voltage vac. Indicates the active current), and the phase of the current I * flowing through the converter 4 is corrected (delayed) by Φ and controlled to make the AC input current iac * only the active current Ip * component. .. Here, if the load 8 for the converter 4 is represented by the resistance R and the output voltage vo, and the loss in the power supply circuit is ignored, the effective current Ip * can be obtained by the following equation (6) from the energy conservation law.

上記の（５）式および（６）式より、交流入力電流ｉａｃを交流入力電圧ｖａｃと同位相とするために、コンバータ４に流す電流Ｉ＊において補正すべき位相差Φは、次の（７）式で算出することができる。

From the above equations (5) and (6), the phase difference Φ to be corrected in the current I * flowing through the converter 4 in order to make the AC input current iac in phase with the AC input voltage vac is as follows (7). ) Can be calculated.

スイッチ制御部１３は、（７）式で算出される位相差Φを用いて、スイッチング素子７の基準オン時間ｔｏｎ１に対する補正量Ｋｐｈａｓｅを決定する。すなわち、入力電圧位相検出部１２で検出される入力電圧位相θ（＝ｖｉｎｐｈａｓｅ）を用いて、次の（８）式に示すように、力率「１」とする理想電流波形（√２・ｉａｃ・ｓｉｎθ）と、（７）式を用いて算出する補正電流波形（√２・ｉａｃ・ｓｉｎ（θ−Φ））との比率を補正量Ｋｐｈａｓｅとする。

Ｋｐｈａｓｅ
＝（√２・ｉａｃ・ｓｉｎ（θ−Φ））／（√２・ｉａｃ・ｓｉｎθ）
＝ｓｉｎ（θ−Φ）／ｓｉｎθ （８）

すなわち、入力コンデンサ６、６ａによる、交流入力電流ｉａｃと、交流入力電圧ｖａｃとの位相差Φに応じた補正電流波形と、上記位相差Φを零とした理想電流波形とを用いて、上記補正電流波形から上記理想電流波形を除算したものが補正量Ｋｐｈａｓｅである。したがって、交流入力電流ｉａｃを交流入力電圧ｖａｃと同位相（Φ＝０）の場合、すなわち力率が「１」の場合には、補正量Ｋｐｈａｓｅ＝１となる。なお、入力電圧位相θ（＝ｖｉｎｐｈａｓｅ）は上述した（２）式から求められる。

The switch control unit 13 determines the correction amount Kphase for the reference on-time ton1 of the switching element 7 by using the phase difference Φ calculated by the equation (7). That is, using the input voltage phase θ (= vinphase) detected by the input voltage phase detection unit 12, the ideal current waveform (√2 · iac) having a power factor of “1” is set as shown in the following equation (8). The ratio between sinθ) and the corrected current waveform (√2 · iac · sin (θ−Φ)) calculated using Eq. (7) is defined as the correction amount Kphase.

Kphase
= (√2 ・ iac ・ sin (θ−Φ)) / (√2 ・ iac ・ sinθ)
= Sin (θ-Φ) / sinθ (8)

That is, the correction current waveform according to the phase difference Φ between the AC input current iac and the AC input voltage vac by the input capacitors 6 and 6a, and the ideal current waveform with the phase difference Φ set to zero are used for the correction. The correction amount Kphase is obtained by dividing the above ideal current waveform from the current waveform. Therefore, when the AC input current iac is in phase with the AC input voltage vac (Φ = 0), that is, when the power factor is “1”, the correction amount Kphase = 1. The input voltage phase θ (= vinphase) can be obtained from the above-mentioned equation (2).

After calculating the correction amount Kphase, the limiter correction amount Kphase * is obtained. When obtaining the limiter correction amount Kphase *, the upper limit threshold value and the lower limit threshold value are set in advance, and when the correction amount Kphase is equal to or more than the upper limit threshold value and when the correction amount Kphase is equal to or less than the lower limit threshold value, the value of the limiter correction amount Kphase *. Is set to zero. When the correction amount Kphase is larger than the lower limit threshold value and less than the upper limit threshold value, the value of the limiter correction amount Kphase * is the value of the correction amount Kphase.

As an example, the case where the upper limit threshold value is set to 2 and the lower limit threshold value is set to 0 is shown in Eq. (9). Hereinafter, a case where the upper limit threshold value is set to 2 and the lower limit threshold value is set to 0 will be described.

Kphase * = 0 (when Kphase ≧ 2)
Kphase * = Kphase (when 0 <Kphase <2) (9)
Kphase * = 0 (when Kphase ≤ 0)

After deriving the limiter correction amount Kphase * as described above, the switch control unit 13 corrects the reference on-time ton1 by the limiter correction amount Kphase * using the above equation (4), and the element for the switching element 7. The on-time ton2 of the drive signal Vg is finally determined. Since the correction of the reference on-time ton1 according to the equation (4) is performed every time the element drive signal Vg is output, the limiter correction amount Kphase required for this correction is also calculated every time the element drive signal Vg is output. Will be implemented in.

As described above, the calculation of the reference on-time ton1 is performed for each cycle of the pulsating current voltage | vac |, while the calculation of the limiter correction amount Kphase and the correction of the reference on-time ton1 by the equation (4) are performed by the element drive signal Vg. Is executed every time is output. Here, from equations (8) and (9), the correction amount Kphase and the limiter correction amount Kphase * are functions of the input voltage phase θ (= vinphase), and the input voltage phase θ is a function of time. During one cycle of voltage | vac |, there is a period in which the value of the correction amount Kphase is equal to or greater than the upper limit threshold value (= 2) or equal to or less than the lower limit threshold value (= 0), and the limiter correction amount Kphase * becomes 0. During the period when the limiter correction amount Kphase * is 0, the on-time ton2 of the element drive signal Vg becomes 0 regardless of the reference on-time ton1, and the switching element 7 is not switched. As described above, in the first embodiment, there is a period in which the switching of the switching element 7 is stopped by the correction by the limiter correction amount Kphase *. Further, no switching loss occurs during the period in which the switching of the switching element 7 is stopped. During the period in which switching is stopped by the limiter correction amount Kphase *, the switching element 7 is fixed off.

Here, the reason why the power factor improvement can be realized even if the period for stopping the switching of the switching element 7 is set will be described. FIG. 8 is an explanatory diagram of the input voltage of the converter 4, and shows the relationship between the AC input voltage vac and the pulsating current voltage | vac |. As shown in FIG. 1, the power supply main circuit unit 2 includes an input capacitor 6 after the rectifier circuit 14. Therefore, the pulsating current voltage | vac | input to the converter 4 has the same magnitude as the AC input voltage vac when the AC input voltage vac is large, but the AC input voltage vac | when the AC input voltage vac is small. And the size is different. During the period when the AC input voltage vac and the pulsating current voltage | vac | are different in magnitude, the energy supplied to the converter 4 is sent from the input capacitor 6, and no current flows through the AC power supply 5. Therefore, the period in which the magnitudes of the pulsating current voltage | vac | and the AC input voltage vac are different is a period that does not contribute to the improvement of the power factor. Since this period is a period in which the AC input voltage vac is small, as shown in FIG. 8, it is a period in which the input voltage phase θ is near 0 degrees or around 180 degrees.

In the first embodiment, the switching of the switching element 7 is stopped during the period when the input voltage phase θ is near 0 degrees or around 180 degrees. Specifically, for example, when the upper limit threshold value for calculating the limiter correction amount Kphase * is set to 2 and the lower limit threshold value is set to 0, and the phase difference Φ calculated by the equation (7) is 10 degrees, (8). The range in which the correction amount Kphase calculated from the equation () is equal to or less than the lower limit threshold value or greater than or equal to the upper limit threshold value is a range in which the input voltage phase θ is 0 to 10 degrees and 170 to 180 degrees. As described above, in the first embodiment, the period in which the switching of the switching element 7 is stopped is the period in which the input voltage phase θ is in the vicinity of 0 degrees or in the vicinity of 180 degrees, which coincides with the period in which the force factor improvement is not contributed. Therefore, even if a period for stopping the switching of the switching element 7 by the limiter correction amount Kphase * is provided, the influence on the power factor improvement is small.

If the input capacitor 6 is not provided, there is no period that does not contribute to the above-mentioned power factor improvement. In this case, the period for stopping the switching of the switching element 7 is the period during which the AC input voltage vac is small, and the AC input voltage. The contribution to power factor improvement is small compared to the period when vac is large. Therefore, even if the switching of the switching element 7 is stopped, the influence on the power factor improvement is small, and the effect of the first embodiment of reducing the switching loss while improving the power factor can be obtained. However, since the effect of improving the power factor is considered to be smaller than that without the input capacitor 6, it is desirable to apply the first embodiment when the input capacitor 6 is present.

Next, the simulation result of the first embodiment will be described. The simulation is performed using a circuit simulation manufactured by Myway Plus. The simulation conditions are AC input voltage vac = 200V (50Hz: pulsating voltage | vac | frequency is 100Hz), output voltage vo = 330V, load 8 resistance R = 2000Ω (power 54.45W), input of input filter 21. The reactance L of the coil 20 is 7 mH, and the combined capacitance C of the input capacitors 6 and 6a of the input filter 21 is 0.42uF. Using equation (7), the phase difference Φ of the AC input current iac with respect to the AC input voltage vac was Φ = 10.98 [degrees], so simulation was performed using equation (8) with Φ = 10.98 degrees. Was done. The upper limit threshold value for determining the limiter correction amount Kphase * was set to 2, and the lower limit threshold value was set to 0.

FIG. 9 shows the simulation results of the reference example to which the first embodiment is not applied, and FIG. 10 shows the simulation results to which the first embodiment is applied. In FIGS. 9 and 10, in order from the top, the output voltage detection value voten and the target voltage value waveform, the limiter correction amount Kphase * according to the input voltage phase θ (= vinphase), the reference on-time ton1 and the limiter correction amount Kphase *. The on-time ton2 of the element drive signal Vg, which is the multiplication value of, and the waveforms of the element drive signal Vg, the AC input voltage vac, and the AC input current iac are shown. In FIG. 9, which is a reference example, the value of the limiter correction amount Kphase * is fixed at 1 for convenience of comparison with FIG. 10. Further, in FIGS. 9 and 10, the waveforms of the AC input voltage vac and the AC input current iac in the lowermost stage are for comparing the phase difference between the two, and the AC input current iac is multiplied by 600 for easy comparison. The value is displayed. Further, in the figure showing the waveform of the element drive signal Vg, it is shown that the shaded area is the area where the switching element 7 is switched.

In the example shown in FIG. 9 and the example shown in FIG. 10, it is common that the on-time ton2 of the element drive signal Vg changes so as to correct the phase difference Φ between the AC input voltage vac and the AC input current iac. Both have achieved a power factor exceeding 0.98, but the mode of change of the on-time ton2 of the element drive signal Vg is different between the two. That is, in FIG. 9 as a reference example, since the value of the limiter correction amount Kphase * is fixed at 1 as described above, the on-time ton2 of the element drive signal Vg outputs the reference on-time ton1 as it is (). ton2 = ton1). In this case, it can be confirmed from the waveform of the element drive signal Vg that the switching element 7 always performs switching. On the other hand, in FIG. 10 showing the simulation result of the first embodiment, the reference on-time ton1 is corrected by the limiter correction amount Kphase * that changes according to the input voltage phase θ, and the on-time ton2 of the element drive signal Vg is calculated. The on-time ton2 of the element drive signal Vg changes significantly with respect to the reference on-time ton1.

By checking the waveforms at the bottom of FIGS. 9 and 10, in the reference example, the phase difference due to the phase lead of the AC input current iac with respect to the AC input voltage vac can be confirmed, but in the first embodiment, the AC input voltage vac and the AC input current It can be confirmed that the phase difference of iac is almost zero. When the power factor was measured by the circuit simulation, it was found that the power factor was 0.985 in the case of the reference example, whereas it was improved to the power factor = 0.997 in the case of the first embodiment.

Further, in the first embodiment, it can be confirmed from FIG. 10 that there is a period in which the element drive signal Vg is fixed to 0, that is, a period in which the switching of the switching element 7 is stopped. From this, it can be confirmed that when the first embodiment is applied, the increase in switching loss can be suppressed as compared with the reference example.

Here, in the calculation of the correction amount Kphase, the method of obtaining the phase difference Φ by the calculation using the equation (7) is shown, but the phase detection means of the AC input current iac is added to the configuration shown in FIG. The phase difference Φ may be obtained from the phases of both the actually measured AC input voltage vac and the AC input current iac, and further, the phase difference Φ may be obtained in advance by simulation or experiment.

According to the first embodiment, a high power factor can be realized while suppressing an increase in switching loss. More specifically, when calculating the on-time of the element drive signal that drives the switching element of the converter, the reference on-time obtained from the switching element output voltage detection value and the target voltage value, and the input voltage phase of the pulsating voltage. The limiter correction amount obtained by applying the limiter processing to the correction amount calculated based on the above was calculated, and the result of multiplying the reference on time by the limiter correction amount was taken as the on time of the element drive signal. Then, by controlling the on / off of the switching element using the on time calculated in this way, the influence of the phase advance of the AC input current on the AC input voltage is suppressed. As a result, the AC input current and the AC input voltage can be accurately in phase and have the same waveform, and the power factor is improved as compared with the conventional case. Further, since the switching is stopped during the period in which the contribution to the power factor improvement is small by the limiter process, the increase in the switching loss is suppressed without significantly affecting the power factor improvement.

Embodiment 2.
Hereinafter, the second embodiment will be described with reference to FIG.
The second embodiment is different from the first embodiment in that the upper limit threshold value and the lower limit threshold value for obtaining the limiter correction amount Kphase * are determined according to the phase difference Φ. FIG. 11 shows an upper limit threshold value and a lower limit threshold value for each when the phase difference Φ is changed from 5 degrees to 45 degrees in increments of 5 degrees. In the second embodiment, as shown in the figure, the smaller the phase difference Φ is, the larger the value of the upper limit threshold value is, and the shorter the period for stopping the switching of the switching element 7. When the phase difference Φ is small, as shown in FIG. 6B, the current Ic * is smaller than the current I *, so that the current flowing through the converter 4 is larger than the current flowing through the input capacitors 6 and 6a of the input filter 21. , The output power is large. In such a state, switching of the switching element 7 is performed in a period outside the ideal period for stopping the switching of the switching element 7 (for example, a period in which the input voltage phase θ is 0 to 10 degrees or 170 to 180 degrees). If the power factor is stopped, the input current iac may be distorted and the power factor may be lowered.

In order to prevent the above-mentioned decrease in the power factor, it is necessary to prevent the period for stopping the switching of the switching element 7 from deviating from the ideal period, but the input voltage phase θ used for calculating the correction amount Kphase is the input voltage. Since the input voltage phase is vinphase detected by the phase detection unit 12, there is a possibility that a deviation from the actual phase of the AC input voltage vac may occur due to a detection error or the like. When such a deviation occurs, the correction amount Kphase calculated by the equation (8) has a phase lead or a phase delay with respect to the AC input voltage vac, and the correction amount Kphase is equal to or more than the upper limit threshold value or less than or equal to the lower limit threshold value. Since the period in which is deviated from the ideal period and the period in which the limiter correction amount Kphase * becomes zero according to Eq. (9) is also deviated from the ideal period, the switching element 7 is in an unexpected period deviating from the ideal period. Will stop the switching of. Therefore, in the second embodiment, when the phase difference Φ is small, the value of the upper limit threshold value is set to a larger value so that the period for stopping the switching of the switching element 7 becomes shorter. As a result, even if a phase advance or a phase delay occurs in the correction amount Kphase, it is possible to prevent the switching from being stopped in a period deviating from the ideal period.

For example, when the phase difference Φ is 10 degrees, the upper limit threshold value is 2.5 as shown in FIG. When the limiter correction amount Kphase * is calculated based on this and the period for stopping the switching of the switching element 7 is obtained, the period when the input voltage phase θ is 0 to 10 degrees or 174 to 180 degrees is the period for stopping the switching. Become. Therefore, if the phase advance is 4 degrees or less, the period for stopping the switching is within the above-mentioned ideal period.

When the phase difference Φ is large, as shown in FIG. 6B, the current Ic * is larger than the current I *, so that the current flowing through the converter 4 is smaller than the current flowing through the input capacitors 6 and 6a of the input filter 21. , The output power is low. In such a state, the switching of the switching element 7 can be stopped for a long period of time, and the distortion of the input current iac generated when the switching is stopped is small. Therefore, the upper limit threshold value is set to a small value and the lower limit threshold value is set to a large value. Is set to, and the period for stopping switching is lengthened.
Others are the same as those in the first embodiment, and the description thereof will be omitted.

According to the second embodiment, the same effect as that of the first embodiment can be obtained. In addition, when the phase difference between the AC input current and the AC input voltage is small, the upper limit threshold for obtaining the limiter correction amount is increased, and the period for stopping the switching of the switching element is set shorter, so that the detection error in the input voltage phase Therefore, even if the correction amount has a phase lead or a phase lag, it is possible to prevent the switching from being stopped in an unexpected period. Therefore, it is possible to prevent the occurrence of distortion of the AC input current and the decrease in the power factor due to this distortion. Further, when the phase difference between the AC input current and the AC input voltage is large, the upper limit threshold value is made smaller and the lower limit threshold value is made larger, and the period for stopping the switching of the switching element is set longer, so that the switching loss is further reduced. Can be done.

The upper limit threshold value and the lower limit threshold value shown in FIG. 11 are examples, and the upper limit threshold value and the lower limit threshold value may be changed by the phase difference Φ. For example, when the phase difference Φ is small, it is conceivable to set the upper limit threshold value small. However, it is desirable to set as shown in FIG.

Embodiment 3.
Hereinafter, the third embodiment will be described with reference to FIGS. 12A to 21. The third embodiment is different from the first embodiment in the method of determining the correction amount Kphase according to the input voltage phase θ (= vinphase). That is, in the first embodiment, the input voltage phase θ (= vinphase) obtained from the input voltage detection value vinsen is used to determine the correction amount Kphase based on the equation (8), but in the third embodiment, the switch is used. The control unit 13 approximates the calculation of the equation (8) to determine the correction amount Kphase.

First, consider converting Eq. (8) into an approximate expression. With the input voltage phase θ as a frequency display (the period of pulsation voltage | vac | is 0 to 180 degrees), in FIG. 12A, the ideal AC input current waveform with a force factor of "1", that is, the AC input voltage vac. The waveform of the ideal AC input current iacref having the same phase as the above is shown by a solid line, and the waveform of the phase-advancing AC input current iaclead in a state where the phase is advanced by 10 degrees with respect to the ideal AC input current iacref is shown by a broken line. Further, in FIG. 12B, the waveform of the ideal AC input current iacref is shown by a solid line, and the waveform of the phase-delayed AC input current iaclague in a state where the phase is delayed by 10 degrees with respect to the ideal AC input current iacref is shown by a broken line.

FIG. 13A shows the limiter correction amount Kphase * obtained by dividing the phase-advancing AC input current iaclead in FIG. 12A by the ideal AC input current iacref based on the equation (8) and performing the limiter processing according to the equation (9). .. Further, in FIG. 13B, the phase-delayed AC input current iaclag in FIG. 12B is divided by the ideal AC input current iacref based on the equation (8), and the limiter correction amount Kphase * obtained by the limiter processing according to the equation (9) is obtained. show. In FIGS. 13A and 13B, the upper limit threshold value and the lower limit threshold value for obtaining the limiter correction amount Kphase * are set to 2 and 0, respectively.

When the AC input current iac is phase-advanced with respect to the AC input voltage vac, that is, when the denominator of Eq. (5) is "positive", correction is performed so as to delay the phase-advancing component. As shown, the limiter correction amount Kphase * increases as the input voltage phase θ increases. On the other hand, when the AC input current iac is in slow phase with respect to the AC input voltage vac, that is, when the denominator of Eq. (5) is "negative", correction is performed so as to advance the slow phase component. As shown in 13B, the limiter correction amount Kphase * decreases as the input voltage phase θ increases.
In the following description, in consideration of the influence of the input capacitors 6 and 6a, the case where the AC input current iac is in phase with respect to the AC input voltage vac (that is, the state of FIG. 12A) is described.

Consider the case where the phase difference Φ of the AC input current iac with respect to the AC input voltage vac changes due to the output power or the like. When the phase difference Φ of the AC input current iac with respect to the AC input voltage vac is changed to "5 degrees", "10 degrees", "15 degrees", "20 degrees", and "25 degrees", respectively, (8) ( The result of obtaining the limiter correction amount Kphase * for the input voltage phase θ based on the equation 9) is shown by a solid line in FIG. Further, the limiter correction amount Kphase * when each limiter correction amount Kphase * shown by the solid line is approximated to a simple linear function is shown by a broken line. In FIG. 14, the upper limit threshold value 2 and the lower limit threshold value for obtaining the limiter correction amount Kphase * are set to 0. Since the approximated limiter correction amount Kphase * is one in which each linear function shown by the broken line in FIG. 14 is periodically repeated, the approximated limiter correction amount Kphase * is a saw having a period of 180 degrees. It becomes a wavy signal. Such an approximation is to approximate the equation (8) to a linear function equation in which the correction amount Kphase simply increases according to the input voltage phase θ as shown in the following equation (10). The time required for the calculation of the limiter correction amount Kphase * and the calculation of the limiter correction amount Kphase * is shortened.

Kphase = A · θ + B (10)

With reference to FIG. 14, the slope A and the intercept B of the linear function equation shown in the equation (10) can be set under the following conditions.
(I) Set the slope A to a "positive" value.
(Ii) The larger the phase difference Φ of the AC input current iac with respect to the AC input voltage vac, the larger the change in the correction amount Kphase with respect to the change in the input voltage phase θ, that is, the larger the inclination A is set.
(Iii) As a method of determining the intercept B, the intercept B is set so that the correction amount Kphase is “1” when the input voltage phase θ is 90 degrees according to FIG. That is, the intercept B is determined so that B = 1-A × (90 degrees).

Regarding (ii) above, the condition that the phase difference Φ of the AC input current iac with respect to the AC input voltage vac becomes large will be considered from the equation (7).
The constants used in equation (7) are the AC input voltage vac = 200V (50Hz: the frequency of the pulsating voltage is 100Hz), the output voltage vo = 400V, the load R = 2938Ω (power 54.45W), and the input filter 21. Based on the reactor L of the input coil 20 = 7 mH and the combined capacity C = 0.42uF of the input capacitors 6 and 6a of the input filter 21, the output power Po to the load 8 of the power supply main circuit unit 2, the AC input voltage vac, and the AC FIGS. 15A to 15C show the calculation results of the phase difference Φ of the AC input current iac with respect to the AC input voltage vac when the input voltage frequency f is used as a parameter and they are changed.

It can be seen from FIG. 15A that the smaller the output power Po, the larger the phase difference Φ, and that the larger the AC input voltage vac, the larger the phase difference Φ. Further, it can be seen from FIG. 15C that the larger the AC input voltage frequency f, the larger the phase difference Φ. In this way, the phase difference Φ increases with changes such as the output power Po to the load 8 decreases, the AC input voltage vac increases, or the AC input voltage frequency f increases. Therefore, the input voltage increases accordingly. The change in the correction amount Kphase with respect to the change in the phase θ is set to be large, that is, the inclination A in the equation (10) is set to be large.

The correction amount Kphase may be determined by reading the values of the slope A and the intercept B determined in advance for each combination of various parameters from the table memory, and the correction amount Kphase calculated in advance for each combination of various parameters. The value may be read from the table memory. As another method, for example, a means capable of detecting one or both of a configuration for detecting the power factor and a configuration for detecting the AC input current iac is added to the configuration of FIG. 1, and the power factor is improved or the AC input is input. By controlling the slope A and the section B to be changed every cycle of the pulsating voltage | vac | so that the current strain is reduced, these values A and B are adjusted so that the power factor is further improved. can do.
Similar to the first embodiment, the limiter correction amount Kphase * is subjected to the limiter processing as shown in the equation (9) after the calculation of the correction amount Kphase, and is used for deriving the on-time ton2 of the element drive signal Vg.

As for the third embodiment, the effect was confirmed by the simulation using the circuit simulation of Myway Plus Co., Ltd. as in the case of the first embodiment. The simulation conditions are AC input voltage vac = 242V (50Hz: pulsating voltage | vac | frequency is 100Hz), output voltage vo = 390V to load 8, reactance L = 7mH of input coil 20 of input filter 21, input filter. The combined capacitance C = 0.42uF of the input capacitors 6 and 6a of 21; the upper limit threshold when obtaining the limiter correction amount Kphase * was set to 2, and the lower limit threshold was set to 0. The output power Po to the load 8 was confirmed in two ways: (i) Po = 50W (load resistance R = 3042Ω) and (ii) Po = 5W (load resistance R = 30kΩ).

(I) When the output power Po = 50 W The simulation result of the reference example to which neither the third embodiment nor the first embodiment is applied is shown in FIG. 16, the simulation result of the third embodiment is shown in FIG. 17, and the first embodiment. The simulation result of is shown in FIG. In FIGS. 16, 17, and 18, the upper row shows the limiter correction amount Kphase * according to the input voltage phase θ, the middle row shows the element drive signal Vg, and the lower row shows the waveforms of the AC input voltage vac and the AC input current iac. Shows. In FIG. 16, the value of the limiter correction amount Kphase * is fixed at 1 for convenience of comparison with FIGS. 17 and 18. Further, in FIG. 17, the slope and intercept of the equation (10) are determined so that the value of the limiter correction amount Kphase * becomes 1 when the input voltage phase θ is 90 degrees. Further, in FIGS. 16, 17, and 18, the waveforms of the AC input voltage vac and the AC input current iac in the lowermost stage are for comparing the phase difference between the two, and the AC input current is for easy comparison. iac displays a value multiplied by 1000.

When the power factor is confirmed by circuit simulation, it is 0.971 in the reference example shown in FIG. 16, 0.995 in the third embodiment shown in FIG. 17, and 0.993 in the first embodiment shown in FIG. By using the approximate expression according to the equation (10) of the embodiment 3, there is a greater effect of improving the power factor than that of the reference example, and it is possible to obtain almost the same effect as the case of the equation (8) shown in the first embodiment. You can check it. Under the condition of output power Po = 50W, the correction amount Kphase approximately calculated by the equation (10) does not reach the upper limit threshold value and the lower limit threshold value for obtaining the limiter correction amount, so that the switching element 7 Is switching in all areas.

(Ii) When the output power Po = 5W The simulation result of the reference example to which neither the third embodiment nor the first embodiment is applied is shown in FIG. 19, the simulation result of the third embodiment is shown in FIG. 20, and the first embodiment. The simulation result of is shown in FIG. In FIGS. 19, 20, and 21, the upper row shows the limiter correction amount Kphase * according to the input voltage phase θ, the middle row shows the element drive signal Vg, and the lower row shows the waveforms of the AC input voltage vac and the AC input current iac. Shows. In FIG. 19, the value of the limiter correction amount Kphase * is fixed at 1 for convenience of comparison with FIGS. 20 and 21. Further, in FIG. 20, the slope and intercept of Eq. (10) are determined so that the value of the limiter correction amount Kphase * becomes 1 when the input voltage phase θ is 90 degrees (advance 25 degrees in FIG. 14). State). Further, in FIGS. 19, 20, and 21, the waveforms of the AC input voltage vac and the AC input current iac at the bottom are for comparing the phase difference between the two, and the AC input current is for easy comparison. iac displays a value multiplied by 6000.

When the power factor is confirmed by circuit simulation, it is 0.676 in the reference example shown in FIG. 19, 0.822 in the third embodiment shown in FIG. 20, and 0.834 in the first embodiment shown in FIG. By using the approximate expression according to the equation (10) of the embodiment 3, the power factor improving effect is larger than that of the reference example, and the same level of effect as that of the equation (8) shown in the first embodiment can be obtained. Can be confirmed.

In the third embodiment shown in FIG. 20, there is a period in which the limiter correction amount Kphase * is 0 and the period in which the element drive signal Vg is fixed to 0 is the same as in the first embodiment shown in FIG. You can check. From this, it can be seen that even when the equation (10) of the third embodiment is used, the increase in the switching loss is suppressed by stopping the switching of the switching element 7.
As described above, from the simulation results (FIGS. 17 and 20) when the output power Po is 50 W and 5 W, the power factor even when a simple linear function equation based on the equation (10) of the third embodiment is used. It is confirmed that improvement is possible. Further, when the output power Po is 5 W, it is confirmed that the power factor can be improved while suppressing the increase in the switching loss as in the first embodiment.

According to the third embodiment, the same effect as that of the first embodiment can be obtained. Further, since the approximate expression of the linear function is used for calculating the correction amount for determining the on time of the element drive signal, it is easy to calculate the correction amount and the limiter correction amount obtained by applying the limiter processing to the correction amount. Therefore, it is not necessary to perform the sin calculation or division required in the first embodiment, and it is easy to implement a calculation processing unit such as a microcomputer.

Embodiment 4.
Hereinafter, the fourth embodiment will be described with reference to FIG. 22. FIG. 22 is a block diagram showing the configuration of the LED lighting device according to the fourth embodiment, and the components corresponding to those in FIG. 1 are designated by the same reference numerals. The LED lighting device 100 is replaced with an LED module 22 in which a load 8 vertically connects a plurality of LEDs 23 with respect to the configuration of the power factor compensating power supply device 1 shown in FIG. Therefore, in the power supply control unit 3, an output current detection unit 24 is provided in place of the output voltage detection unit 9, and an output current control unit 26 is provided in place of the output voltage control unit 11, and the target current value ioref is set. It has been changed to be. Other configurations are the same as those of the power factor compensating power supply device 1 shown in FIG.

For example, the output current detection unit 24 installs a current detection resistor (not shown) and detects the potential difference generated between both ends of the current detection resistor as a voltage conversion value iosen corresponding to the output current flowing through the LED module 22. Further, in the connection configuration of the LED module 22, all the LEDs 23 are connected in series here, but a plurality of LEDs 23 may be connected in series or in series or parallel. Further, although the light source is LED23 here, the light source is not limited to this, and can be changed to an organic EL or a laser diode.

In the fourth embodiment, considering that the current control is usually suitable for the LED 23 due to its VI characteristics, the output voltage control applied to the first embodiment is replaced with the output current control. With this configuration, the current flowing through the LED module 22 can be controlled by performing the same control as in the first to third embodiments. Further, when the dimming function is installed and the amount of light is adjusted, the dimming function can be realized by making the target current value ioref input from the outside variable.

When the LED module 22 is used as a load, the output voltage v does not change even when the current flowing through the LED module 22 is changed by the dimming function. In the first and third embodiments, the output power Po is one parameter for determining the correction amount Kphase. Therefore, ideally, it is necessary to detect both the output voltage vo and the output current io to calculate the power. rice field. On the other hand, in the fourth embodiment, since the load 8 is the LED module 22 as shown in FIG. 22, the correction amount Kphase is obtained by detecting the output current io as the voltage conversion value iosen by the output current detection unit 24. And the limiter correction amount Kphase * can be adjusted, and it is not necessary to detect the output voltage vo. The correction amount Kphase and the limiter correction amount Kphase * may be calculated in the same manner as in the cases of the first to third embodiments.

According to the fourth embodiment, in the LED lighting device to which the LED module is applied as a load, the AC input current and the AC input voltage can have the same phase and the same waveform, and the power factor can be improved. More specifically, an output current detection unit is provided in the power supply main circuit unit to detect the output current as a voltage conversion value, and an output current control unit that calculates the reference on time from the output current detection value and the target current value. Since the switching element is controlled on / off using the on-time provided in the power supply control unit and the correction amount according to the input voltage phase multiplied by the reference on-time, the phase lead of the AC input current with respect to the AC input voltage by the input capacitor The effect can be corrected. Therefore, the AC input current and the AC input voltage can be made to have the same phase and the same waveform, and the force factor can be improved. Further, as in the first to third embodiments, the switching is stopped during the period when the contribution to the power factor improvement is small, so that the increase in the switching loss is suppressed without significantly affecting the power factor improvement.

Further, since the LED module is used as a load, the output current can be detected as a voltage conversion value by the output current detection unit, and the correction amount can be adjusted by using this voltage conversion value, and it is necessary to detect the output voltage vo. No. Therefore, the advantages of reducing the number of pins of the microcomputer, shortening the processing time, reducing the wiring, and reducing the size and cost due to these can be obtained.
Further, since the dimming signal is usually input to the microcomputer from the outside to perform dimming control, even when the dimming signal is variable, the optimum correction amount Kphase after dimming can be obtained in advance. Therefore, it is possible to perform power factor improvement control with a high response speed in response to load fluctuations.

Embodiment 5.
Hereinafter, the fifth embodiment will be described with reference to FIG. 23. FIG. 23 is a block diagram showing the configuration of the LED lighting device according to the fifth embodiment, and the components corresponding to those in FIGS. 1 and 22 are designated by the same reference numerals. In the LED lighting device 101, for the configuration shown in FIG. 1, in addition to the load 8 being the LED module 22, the output current adjustment is performed between the LED module 22 and the converter 4 in the power supply main circuit unit 2. The LED current adjustment circuit 41 as a DC / DC converter, that is, a second converter is provided and a two-stage converter is applied, and an output that detects the voltage conversion value iosen corresponding to the current flowing through the LED module 22. A current detection unit 24 is provided. Further, in the power supply control unit 3, an output current control unit 26 is provided in addition to the output voltage control unit 11.

According to the fifth embodiment, the same effect as that of the fourth embodiment can be obtained. Further, since the two-stage converter is applied, the power factor improvement control on the input side and the current stability control on the output side can be realized at the same time. That is, it is possible to remove the commercial frequency ripple of the AC power supply on the output side while controlling the power factor improvement on the input side, and prevent the occurrence of light flicker in the LED module.

In the above embodiments 1 to 5, in order to reduce the size of the power factor compensation power supply device or the LED lighting device, all or part of the device is mounted in one integrated circuit and housed in one package. It may be an IC. For example, it is preferable to house the power supply control unit in one control IC package.

Although the present application describes various exemplary embodiments and examples, the various features, embodiments, and functions described in one or more embodiments are applications of a particular embodiment. It is not limited to, but can be applied to embodiments alone or in various combinations.
Therefore, innumerable variations not exemplified are envisioned within the scope of the techniques disclosed in the present application. For example, it is assumed that at least one component is modified, added or omitted, and further, at least one component is extracted and combined with the components of other embodiments.

1 Power factor compensating power supply, 2 Power supply main circuit, 3 Power supply control, 4 Converter, 5 AC power supply, 6 Input capacitor, 7 Switching element, 8 Load, 9 Output voltage detector, 10 Input voltage detector, 11 Output Voltage control unit, 12 input voltage phase detection unit, 13 switch control unit, 14 full-wave rectifier circuit, 15 output capacitor, 22 LED module, 24 output current detection unit, 26 output current control unit, 41 LED current adjustment circuit, 100, 101 LED lighting device, vac AC input voltage, iac AC input current, vinsen input voltage detection value, vo output voltage, vosen output voltage detection value, voref target voltage value, io output current, iosen voltage conversion value, ioref target current value, ton1 reference on time, ton2 on time, Kphase * limiter correction amount

Claims (5)

A power supply main circuit unit and a power supply control unit that controls the power supply main circuit unit are provided.
The power supply main circuit section is
A full-wave rectifier circuit that full-wave rectifies the AC voltage supplied from the AC power supply,
A converter having a switching element, the AC voltage fully wave rectified by the full wave rectifier circuit is input as an input voltage, and the input voltage is converted into a target output voltage.
An input capacitor that suppresses the noise generated by switching of the switching element from being conducted to the AC power supply,
An input voltage detector that detects the input voltage and
It has an output voltage detection unit that detects the output voltage.
The power supply control unit
The reference on time of the switching element is determined based on the output voltage, and the reference on time is determined.
The reference on-time is corrected by the phase difference between the input current from the AC power supply and the input voltage and the correction amount calculated based on the phase of the input voltage to calculate the on-time of the switching element.
A power supply control unit that controls the switching element based on the on-time.
The power supply control unit compares the correction amount with the predetermined upper limit threshold value and lower limit threshold value, and sets the on-time when the correction amount is equal to or more than the upper limit threshold value and when the correction amount is equal to or less than the lower limit threshold value. A power factor compensating power supply device characterized by stopping the switching of the switching element as zero. The power factor compensating power supply device according to claim 1, wherein the upper limit threshold value and the lower limit threshold value are each determined based on the phase difference. It is provided with a power supply main circuit unit to which an LED module is connected on the output side and a power supply control unit that controls the power supply main circuit unit.
The power supply main circuit section is
A full-wave rectifier circuit that full-wave rectifies the AC voltage supplied from the AC power supply,
A converter having a switching element and converting an input current obtained by full-wave rectification of the AC voltage by the full-wave rectifier circuit into a target output current.
An input capacitor that suppresses the noise generated by switching of the switching element from being conducted to the AC power supply,
An input voltage detection unit that detects an input voltage obtained by the full-wave rectifier circuit performing full-wave rectification of the AC voltage.
It has an output current detection unit that detects the output current, and has an output current detection unit.
The power supply control unit
The reference on time of the switching element is determined based on the output current, and the reference on time is determined.
The reference on-time is corrected by the phase difference between the input current from the AC power supply and the input voltage and the correction amount calculated based on the phase of the input voltage to calculate the on-time of the switching element.
A power supply control unit that controls the switching element based on the on-time.
The power supply control unit compares the correction amount with the predetermined upper limit threshold value and lower limit threshold value, and sets the on-time when the correction amount is equal to or more than the upper limit threshold value and when the correction amount is equal to or less than the lower limit threshold value. An LED lighting device characterized by stopping the switching of the switching element as zero. The LED lighting device according to claim 3, wherein the upper limit threshold value and the lower limit threshold value are each determined based on the phase difference. The LED lighting device according to claim 3 or 4, wherein the power supply main circuit unit has a second converter for adjusting the output current connected between the converter and the LED module.

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