JP6711170B2 - Motor controller - Google Patents

Description

The present invention relates to a motor control device.

Generally, a motor control device that drives and controls a motor by vector control generates a d-axis current command value and a q-axis current command value according to a motor speed command value (target speed), and then outputs the d-axis current command value. And the q-axis current command value is converted into a d-axis voltage command value and a q-axis voltage command value. Further, the d-axis voltage command value and the q-axis voltage command value are converted into a three-phase voltage, and the motor is driven by the output according to the three-phase voltage. Here, the speed (rotation speed) of the motor and the rotation speed are equivalent.

Further, the motor control device detects a three-phase current of the motor (current flowing in the motor) as control for the motor to reach the speed command value, and the three-phase current is used as the d-axis current of the two-phase rotating coordinate system. It is converted into a q-axis current, the actual speed of the motor is calculated based on the d-axis current and the q-axis current, and feedback control is performed so that the difference between the actual speed and the speed command value becomes zero.

Further, the motor control device controls the d-axis current and the q-axis current, and the d-axis current command value and the q-axis current obtained based on the difference between the actual speed and the speed command value, as control for the motor to reach the speed command value. Feedback control is performed so that the difference between the respective axis current command values becomes zero.

As described above, the motor control device that drives the motor by the vector control has the two-system feedback loop, and the drive accuracy of the motor depends on the detection accuracy of the three-phase current, so that the three-phase current is accurately detected. Is required.

The motor control device performs A/D (Analog/Digital) conversion for converting the three-phase current from an analog value to a digital value in order to generate a signal necessary for motor control from the detected three-phase current. In order to realize the high precision and high responsiveness of this A/D conversion, a motor control device provided with a ΔΣ modulation type A/D converter has been proposed (for example, refer to Patent Document 1).

JP, 2013-198229, A

Here, in the above-mentioned Patent Document 1, even if three-phase current is detected with high accuracy and A/D conversion is performed, conversion to d-axis current and q-axis current, or three-phase from d-axis voltage and q-axis voltage. It is not possible to compensate for the conversion accuracy when converting to voltage. It is possible to add a control for compensating the conversion accuracy to these conversions, but there are problems such as an increase in the amount of calculation related to the conversion process and a decrease in responsiveness due to the increase in the amount of calculation. ..

An example of the embodiment of the present application has been made in view of the above, and an object thereof is to provide, for example, a motor control device capable of further reducing an error caused by current or voltage conversion performed in motor control.

In an example of the technology disclosed in the present application, for example, a motor control device includes a first ΔΣ unit, a second ΔΣ unit, a drive unit, and a detection unit. The first ΔΣ unit performs a ΔΣ calculation on the motor speed command value and outputs it. The second ΔΣ unit performs a ΔΣ operation on the output of the first ΔΣ unit and outputs it. The drive unit applies a voltage obtained by quantizing the output of the second ΔΣ unit to the motor to drive it. The detector detects the motor current. Then, the first ΔΣ unit subtracts the actual speed of the motor estimated from the motor current detected by the detection unit from the motor rotation speed command value, and integrates and outputs the integrated value to the second ΔΣ unit. The second ΔΣ unit subtracts the d-axis current and the q-axis current obtained based on the motor current detected by the detection unit from the output of the first ΔΣ unit, integrates them, and outputs the integrated result to the drive unit.

According to an example of the technology disclosed in the present application, for example, an error caused by conversion of current or voltage performed in motor control can be further reduced. In addition, an increase in the amount of calculation related to the conversion process and a decrease in responsiveness due to the increase in the amount of calculation will not occur.

FIG. 1 is a diagram showing a basic configuration example of a conventional motor control device that performs vector control. FIG. 2 is a diagram showing elements that can be replaced with a quantizer in a conventional motor control device. FIG. 3 is a diagram showing replacement of the position estimator in the conventional motor control device. FIG. 4 is a diagram showing a configuration in which elements are replaced in the conventional motor control device. FIG. 5 is a diagram showing a configuration model of a second-order ΔΣ modulation type A/D converter. FIG. 6 is a diagram showing a configuration of a secondary ΔΣ modulation type motor control device according to the first embodiment. FIG. 7 is a diagram showing a configuration model of a secondary ΔΣ modulation type A/D converter when the motor is regarded as a frequency converter. FIG. 8 is a diagram showing a configuration model of a second-order ΔΣ modulation type A/D converter that copes with an error due to load torque fluctuation. FIG. 9A is a diagram showing a configuration of a secondary ΔΣ modulation type motor control device according to the second embodiment. FIG. 9B is a diagram showing the configuration of the current corrector of the second-order ΔΣ modulation type motor control device according to the second embodiment.

Embodiments of a motor control device according to the present application will be described below with reference to the accompanying drawings. The following embodiments relate to a motor control device such as an air conditioner or a cryogenic storage device that performs torque control of a motor that drives a compressor having periodic load torque fluctuations by position sensorless vector control. However, the disclosed technique is widely applicable to motor control devices that perform position sensorless vector control.

The embodiments described below are merely examples and do not limit the disclosed technology. Further, the embodiments and the modifications thereof described below can be appropriately combined within a range that does not contradict. In addition, in the embodiments and modifications thereof described below, configurations and processes according to the disclosed technology are mainly shown, and description of the other configurations and processes is simplified or omitted. Further, in the embodiments and the modifications thereof described below, the same configurations and processes are denoted by the same reference numerals, and the description of the configurations and processes already described is omitted.

[Embodiment 1]
First, the background and outline of the first embodiment will be described. FIG. 1 is a diagram showing a basic configuration example of a conventional motor control device that performs vector control. The motor controller 100X includes a subtractor 11, a speed controller 12, an exciting current controller 13, a subtractor 14, a subtractor 15, a d-axis current controller 16, a q-axis current controller 17, a decoupling controller 18, A subtractor 19, an adder 20, a dq/3φ converter 21, a PWM (Pulse Width Modulation) generator 22, an IPM (Intelligent Power Module) 23, a resistor R constituting a shunt current detector, a 3φ current calculator 24, It has a 3φ/dq converter 25, an axis error calculation processing unit 26, a PLL controller 29, a position estimator 30, and a 1/Pn processor 31.

The subtractor 11 is the estimated current angular velocity output from the 1/Pn processor 31 from the velocity command value (target velocity, here, the mechanical angular target velocity) ωm ^* input to the motor control device 100X. The speed deviation (mechanical angular speed deviation) Δω obtained by subtracting the speed (actual mechanical angular speed) ωm is output to the speed controller 12.

The speed controller 12 generates a q-axis current command value Iq ^* that reduces the speed deviation Δω output from the subtractor 11, and outputs the q-axis current command value Iq ^* to the exciting current controller 13 and the subtractor 15. The excitation current controller 13 generates a d-axis current command value Id ^* from the q-axis current command value Iq ^* output from the speed controller 12, and outputs it to the subtractor 14. Here, the speed controller 12 and the exciting current controller 13 are also referred to as a current command value generation unit. Further, the d-axis and the q-axis represent coordinate axes of a two-phase rotating coordinate system, and Id, Iq and Vd, Vq described later are current and voltage on this coordinate axis.

The subtractor 14 subtracts the d-axis current Id output by the 3φ/dq converter 25 from the d-axis current command value Id ^* output from the excitation current controller 13 to generate a d-axis current deviation ΔId. Output to the axis current controller 16. The subtractor 15 subtracts the q-axis current Iq output from the 3φ/dq converter 25 from the q-axis current command value Iq ^* output from the speed controller 12 to generate a q-axis current deviation ΔIq to generate the q-axis. Output to the current controller 17.

The d-axis current controller 16 generates a d-axis voltage command value Vd ^** from the d-axis current deviation ΔId output from the subtractor 14. The q-axis current controller 17 generates a q-axis voltage command value Vq ^** from the q-axis current deviation ΔIq output from the subtractor 15.

The decoupling controller 18 cancels the interference of the d-axis and the q-axis and generates a decoupling correction value for controlling each independently. Specifically, for decoupling the d-axis voltage command value Vd ^** from the d-axis current Id output from the 3φ/dq converter 25 and the estimated electrical angle ωe output from the PLL controller 29. The d-axis decoupling correction value Vda is generated and output to the subtractor 19. The decoupling controller 18 determines the q-axis voltage command value Vq ^** from the q-axis current Iq output from the 3φ/dq converter 25 and the estimated electrical angle ωe output from the PLL controller 29. A q-axis decoupling correction value Vqa for interfering is generated and output to the adder 20.

The subtractor 19 subtracts the d-axis decoupling correction value Vda output from the decoupling controller 18 from the d-axis voltage command value Vd ^** output from the d-axis current controller 16 to obtain the d-axis voltage. The d-axis voltage command value Vd ^* is generated by decoupling the command value Vd ^** and output to the dq/3φ converter 21. The adder 20 adds the q-axis decoupling correction value Vqa output from the decoupling controller 18 to the q-axis voltage command value Vq ^** output from the q-axis current controller 17, and adds the q-axis voltage. A q-axis voltage command value Vq ^* is generated by decoupling the command value Vq ^** and output to the dq/3φ converter 21.

The dq/3φ converter 21 uses the electrical angle phase (dq axis phase) θe, which is the current rotor position output by the position estimator 30, to decoupling the two-phase d axis voltage command value Vd. ^{The *} and q-axis voltage command values Vq ^* are converted into U-phase output voltage command values Vu ^* , V-phase output voltage command values Vv ^* , and W-phase output voltage command values Vw ^{* of} three phases. Then, the dq/3φ converter 21 outputs the U-phase output voltage command value Vu ^* , the V-phase output voltage command value Vv ^* , and the W-phase output voltage command value Vw ^* to the PWM generator 22. Note that Vu ^* , Vv ^* , Vw ^*, and Iu, Iv, Iw, which will be described later, are voltages and currents in a three-phase fixed coordinate system.

The PWM generator 22 generates a 6-phase PWM signal from the U-phase output voltage command value Vu ^* , the V-phase output voltage command value Vv ^* , the W-phase output voltage command value Vw ^*, and the PWM carrier signal, and sends it to the IPM 23. Output.

The IPM 23 converts the AC voltage applied to each of the U phase, V phase, and W phase of the motor M from the DC voltage Vdc supplied from the outside, based on the 6-phase PWM signal output from the PWM generator 22. Then, the respective AC voltages are applied to the U phase, V phase, and W phase of the motor M.

The 3φ current calculator 24 calculates the U-phase current Iu and the V-phase current Iv of the motor M from the 6-phase PWM switching information output from the PWM generator 22 and the bus current detected by the resistor R by the 1-shunt current detection method. , W-phase current Iw is calculated. Alternatively, for the U-phase current Iu, V-phase current Iv, and W-phase current Iw of the motor M, two CTs (Current Transformers) detect the U-phase current Iu and V-phase current Iv, and the remaining W-phase current Iw is The 2CT method may be used, which is calculated from the relational expression of Iu+Iv+Iw=0. The 3φ current calculator 24 outputs the calculated U-phase current Iu, V-phase current Iv, and W-phase current Iw of the motor M to the 3φ/dq converter 25.

The 3φ/dq converter 25 uses the electrical angle phase θe output from the position estimator 30 to output the three-phase U-phase current Iu, V-phase current Iv, and W-phase current Iw output from the 3φ current calculator 24. Is converted into a two-phase d-axis current Id and a q-axis current Iq. Then, the 3φ/dq converter 25 transfers the d-axis current Id to the subtractor 14, the decoupling controller 18, the axis error calculation processing unit 26, and the q-axis current Iq to the subtractor 15, the decoupling controller 18, It is output to the axis error calculation processing unit 26, respectively.

The axis error calculation processing unit 26 outputs the d-axis voltage command value Vd ^* output from the subtractor 19 and the q-axis voltage command value Vq ^* output from the adder 20 and the d-axis output from the 3φ/dq converter 25. The axis error variation Δθ is calculated from the current Id and the q-axis current Iq and output to the PLL controller 29. Here, the axis error is a deviation between the actual dq axis and the control dq axis.

The PLL controller 29 calculates an electrical angle estimated speed ωe, which is the estimated angular velocity of the current rotation of the motor, from the axis error variation Δθ output from the axis error calculation processing unit 26, and the decoupling controller 18, It outputs to the position estimator 30 and the 1/Pn processor 31, respectively.

The position estimator 30 calculates an electrical angle phase (dq axis phase) θe for estimating the rotor position from the electrical angle estimation speed ωe output from the PLL controller 29. Then, the position estimator 30 outputs the electrical angle phase θe to the dq/3φ converter 21 and the 3φ/dq converter 25, respectively.

The 1/Pn processing unit 31 calculates the actual angular velocity (mechanical angular velocity) that is the estimated current angular velocity calculated by dividing the estimated electrical angle velocity ωe output from the PLL controller 29 by the pole pair number Pn of the motor M. ) Ωm is output to the subtractor 11.

Here, the motor control device 100X shown in FIG. 1 has three features. The first feature is that it has two feedback paths, a first feedback (FB) path FB1 and a second feedback path FB2 shown in FIG. 1, and each feedback path has its own command value (target value). ) Is subtracted from each detected value (estimated value). In the embodiment of the present invention, the first feedback path FB1 is the feedback path of the actual speed ωm, and the second feedback path FB2 is the feedback path of the q-axis current Iq and the d-axis current Id. The second feature is that each converter has a dq/3φ converter 21, a 3φ/dq converter 25, a PWM generator 22, and a 3φ current calculator 24. The third feature is that the rotation phase (electrical angle phase θe) of the motor M by the position estimator 30 is feed-forwarded (FF: Feed Forward in FIG. 1) to the dq/3φ converter 21 and the 3φ/dq converter. 25 is being fed back (third feedback path FB3 in FIG. 1).

Focusing on the second feature, as shown in FIG. 2, the path from the d-axis current controller 16 and the q-axis current controller 17 to the 3φ/dq converter 25 in FIG. Can be considered and replaced. FIG. 2 is a diagram showing elements that can be replaced with a quantizer in a conventional motor control device. As a result, the input/output of the quantizer Q becomes the d-axis current Id and the q-axis current Iq.

In the third feature, as shown in (a) of FIG. 3, since the output from the position estimator 30 is fed back to the 3φ/dq converter 25, as shown in (b) of FIG. The position estimator 30 can be replaced with the delay unit 32 and the delay unit 33 by regarding the process of the position estimator 30 as a process with one carrier delay. The delay device 32 is a d-axis first-order delay device, and the delay device 33 is a q-axis first-order delay device. FIG. 3 is a diagram showing replacement of the position estimator in the conventional motor control device.

Therefore, when the replacement of FIGS. 2 and 3 is applied to the configuration of FIG. 1, it can be expressed as shown in FIG. FIG. 4 is a diagram showing a configuration in which elements are replaced in the conventional motor control device.

Here, the configuration of the motor control device 100Y shown in FIG. 4 is compared with the configuration model of the secondary ΔΣ modulation type A/D converter shown in FIG. FB1 in FIG. 4 and FB1 in FIG. 5, and FB2 in FIG. 4 and FB2 in FIG. 5 are corresponding feedback paths. Further, the subtractor 11 in FIG. 4 and the subtractor S1 in FIG. 5, the subtractor 14 and the subtractor 15 in FIG. 4, and the subtractor S2 in FIG. 4A and FIG. 5A each include a quantizer Q and a delay device D.

From the above, the motor control device 100Y shown in FIG. 4 has a portion similar to the second-order ΔΣ modulation type A/D converter shown in FIG. Then, as described above, the subtractor 11, the subtractor 14, and the subtractor 15 of the motor control device 100Y shown in FIG. 4 are regarded as the subtractor S1 and the subtractor S2 of FIG. 5, respectively, and are the same as those of FIG. By adding the integrator I1 between the subtractor S1 and the subtractor S2 and the integrator I2 between the subtractor S2 and the quantizer Q, the configuration of the second-order ΔΣ modulation type A/D converter is obtained. be able to.

As described above, the quantizer Q shown in FIG. 5A includes the elements shown in FIG. FIG. 5B is a diagram in which an adder A1 for adding the quantization error Q1 is added to the elements of FIG. 2 included in the quantizer Q of FIG. 5A. This quantization error Q1 is the conversion of current and voltage in the elements from the d-axis current controller 16 and the q-axis current controller 17 to the 3φ/dq converter 25 in FIG. 2, PWM modulation, IPM driving, 1 shunt current. A conversion error caused by detection or the like is regarded as a quantization error, and this quantization error Q1 is added to the quantizer Q by the adder A1.

Here, the input/output relationship of the secondary ΔΣ modulation type A/D converter shown in FIG. 5B is shown in the following expression (1). “X”, “Y”, “Z ⁻¹ ”, and “Q1” in the following equation (1) are the same as those shown in FIG. 5B.

The quantization error Q1 in the above equation (1) represents the conversion error by the quantizer Q as described above, and the coefficient (1-Z- ¹ ) related to the quantization error Q1 becomes the attenuation coefficient (differential term). Z ⁻¹ included in this attenuation coefficient can be regarded as ¹ in the low frequency region. This is because Z=exp(jωn) (where n is the number of delay clocks, exp is an exponential function, and j is an imaginary unit). With this characteristic, the conversion error (quantization error) can be reduced in the low frequency region (driving speed region) (noise shaping characteristic).

In order to apply this characteristic to vector control, in the present invention, as shown in FIG. 6, the integrator 35 corresponding to the integrators I1 and I2 shown in FIG. The configuration is such that a device 36 and an integrator 37 are added. FIG. 6 is a diagram showing a configuration of a secondary ΔΣ modulation type motor control device according to the first embodiment. The second-order ΔΣ modulation type motor control device 100A according to the first embodiment shown in FIG. 6 has a function of reducing the conversion error by integrating the quantization error Q1 by the integrator 35, the integrator 36, and the integrator 37. Demonstrate. In other words, the integrator 35, the integrator 36, and the integrator 37 add up the difference between the input and output by the subtractor 11, the subtractor 14, and the subtractor 15 to obtain the current sampling data (data processed by the current carrier). And the sampling data before that (data processed by the carrier before the present) has the information, the reproducibility of the input signal (speed command value, target speed) is improved (the conversion error is reduced and the accuracy is improved. To).

In FIG. 6, the subtractor 11 is the first subtractor of the present invention, the integrator 35 is the first integrator of the present invention, and the first ΔΣ unit is an example of the first subtractor and the first subtractor. It has an integrator of 1. Further, the subtractors 14 and 15 are the second subtractors of the present invention, the integrators 36 and 37 are the second integrators of the present invention, and the second ΔΣ section of the present invention is an example. As the second subtractor and the second integrator. Further, the d-axis current controller 16, the q-axis current controller 17, the decoupling controller 18, the subtractor 19, the adder 20, the dq/3φ converter 21, the PWM generator 22, and the IPM 23 are the driving units of the present invention. It is an example of a part. Further, the resistance R, the 3φ current calculator 24, and the 3φ/dq converter 25 are an example of the detection unit of the present invention. The delay device 34 is an example of the first delay unit of the present invention. The delay device 32 and the delay device 33 are examples of the second delay unit of the present invention. The axis error calculation processing unit 26, the PLL controller 29, and the 1/Pn processing unit 31 are examples of the position/speed estimating unit of the present invention. Then, the first ΔΣ section is a speed deviation (mechanical angular speed) that is a difference between the input speed command value (target speed) ωm ^* and the actual speed (mechanical angular speed) ωm of the motor detected by the position/speed estimation section. Deviation) Δω is integrated and output, and the second ΔΣ unit outputs the current command value generated by the speed controller 12 and the excitation current controller 13, which are current command value generators, based on the output of the first ΔΣ unit. I ^* (d-axis current command value Id ^* , q-axis current command value Iq ^* ) and motor current I detected by the detection unit (here, from the current of the three-phase fixed coordinate system detected by the detection unit to the two-phase current) The difference ΔI (here, the d-axis current deviation ΔId and the q-axis current deviation ΔIq) between the d-axis current Id and the q-axis current Iq converted into the current of the rotating coordinate system is integrated and output to the drive unit.

The relationship between input and output in FIG. 6 is as shown in the following expression (2). In the following equation (2), Q1 is the quantization error of the element replaced by the quantizer Q in FIG. “Ω”, “ω ^* ”, “Z ⁻¹ ”, and “Q1” in the following equation (2) are the same as those shown in FIG. 5B and FIG. 6.

According to the first embodiment, the configuration of the motor control device that performs vector control is a ΔΣ modulation type, which brings about a reduction effect on a conversion error that occurs in the control system of vector control, and improves the accuracy of the entire vector control. Can be planned. For example, the motor control device 100A according to the first embodiment can reduce the quantization error Q1 of the element replaced with the quantizer Q shown in FIG. Further, since only an integrator that performs an integration process for adding the value of one carrier before to the current value is added to the configuration of the motor control device that performs vector control, an increase in the amount of calculation related to the conversion process, There is no decrease in responsiveness or increase in processing time due to the increase in the amount of calculation.

[Embodiment 2]
In the first embodiment, the quantization error Q1 of the element replaced with the quantizer Q can be reduced, but the speed fluctuation due to the load torque pulsation of the motor M driven by the motor control device cannot be reduced. That is, in the first embodiment, conversion errors such as A/D conversion error and D/A conversion error in the motor control device that executes vector control are only reduced, and speed fluctuation due to load torque pulsation of the motor is also reduced. Can not. That is, it is impossible to deal with an error (external factor) that does not exist in the control system.

That is, when a load having a periodic load torque pulsation such as a compressor is driven, the speed (rotation speed) of the motor fluctuates due to the load torque pulsation, causing noise and vibration, which is a problem. Therefore, the motor control device is required to suppress the speed fluctuation due to the load torque pulsation.

Therefore, in the second embodiment, attention is paid to the fact that the speed (rotation speed) of the motor fluctuates, the motor M is regarded as a frequency converter, and the present invention is applied to reduce the speed fluctuation due to load torque pulsation.

Here, the configuration of the motor control device 100A shown in FIG. 6 will be compared with the configuration model of the second-order ΔΣ modulation type A/D converter shown in FIG. 7. FIG. 7 is a diagram showing a configuration model of a secondary ΔΣ modulation type A/D converter when the motor is regarded as a frequency converter. 7A corresponds to FIG. 5A and is a configuration model in which the motor M is regarded as the frequency converter FC in the motor control device 100A of FIG.

The quantizer Q shown in (a) of FIG. 7 is obtained by adding the quantization error Q1 included in the quantizer Q of FIG. Further, when the frequency converter FC regards the motor M in FIG. 6 as the frequency converter FC, the frequency converter FC adds the speed fluctuation due to the load torque pulsation of the motor M as a frequency conversion error Q2 (hereinafter, also referred to as speed error Q2). It is a thing. From this, (a) of FIG. 7 can be replaced as shown in (b) of FIG. 7 ((the frequency converter FC of FIG. 7(a) can be replaced with the adder A2 of FIG. 7(b). At this time, the characteristic of the ΔΣ modulation type A/D converter can be obtained at the input/output in (b) of Fig. 7. The total conversion error of the element replaced with the quantizer Q in Fig. 6 is quantized. If the error is Q1 and the speed fluctuation (speed error) of the motor M is Q2, the relationship between the input X and the output Y in (a) of FIG. 7 is expressed by the following equation (3).

Here, in the above formula (3), since the speed error Q2 due to the load torque pulsation of the motor M does not have the damping coefficient as described in the first embodiment, the speed error Q2 due to the load torque pulsation of the motor M is reduced. I see that I can't. A characteristic of the ΔΣ modulation type A/D converter is that when an error is included in the control system like the quantization error Q1, the attenuation coefficient is involved and the error can be reduced. Therefore, as shown in FIG. 8, an adder A3 for adding the same amount as the speed error (frequency conversion error) Q2 is added in the control system. FIG. 8 is a diagram showing a configuration model of a second-order ΔΣ modulation type A/D converter that copes with an error due to load torque fluctuation. The relationship between the input X and the output Y shown in FIG. 8 is expressed by the following equation (4).

According to the above equation (4), since the speed error (frequency conversion error) Q2 is related to the damping coefficient, it is possible to reduce the speed error Q2.

A motor control device 100B according to the second embodiment to which the configuration model of FIG. 8 is applied is as shown in FIG. 9A. Compared with the conventional motor control device 100X shown in FIG. 1, the motor control device 100B has a position estimator 30B instead of the position estimator 30, and has an integrator 35 between the subtractor 11 and the speed controller 12. And an integrator 36 between the subtractor 14 and the d-axis current controller 16, an integrator 37 between the subtractor 15 and the q-axis current controller 17, and a 3φ/dq converter 25. And a current corrector 38 between the subtractors 14 and 15.

The position estimator 30B calculates an electrical angle phase (dq axis phase) θe and a mechanical angle phase θd from the electrical angle estimated speed ωe output from the PLL controller 29, and the electrical angle phase θe is converted into a dq/3φ converter 21. And the 3φ/dq converter 25 and the mechanical angle phase θd to the current corrector 38.

In FIG. 9A, the subtractor 11 is the first subtractor of the present invention, the integrator 35 is the first integrator of the present invention, and the first ΔΣ unit of the present invention is, for example, the first subtractor. And a first integrator. Further, the subtractors 14 and 15 are the second subtractors of the present invention, the integrators 36 and 37 are the second integrators of the present invention, and the second ΔΣ section of the present invention is an example. As the second subtractor and the second integrator. Further, the d-axis current controller 16, the q-axis current controller 17, the decoupling controller 18, the subtractor 19, the adder 20, the dq/3φ converter 21, the PWM generator 22, and the IPM 23 are the driving units of the present invention. It is an example of a part. Further, the resistance R, the 3φ current calculator 24, and the 3φ/dq converter 25 are an example of the detection unit of the present invention. The current corrector 38 is an example of a current corrector. The axis error calculation processing unit 26, the PLL controller 29, and the 1/Pn processing unit 31 are examples of the position/speed estimating unit of the present invention.

FIG. 9B is a diagram showing the configuration of the current corrector of the second-order ΔΣ modulation type motor control device according to the second embodiment. The current corrector 38 of the motor control device 100B includes a subtractor 38a, an adder 38b, and a correction current calculator 38c. Further, it has a delay device 39a and a delay device 39b for executing processing with a delay of one carrier.

The correction current calculator 38c generates a current correction value ΔIc from the mechanical angle phase θd output from the position estimator 30B and the speed deviation Δω (=ωm ^* −ωm) output from the subtractor 11.

Since the speed error (frequency conversion error) Q2 is added to the current feedback path, the amplitude and phase of the speed error Q2 are converted from the speed dimension to the current dimension. Therefore, the correction current amplitude Aω and the correction phase θω are calculated from the generated mechanical angle phase θd and the speed deviation Δω. In the equation (5) below, a known method using a correlation calculation can be used for the phase detection and the amplitude detection.

In the current corrector 38 shown in FIGS. 9A and 9B based on the above equation (5), the correction current calculator 38c outputs the mechanical angle phase θd output from the position estimator 30B and the subtractor 11 outputs. The current correction value ΔIc is generated from the speed deviation (mechanical angular speed deviation) Δω (=ωm ^* −ωm). The subtractor 38 a outputs a corrected d-axis current Id′, which is the result of subtracting the current correction value ΔIc from the d-axis current Id output from the 3φ/dq converter 25, to the subtractor 14. The adder 38 b outputs a corrected q-axis current Iq′, which is the result of adding the current correction value ΔIc to the q-axis current Iq output from the 3φ/dq converter 25, to the subtractor 15.

The speed error Q2 cannot be detected from the d-axis current Id and the q-axis current Iq output from the detector (the d-axis current Id and the q-axis current Iq are currents at the actual speed, and the speed error cannot be recognized). Therefore, in the second embodiment, as the amount representing the speed error Q2, the current corrector 38 generates the current correction value ΔIc based on the speed deviation Δω which is the difference between the speed command value ωm ^* and the actual speed ωm. It is subtracted from the axis current Id and the q-axis current Iq and fed back to the second ΔΣ unit. That is, the current corrector 38 has the function of detecting the speed error Q2 and the function of the adder A3 shown in FIG.

As described above, in the second embodiment, the first ΔΣ unit has the input speed command value (target speed) ωm ^* and the actual speed (mechanical angular speed) of the motor detected by the position/speed estimation unit. The velocity deviation (mechanical angular velocity deviation) Δω, which is the difference from ωm, is integrated and output, and the second ΔΣ unit is excited based on the output of the first ΔΣ unit with the speed controller 12 that is the current command value generating unit. The current command value I ^* (d-axis current command value Id ^* , q-axis current command value Iq ^* ) generated by the current controller 13 and the correction current I′ (here, three-phase fixed coordinates detected by the detection unit) A current correction value ΔIc is generated by the current corrector 38 from the d-axis current Id and the q-axis current Iq, which are obtained by converting the system current into the current of the two-phase rotating coordinate system, and the current is further calculated from the d-axis current Id and the q-axis current Iq. The difference ΔI (here, the d-axis current deviation ΔId and the q-axis current deviation ΔIq) from the corrected d-axis current Id′ and the corrected q-axis current Iq′ obtained by subtracting the correction value ΔIc is integrated and output to the drive unit.

(Modification of Embodiment 2)
In the second embodiment, the motor M is regarded as a frequency converter and the frequency conversion error is reduced. However, the present invention is not limited to this, and an error such as an output torque ripple that is an external factor of the motor M, a variation in electric characteristics, and other elements that may be connected to the subsequent stage of the quantizer Q in FIG. Various errors can be reduced similarly to the frequency conversion error by adding the errors in the control system as shown in FIG.

According to the second embodiment, in the configuration of the motor control device that performs vector control, the motor is regarded as a frequency converter, and the rotation speed fluctuation (speed fluctuation) due to load torque pulsation causes the rotation speed command value (target rotation speed, target speed). By adopting the ΔΣ modulation type, which is regarded as an error with, the speed fluctuation (frequency conversion error) due to the load torque pulsation of the motor can be reduced, and the accuracy of the entire vector control including the reduction of frequency conversion error can be improved. it can. For example, the motor control device 100B according to the second embodiment can reduce the quantization error Q1 of the element replaced with the quantizer Q shown in FIG. 9A and the frequency conversion error Q2 of the motor M. In addition, since only the integrator and the current corrector that do not increase the processing time are added to the configuration of the motor control device that performs vector control, the amount of calculation related to the conversion process and the responsiveness associated with the increase in the amount of calculation are increased. Does not cause a decrease in temperature and an increase in processing time.

The above-described embodiment and the specific names, processes, controls, and information including various data and parameters shown in the drawings are merely examples, and can be appropriately changed unless otherwise specified. Further, the configuration of each unit or each device in the above-described embodiments may be appropriately distributed and integrated in view of processing load, mounting efficiency, and the like.

The broader aspects of the embodiments described above are not limited to the particular details and representative embodiments described and described above. Therefore, various changes may be made without departing from the spirit or scope of the general inventive concept defined by the appended claims and their equivalents.

S1, S2 Subtractors I1, I2 Integrators A1, A2, A3 Adder Q Quantizer D Delay device FC Frequency converter 100X, 100A, 100B Motor controller 11, 14, 15, 19 Subtractor 12 Speed controller 13 Excitation current controller 16 d-axis current controller 17 q-axis current controller 18 Decoupling controller 20 Adder 21 dq/3φ converter 22 PWM generator 23 IPM
24 3φ current calculator 25 3φ/dq converter 26 axis error calculation processing unit 29 PLL controller 30, 30B position estimator 31 1/P _n processor 32, 33, 34 delay device 35, 36, 37 integrator 38 current Corrector 38a Subtractor 38b Adder 38c Correction current calculator 39a, 39b Delay device

Claims (3)

A motor control device that has a detection unit that detects the motor current, a position/speed estimation unit that detects the actual speed of the motor, and a drive unit that drives the motor, and that controls the speed of the motor based on the speed command value. And
The motor control device includes a first ΔΣ unit including a first subtractor and a first integrator, and a second ΔΣ unit including a second subtractor and a second integrator. ,
The first ΔΣ unit integrates and outputs the difference between the speed command value and the actual speed of the motor detected by the position/speed estimation unit, and the second ΔΣ unit outputs the integrated value of the first ΔΣ unit. A motor control device characterized by integrating a difference between a current command value generated based on an output and a motor current detected by the detection unit and outputting the result to the drive unit. Further comprising a current correction unit that generates a correction current for correcting the speed error of the motor based on the current of the motor detected by the detection unit,
The second ΔΣ unit subtracts the correction current generated by the current correction unit from the current command value generated based on the output of the first ΔΣ unit, integrates the current, and outputs the integrated value to the drive unit. The motor control device according to claim 1, wherein the motor control device is a motor control device. The current correction unit,
The difference between the mechanical angle phase of the motor estimated based on the current value of the motor and the actual speed of the motor estimated from the speed command value of the motor and the current value of the motor detected by the detection unit. The motor control device according to claim 2, wherein the correction current is generated from

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